INVESTIGATION OF HIGH-PERFORMANCE DC-DC CONVERTERS FOR PLUG-

Transcription

INVESTIGATION OF HIGH-PERFORMANCE DC-DC CONVERTERS FOR PLUGIN HYBRID ELECTRIC VEHICLE BATTERY CHARGERS
by
Deepak Gautam
M.A.Sc., University of Victoria, 2007
A THESIS SUBMITTED IN PARTIAL FULFILLMENT OF
THE REQUIREMENTS FOR THE DEGREE OF
DOCTOR OF PHILOSOPHY
in
THE FACULTY OF GRADUATE AND POSTDOCTORAL STUDIES
(Electrical and Computer Engineering)
THE UNIVERSITY OF BRITISH COLUMBIA
(Vancouver)
April 2014
© Deepak Gautam, 2014
Abstract
Plug-in Hybrid Electric Vehicles (PHEVs) and Electric Vehicles (EVs) are an emerging trend
in automotive circles, and consumer interest is growing rapidly. With the development of
PHEVs, battery chargers for automotive applications are becoming a large market for the
power supply industry. The most common charger power architecture includes an ac-dc
converter with power factor correction (PFC) followed by an isolated dc-dc converter. As a
key component of a charger system, the dc-dc converter must achieve high efficiency and
power density.
This dissertation mainly focuses on the dc-dc converter stage only and in order to meet high
efficiency, high power density and a cost-effective solution, various dc-dc topologies have
been investigated and proposed for battery charging application. In this research work two
new full-bridge dc-dc converter topologies (one with inductive and another with capacitive
output filter) operating with a trailing edge pulse width modulation (PWM) gating scheme
are investigated. Also for higher power (>2 kW) battery charging application, another two
new interleaved dc-dc converter topologies using full-bridge with capacitive output filter
(one with bridge rectifier diodes and another with voltage doubler rectifier) are also
investigated. Detailed operating principle and steady state analysis for different modes of
operation, step-by-step design procedure, simulation, experimental results and performance
evaluation with various semiconductor devices for each of these topologies are presented in
this thesis. The results show that the performance, in terms of efficiency, size and cost for the
full-bridge converter with capacitive output filter is superior to that with inductive output
filter. Moreover the dc-dc converter with capacitive output filter overcomes some of the
major issues such as high voltage ringing on the rectifier diodes and duty-cycle loss, which
are present in the converter with inductive output filter.
ii
Preface
I am the lead investigator for this research work, responsible for performing literature survey,
topology investigation, theoretical analysis, design, simulation and experimentation. This
work was done under the guidance of my thesis supervisor Dr. William G. Dunford of UBC,
Vancouver and co-supervisor Dr. Wilson Eberle of UBC, Okanagan campus. This work was
also supervised by Dr. Fariborz Musavi and Mr. Murray Edington of Delta-Q Technologies
Corp.
Chapter 1: In this chapter, most of the figures and tables are obtained from various sources
and they have been appropriately cited. Some sections of the chapter are also modified from
previously written introductory material from my master’s thesis entitled “Soft-Switched
DC-to-DC Converters for Power Conditioning of Electrolyzer in a Renewable Energy
System” (2006) completed at the University of Victoria.
Chapter 2: Content in this chapter has been published in:
[1] D.S. Gautam, Fariborz Musavi, Murray Edington, W. Eberle and W.G. Dunford, "An
Automotive On-Board 3.3 kW Battery Charger for PHEV application," Proceedings
of IEEE Vehicular Power and Propulsion Conference (VPPC 2011), Chicago, pp. 16, Sep. 2011
Automotive On-Board 3.3 kW Battery Charger for PHEV application," IEEE
Transactions on Vehicular Technology, vol. 61, no. 8, pp. 3466-3474, Oct. 2012
[3] D.S. Gautam, Fariborz Musavi, Murray Edington, W. Eberle and W.G. Dunford, "A
Zero Voltage Switching Full-Bridge DC-DC Converter for an On-Board PHEV
iii
Battery Charger," Proceedings of IEEE Transportation Electrification Conference
and Expo (ITEC 2012), Dearborn, pp. 1-6, Jun. 2012
Zero Voltage Switching Full-bridge DC-DC Converter with Capacitive Output Filter
for a Plug-in-Hybrid Electric Vehicle Battery Charger," Proceedings of IEEE Applied
Power Electronics Conference and Exposition (APEC 2012), Orlando, pp. 13811386, Feb. 2012
for a Plug-in-Hybrid Electric Vehicle Battery Charger," IEEE Transactions on Power
Electronics, vol. 28, no. 12, pp. 5728-5735, Dec. 2013
Interleaved Zero Voltage Switching Full-Bridge DC-DC Converter with Capacitive
Output Filter for a Plug-in-Hybrid Electric Vehicle Battery Charger," Proceedings of
IEEE Energy Conversion Congress and Exposition (ECCE 2012), Raleigh, pp. 28272832, September 2012
[7] D.S. Gautam, Fariborz Musavi, Murray Edington, W. Eberle and W.G. Dunford, "
An Isolated Interleaved DC-DC Converter with Voltage Doubler Rectifier for PHEV
Battery Charger, "Proceedings of IEEE Applied Power Electronics Conference and
Exposition (APEC 2013), Long Beach, pp. 3067-3072, Mar. 2013
iv
In all of the publications listed above [1-7], I was the lead investigator, responsible for
performing theoretical analysis, design, simulation and experimentation. Dr. Musavi was
involved in the early stages of concept formation and contributed to manuscript edits. Dr.
Dunford, Dr. Eberle and Mr. Edington were the supervisory authors and were involved
throughout the project in concept formation and manuscript composition.
The first page of each chapter includes footnotes with references to existing publications on
the related work.
v
Table of Contents
Abstract .................................................................................................................................... ii
Table of Contents ................................................................................................................... vi
List of Tables ......................................................................................................................... xii
List of Figures ....................................................................................................................... xiii
List of Abbreviations ........................................................................................................... xxi
Acknowledgements ............................................................................................................ xxiii
Dedication ........................................................................................................................... xxiv
Chapter 1: Introduction ........................................................................................................ 1
1.1
Plug-in Hybrid Electric Vehicle................................................................................ 1
1.1.1
Series Hybrid Electric Vehicle.............................................................................. 3
1.1.2
Parallel Hybrid Electric Vehicle ........................................................................... 4
1.2
Conductive Battery-Charging System Standard (SAE J1772TM) [8] ...................... 5
1.2.1
AC Level 1 Charging ............................................................................................ 6
1.2.2
AC Level 2 Charging ............................................................................................ 7
1.3
On-Board AC-DC Battery Charger .......................................................................... 7
1.4
DC-DC Converter for Battery Charging Application ............................................... 9
1.4.1
Hard-Switched Converter ..................................................................................... 9
1.4.2
Soft-Switched Converter ..................................................................................... 10
1.5
Interleaving of DC-DC Converters ......................................................................... 11
1.6
Literature Review on ZVS Soft-Switched DC-to-DC Converters .......................... 12
1.6.1
Fixed-Frequency Series Resonant Converter (SRC) .......................................... 13
vi
1.6.2
Fixed-Frequency Parallel Resonant Converter (PRC) ........................................ 14
1.6.3
Fixed-Frequency Series-Parallel or LCC Resonant Converter (SPRC) ............. 14
1.6.4
Fixed-Frequency LCL Series Resonant Converter (SRC) with a Capacitive
Output Filter .................................................................................................................... 14
1.6.5
Fixed-Frequency LCL SRC with an Inductive Output Filter ............................. 15
1.6.6
Fixed-Frequency Phase-Shifted ZVS PWM Full-Bridge Converter with
Inductive Output Filter .................................................................................................... 16
1.6.7
Capacitive Output Filter .................................................................................................. 19
1.7
Thesis Motivation ................................................................................................... 20
1.8
Thesis Outline ......................................................................................................... 21
Chapter 2: Full-Bridge DC-DC Converter with Inductive Output Filter Operated with
Trailing-Edge PWM Gating................................................................................................. 23
2.1
Introduction ............................................................................................................. 23
2.2
Operating Principle ................................................................................................. 24
2.2.1
Interval 1 (T0 – T1) .............................................................................................. 28
2.2.2
Interval 2 (T1 – T2) .............................................................................................. 29
2.2.3
Interval 3 (T2 – T3) .............................................................................................. 30
2.2.4
Interval 4 (T3 – T4) .............................................................................................. 31
2.2.5
Interval 5 (T4 – T5) .............................................................................................. 32
2.2.6
Interval 6 (T5 – T6) .............................................................................................. 33
2.3
2.3.1
Design Procedure .................................................................................................... 34
Selection of Switching Frequency (fs) ................................................................ 34
vii
2.3.2
Selection of Transformer Turns Ratio (nt) .......................................................... 35
2.3.3
Selection of Output Filter Inductor (Lo) .............................................................. 35
2.3.4
Selection of Resonant Inductor (Lr) .................................................................... 36
2.3.5
Selection of MOSFETs (Q1 – Q4) ....................................................................... 37
2.3.6
Selection of Rectifier Diodes (DR1 – DR4)........................................................... 37
2.3.7
Selection of Trailing-edge PWM Controller and MOSFET Gate Driver ........... 38
2.3.8
Selection of Output filter capacitor (Co2) .......................................................... 38
2.4
Experimental Results .............................................................................................. 39
2.5
Performance Evaluation .......................................................................................... 44
2.6
Conclusions ............................................................................................................. 45
Chapter 3: Full-Bridge DC-DC Converter with Capacitive Output Filter Operated
with Trailing-Edge PWM Gating ........................................................................................ 47
3.1
Introduction ............................................................................................................. 47
3.2
3.2.1
Interval 1 (T0 – T1) .............................................................................................. 53
3.2.2
Interval 2 (T1 – T2) .............................................................................................. 53
3.2.3
Interval 3 (T2 – T3) .............................................................................................. 55
3.2.4
Interval 4 (T3 – T4) through Interval 6 (T5 – T6) ................................................. 56
3.3
3.3.1
Selection of Operating Mode .............................................................................. 57
3.3.2
Selection of Switching Frequency (fs) ................................................................ 58
3.3.3
3.3.4
viii
3.3.5
Selection of MOSFETs (Q1 – Q4) ....................................................................... 61
3.3.6
3.3.7
Selection of Output filter capacitor (Co2) ............................................................ 62
3.3.8
Selection of Trailing-edge PWM Controller and MOSFET Gate Driver ........... 62
3.4
Simulation and Experimental Results ..................................................................... 63
3.5
3.6
Conclusions ............................................................................................................. 70
Chapter 4: An Interleaved Full-Bridge DC-DC Converter with Capacitive Output
Filter Operated with Trailing-Edge PWM Gating ............................................................ 71
4.1
Introduction ............................................................................................................. 71
4.2
4.3
4.3.1
HF Transformer Design ...................................................................................... 76
4.3.2
Selection of Output Filter Capacitor (Co2) .......................................................... 76
4.4
Simulation and Experimental Results ..................................................................... 78
4.5
4.6
Conclusions ............................................................................................................. 84
Chapter 5: An Interleaved, Full-Bridge DC-DC Converter with Voltage-Doubler
Rectifier and Capacitive Output Filter Operated with Trailing-Edge PWM Gating .... 86
5.1
Introduction ............................................................................................................. 86
5.2
5.3
5.3.1
ix
5.3.2
5.3.3
5.3.4
Selection of Output Filter Capacitors (Co1 and Co2) ........................................... 99
5.4
Simulation and Experimental Results ................................................................... 100
5.5
Performance Evaluation ........................................................................................ 106
5.6
Conclusions ........................................................................................................... 107
Chapter 6: Conclusion and Future Work ........................................................................ 109
6.1
Introduction ........................................................................................................... 109
6.2
Summary of Contributions .................................................................................... 109
6.2.1
DC-DC Converter with Inductive Filter Operated with Trailing-Edge PWM
Gating 109
6.2.2
DC-DC Converter with Capacitive Filter Operated with Trailing-Edge PWM
Gating 110
6.2.3
Interleaved DC-DC Converter with Capacitive Filter Operated with Trailing-
Edge PWM Gating ........................................................................................................ 111
6.2.4
Interleaved DC-DC Converter with Capacitive Filter and Voltage Doubler
Rectifier......................................................................................................................... 111
6.2.5
6.3
Comparison of Proposed Topologies ................................................................ 112
6.3.1
Suggestions for Future Work ................................................................................ 113
Full-Bridge DC-DC Converter with Clamp Diodes to Reduce Rectifier Ringing
Issues 113
6.3.2
Full-bridge DC-DC Converter with Lossless Snubber ..................................... 113
x
6.3.3
Feedback Control Analysis for the Interleaved DC-DC Converter with
Capacitive Output Filter ................................................................................................ 114
Bibliography ........................................................................................................................ 115
xi
List of Tables
Table 1.1
Charge method electrical ratings (North America) [8] .......................................... 5
Table 1.2
Power levels of on-board battery chargers [11] ..................................................... 8
Table 1.3
Major Specifications of dc-dc converter .............................................................. 11
Table 2.1 Design specification of the Trailing-edge PWM Full-bridge dc-dc converter ..... 34
Table 3.1
Design specification of the Trailing-edge PWM Full-bridge dc-dc converter with
capacitive filter........................................................................................................................ 57
Table 3.2
Comparison of various parameters obtained from simulation and analysis at 5.5
A and 0.7 A load current and 400 V input voltage ................................................................. 63
Table 3.3
Components Used In the Benchmark Converter ................................................. 69
Table 4.1
Design specification of the Trailing-edge PWM Full-bridge dc-dc converter .... 76
Table 4.2
Components Selection ......................................................................................... 77
Table 4.3 Components Used In the Benchmark Converter ................................................. 84
Table 5.1
Design specification of the Trailing-edge PWM Full-bridge dc-dc converter .... 96
Table 5.2
Components Selection ......................................................................................... 99
Table 6.1
Performance comparison of the proposed dc-dc converter topologies .............. 112
xii
List of Figures
Figure 1.1
Typical diagram of a Plug-in Hybrid Electric Vehicle ......................................... 2
Figure 1.2
Typical layout of a series HEV drive train [4] ..................................................... 3
Figure 1.3
Schematic of a parallel HEV drive train configuration [4] .................................. 4
Figure 1.4
AC Level 1 System Configuration [8].................................................................. 6
Figure 1.5
AC Level 2 System Configuration [8].................................................................. 7
Figure 1.6
Simplified block diagram of an ac-dc battery charger ......................................... 8
Figure 1.7
Turn-on and turn-off transition in a hard-switched converter ............................ 10
Figure 1.8
Zero Voltage Switching (ZVS) .......................................................................... 10
Figure 1.9
Zero Current Switching (ZCS) ........................................................................... 11
Figure 1.10
Clamped Mode Series Resonant converter circuit [30] .................................... 13
Figure 1.11
Clamped-Mode Parallel Resonant converter [31] ............................................ 14
Figure 1.12
Fixed-Frequency Series-Parallel Resonant converter ...................................... 14
Figure 1.13
Fixed Frequency LCL SRC with capacitive output filter [33],[34] ................. 15
Figure 1.14
Fixed Frequency LCL SRC with inductive output filter [35] .......................... 15
Figure 1.15
Full-bridge Phase-shifted converter with inductive output filter [36]-[38] ...... 16
Figure 1.16
Trailing-edge PWM gating scheme .................................................................. 18
Figure 1.17
Full-bridge Phase-shifted converter with capacitive output filter [39]-[42] .... 19
Figure 2.1
Trailing-edge PWM Full-bridge dc-dc converter with inductive output filter ... 24
Figure 2.2
Trailing-edge PWM gating scheme .................................................................... 25
Figure 2.3
Typical operating waveforms for an arbitrary pulse width ‘δ’ to illustrate the
operation of the trailing-edge PWM full-bridge converter ..................................................... 27
Figure 2.4
Equivalent circuit for Interval 1 (T0 – T1) .......................................................... 28
xiii
Figure 2.5
Figure 2.6
Figure 2.7
Figure 2.8
Figure 2.9
Figure 2.10
Comparison of measured efficiency as a function of output power for different
switching frequencies at Vo = 300V and Po = 1.65 kW ......................................................... 35
Figure 2.11
Prototype unit of trailing-edge PWM full-bridge converter with inductive
output filter.............................................................................................................................. 39
Figure 2.12
Measured efficiency versus output power at different output voltages with Vin =
400 V ....................................................................................................................................... 39
Figure 2.13
Experimental waveforms of output voltage and current Ch1= Vo 100V/div.
Ch4= Io 2A/div. ....................................................................................................................... 40
Figure 2.14
Experimental waveforms obtained for (Ch1) Q3 gating signal, Vg3 (Ch2) Q3
drain to source voltage, VDSQ3 (Ch3) Transformer secondary current, Isec (Ch4) Rectifier
output voltage, Vrectout at light-load (150 W) with Vin = 400 V and Vo = 300 V ..................... 41
Figure 2.15
Experimental waveforms of Figure 2.14 repeated for half-load (1.65 kW) with
Vin = 400 V and Vo = 300 V .................................................................................................... 41
Figure 2.16
Experimental waveforms of Figure 2.14 repeated for full-load (3.3 kW) with
Vin = 400 V and Vo = 300 V .................................................................................................... 42
Figure 2.17
Experimental waveforms of MOSFET Q1 voltage and current during Turn-ON
at Vo = 300 V and Io = 11 A .................................................................................................... 43
xiv
Figure 2.18
at Vo = 300 V and Io = 11 A .................................................................................................... 43
Figure 2.19
at Vo = 300 V and Io = 1 A ...................................................................................................... 44
Figure 2.20
Measured Efficiency comparison with different combination of primary
MOSFETs and secondary diodes at Vo = 300 V and Io = 11 A ............................................. 44
Figure 3.1
Trailing-edge PWM Full-bridge dc-dc converter with capacitive output filter . 48
Figure 3.2
Typical operating waveforms to illustrate the operation of the trailing-edge
PWM full-bridge converter in DCM mode ............................................................................. 50
Figure 3.3
PWM full-bridge converter in BCM mode ............................................................................. 51
Figure 3.4
PWM full-bridge converter in CCM mode ............................................................................. 52
Figure 3.5
Equivalent circuit for Interval 1 (T0-T1) for DCM, BCM and CCM.................. 53
Figure 3.6
Equivalent circuit for Interval 2 (T1 – T2) for DCM, BCM and CCM and Interval
3 (T2 – T3) for BCM ................................................................................................................ 53
Figure 3.7
Equivalent circuit for Interval 3 (T2 – T3) for CCM ........................................... 55
Figure 3.8
Equivalent circuit for Interval 4 (T3 – T4) for DCM, BCM and CCM ............... 56
Figure 3.9
Equivalent circuit for Interval 5 (T4 – T5) DCM, BCM and CCM and Interval 6
(T5 – T6) for BCM ................................................................................................................... 56
Figure 3.10
Equivalent circuit for Interval 6 (T5 – T6) for CCM ......................................... 57
Figure 3.11
Comparison of measured efficiency as a function of output power for different
switching frequencies at Vo = 300V and Po = 1.65 kW .......................................................... 59
xv
Figure 3.12
Design Curve obtained for Gain versus Duty cycle for various values of k in
DCM and BCM ....................................................................................................................... 60
Figure 3.13
Experimental prototype of 1.65 kW ZVS full-bridge dc-dc converter with
capacitive output filter ............................................................................................................ 64
Figure 3.14
Experimental measurement of efficiency of the proposed converter as a
function of output power at 400 V input and different output voltages .................................. 64
Figure 3.15
Experimental waveforms of output voltage and current Ch1= Vo 100 V/div.
Ch4= Io 2 A/div. ...................................................................................................................... 65
Figure 3.16
Experimental waveforms of the MOSFET Q3 voltage and resonant inductor Lr
current at Vin = 400 V, Vo = 300 V, Po = 200 W and fs = 100 kHz. Ch1=VDS-Q3 200 V/div.
Ch2= iLr 5 A/div. Ch3= VGS-Q3 10 V/div. Time scale=1.16 µs/div. ........................................ 66
Figure 3.17
Experimental waveforms of Figure 3.16 repeated for half-load (800 W) with
Vin = 400 V and Vo = 300 V .................................................................................................... 66
Figure 3.18
Experimental waveforms of Figure 3.16 repeated for full-load (1.65 kW) with
Vin = 400 V and Vo = 300 V .................................................................................................... 67
Figure 3.19
Proposed converter experimental waveforms of the diode DR3 voltage and
current at Vin = 400 V, Vo = 300 V, Po = 200 W and fs = 100 kHz. Ch1=VDR3 200 V/div.
Ch2= IDR3 5 A/div. Time scale=900 ns/div. ............................................................................ 67
Figure 3.20
Experimental waveforms of the diode DR3 voltage and current at Vin = 400 V,
Vo = 300 V, Po = 1650 W and fs = 100 kHz. Ch1=VDR3 100 V/div. Ch2= IDR3 5 A/div. Time
scale=900 ns/div...................................................................................................................... 68
xvi
Figure 3.21
Efficiency comparison for the proposed converter as a function of output
power at 400 V input and 300V output voltage for different rectifier diodes and benchmark
converter ................................................................................................................................. 69
Figure 3.22
Schematic of the benchmark ZVS full-bridge converter with inductive output
filter ......................................................................................................................................... 69
Figure 4.1
A 2-cell interleaved trailing-edge PWM full-bridge converter with capacitive
output filter.............................................................................................................................. 72
Figure 4.2
PWM 2-cell, interleaved, full-bridge converter in DCM mode .............................................. 73
Figure 4.3
PWM 2-cell, interleaved, full-bridge converter in BCM mode .............................................. 74
Figure 4.4
Simulation results of resonant inductor LrA and LrB with current through the
output filter capacitor Co2 at Vin = 400 V and Vo = 300 V and Io = 1 A .................................. 78
Figure 4.5
Simulation results of Figure 4.4 repeated at at Vin = 400 V and Vo = 300 V and Io
= 11 A ..................................................................................................................................... 78
Figure 4.6
Experimental prototype of 3.3 kW, 2-cell, interleaved, full-bridge dc-dc
converter with capacitive output filter .................................................................................... 79
Figure 4.7
An inner-loop, current-sharing control scheme .................................................. 80
Figure 4.8
Experimental measured efficiency of the proposed converter as a function of
output power at 400 V input and different output voltages .................................................... 81
Figure 4.9
Experimental waveforms of the MOSFET Q3B voltage and transformer
secondary winding current at Vin = 400 V, Vo = 300 V, Po = 300 W and fs = 100 kHz.
xvii
Ch1=VDS-Q3B 200 V/div. Ch2= VGS-Q3 10 V/div. Ch3= Tx. B Sec. current 2 A/div. Ch4= Tx.
A Sec. current 2 A/div.Time scale=2 µs/div. .......................................................................... 82
Figure 4.10
Experimental waveforms of Figure 4.10 repeated for Vin = 400 V, Vo = 300 V,
Po = 3300 W and fs = 100 kHz. Ch1=VDS-Q3B 200 V/div. Ch2= VGS-Q3 10 V/div. Ch3= Tx. B
Sec. current 10 A/div. Ch4= Tx. A Sec. current 10 A/div.Time scale=2 µs/div. ................... 82
Figure 4.11
power at 400 V input and 300V output voltage and benchmark converter ............................ 83
Figure 4.12
Benchmark 2-Cell Interleaved PWM ZVS full-bridge converter topology with
inductive output filter .............................................................................................................. 84
Figure 5.1
Trailing-edge PWM Full-bridge dc-dc converter with voltage-doubler rectifier
and capacitive output filter...................................................................................................... 87
Figure 5.2
PWM full-bridge converter with voltage doubler-rectifier in DCM mode ............................. 88
Figure 5.3
PWM full-bridge converter with voltage-doubler rectifier in BCM mode ............................. 89
Figure 5.4
Equivalent circuit for Interval 1 for DCM and BCM ......................................... 90
Figure 5.5
Equivalent circuit for Interval 2 for DCM and Interval 2 and 3 for BCM ......... 91
Figure 5.6
A 2-cell, interleaved, trailing-edge PWM, full-bridge converter with voltage-
doubler rectifier and capacitive output filter ........................................................................... 92
Figure 5.7
PWM, 2-cell, interleaved, full-bridge converter with voltage-doubler rectifier in DCM mode
................................................................................................................................................. 93
xviii
Figure 5.8
PWM, 2-cell, interleaved, full-bridge converter with voltage-doubler rectifier in BCM mode
................................................................................................................................................. 94
Figure 5.9
Design Curve obtained for Gain versus Duty cycle for various values of k in
DCM and BCM ....................................................................................................................... 97
Figure 5.10
Output filter capacitor C01 and C02 ................................................................... 99
Figure 5.11
Simulation results of resonant inductor LrA and LrB with current through the
output filter capacitors Co1 and Co2 at Vin = 400 V and Vo = 300 V and Io = 1 A ................. 100
Figure 5.12
Simulation results of Figure 5.11 repeated at Vin = 400 V and Vo = 300 V and Io
= 11 A ................................................................................................................................... 100
Figure 5.13
Simulation results of voltage across and current through output rectifier diodes
DR2A and DR2B at Vin = 400 V and Vo = 300 V and Io = 1 A ............................................ 101
Figure 5.14
Simulation results of Figure 5.13 repeated at Vin = 400 V and Vo = 300 V and Io
= 11 A ................................................................................................................................... 101
Figure 5.15
Experimental prototype of 3.3 kW 2-cell interleaved full-bridge dc-dc
converter with voltage-doubler rectifier and capacitive output filter ................................... 102
Figure 5.16
An inner-loop current-sharing control scheme ............................................... 102
Figure 5.17
Experimental measured efficiency of the proposed converter as a function of
output power at 400 V input and different output voltages .................................................. 103
Figure 5.18
Experimental waveforms of current through resonant inductor LRA and LRB
and MOSFET Q3B voltage at Vin = 400 V and Vo = 300 V, Po = 300 W and fs = 100 kHz.
Ch1=VDS-Q3B 200 V/div. Ch2= VGS-Q3 10 V/div. Ch3= Resonant inductor LrA current 5 A/div.
Ch4= Resonant inductor LrB 5 A/div. Time scale=2 µs/div.................................................. 104
xix
Figure 5.19
Experimental waveforms of Figure 5.18 repeated for Vin = 400 V and Vo = 300
V, Po = 3300 W and fs = 100 kHz. Ch1=VDS-Q3B 200 V/div. Ch2= VGS-Q3 10 V/div. Ch3=
Resonant inductor LrA current 10 A/div. Ch4= Resonant inductor LrB 10 A/div. Time scale=2
µs/div. ................................................................................................................................... 105
Figure 5.20
Experimental waveforms of current through resonant inductor LrB and
transformer B secondary winding and voltage across diode DR2B at Vin = 400 V and Vo = 300
V, Po = 300 W and fs = 100 kHz. Ch1= VDR2B 100 V/div. Ch3= Tx. B Sec. winding current 5
A/div. Ch4= LRB current 5 A/div. Time scale=2 µs/div. ...................................................... 105
Figure 5.21
Experimental waveforms of Figure 5.20 repeated for Vin = 400 V and Vo = 300
V, Po = 3300 W and fs = 100 kHz. Ch1= VDR2B 100 V/div. Ch3= Tx. B Sec. winding current
20 A/div. Ch4= LRB current 10 A/div. Time scale=2 µs/div. ............................................... 106
Figure 5.22
power at 400 V input and 300V output voltage and benchmark converter .......................... 106
Figure 6.1
Trailing-edge PWM Full-bridge dc-dc converter with inductive output filter and
clamp diodes ......................................................................................................................... 113
xx
List of Abbreviations
AC, ac
Alternating Current
AER
All Electric Range
BCM
Boundary Conduction Mode
CCM
Continuous Conduction Mode
DC, dc
Direct Current
DCM
Discontinuous Conduction Mode
Div.
Division
EMI, emi
Electro-Magnetic Interference
ESR, esr
Equivalent Series Resistance
ESS
Energy Storage System
EV
Electric Vehicle
HF
High Frequency
IGBT
Insulated Gate Bipolar Transistor
MOSFET
Metal Oxide Semiconductor Field Effect Transistor
PHEV
Plug-in Hybrid Electric Vehicle
PRC
Parallel Resonant Converter
PWM
Pulse Width Modulation
RC
Resistor Capacitor
RMS, rms
Root Mean Square
SiC
Silicon Carbide
SPRC
Series-Parallel Resonant Converter
SRC
Series Resonant Converter
xxi
VA
Volt-Ampere
ZCS
Zero Current Switching
ZVS
Zero Voltage Switching
ZVT
Zero Voltage Transition
Prefixes for SI Units
G
Giga (109)
k
Kilo (103)
M
Mega (106)
m
Milli (10-3)
n
Nano (10-9)
p
Pico (10-12)
μ
Micro (10-6)
SI Units
A
Amperes
C
Coulombs
F
Farads
H
Henries
Hz
Hertz
s
seconds
V
Volts
W
Watts
Ω
Ohms
°
Degrees
xxii
Acknowledgements
I would like to thank my supervisors, Dr. William G. Dunford and Dr. Wilson Eberle, for
their valuable guidance during the course of this research work, preparation of the thesis and
for financial support.
I would like to also thank Mr. Murray Edington and Dr. Fariborz Musavi of Delta-Q
Technologies Corp. for their valuable guidance and financial support.
I would like to express my sincere appreciation to the members of my supervisory committee
for their time and suggestions.
I am grateful to Mr. David Mitalpi and Mr. Ryan Truss from Delta-Q Technologies for their
help in preparation of boards for building experimental laboratory prototypes.
I also appreciate the great support of my colleagues Mr. Jon Stroud, Mr. Marian Craciun, Mr.
Dan O’Leary and Mr. Dale Wager at Delta-Q Technologies Corp.
Finally, I appreciate the love and wonderful support from my parents, wife, son and friends.
This work was supported by grants from NSERC, Canada and Delta-Q Technologies Corp.,
Canada.
xxiii
Dedication
This thesis is dedicated to my parents who have supported me all the way since the beginning
of my studies.
Also, this thesis is dedicated to my wife Annalakshmi, who has been a great source of
motivation and inspiration.
And finally I would like to dedicate this dissertation to my son, Akash, who has grown into a
wonderful three-and-a-half-year-old kid in spite of his dad’s spending so much time away
from him while working on this dissertation.
xxiv
Chapter 1: Introduction1
This chapter presents literature review of soft-switched dc-dc converters used in a two-stage
AC-DC on-board battery charger for plug-in hybrid electric vehicle (PHEV) application.
The outline of this chapter is as follows. Section 1.1 briefly discusses the powertrain
components of a PHEV. Sections 1.2 and 1.3 present the conductive charging system
standard (SAE J1772TM) and topology considerations for an on board battery charger.
Sections 1.4 and 1.5 briefly discuss High–Frequency (HF) switching and interleaved dc-dc
converters for battery charger application. Section 1.6 presents the literature survey on softswitched dc-to-dc converters suitable for the desired application. Thesis motivation and
outline are given in Sections 1.7 and 1.8, respectively.
1.1
Plug-in Hybrid Electric Vehicle
A PHEV as shown in Figure 1.1 is a hybrid vehicle that utilizes rechargeable batteries, or
another energy storage device, that can be restored to full charge by connecting a plug to an
external electric power source (usually a normal electric wall socket). A PHEV shares the
characteristics of both a conventional hybrid electric vehicle, having an electric motor and an
Internal Combustion Engine (ICE); and of an all-electric vehicle, having a plug to connect to
the electrical grid. Most PHEVs on the road today are passenger cars, but there are also
PHEV versions of commercial vehicles and vans, utility trucks, buses, trains, motorcycles,
scooters, and military vehicles.
1
Most of the figures and tables used in this chapter are obtained from various sources and they have been
appropriately cited. Some sections of the chapter are also modified from previously written introductory
material from my master’s thesis entitled “Soft-Switched DC-to-DC Converters for Power Conditioning of
Electrolyzer in a Renewable Energy System” (2007) completed at the University of Victoria.
1
Source: Argonne National Laboratory
Figure 1.1
Typical diagram of a Plug-in Hybrid Electric Vehicle
Living in the era of increasing environmental sensibility and rising fuel price makes it
necessary to develop a generation of vehicles that are more fuel efficient and environmental
friendly. Hybrid electric vehicles could meet these demands [1]. Plug-in hybrid vehicles have
recently created interest among leading automotive industry manufactures because of their
potential to replace fuel-generated energy with battery-stored electricity in short daily
journeys, and also continuing extended range as a HEV afterwards. This feature makes
PHEV a very low or zero-emission vehicle during its Charge Depletion or All-Electric Range
(AER) [2].
One of the unique advantages of PHEVs is its capability to integrate the transportation and
electric power generation sectors in order to improve the efficiency, fuel economy, and
reliability of both systems. This goal is performed via integration of the onboard energy
storage units of plug-in vehicles with the power grid by power electronic converters and
communication systems. Employing energy storage systems improves the efficiency and
reliability of the electric power generation, transmission, and distribution. Similarly by
combining Energy Storage System (ESS) with the power train of a conventional vehicle can
result in a hybrid vehicle with higher fuel efficiency [3].
2
The operational characteristics of various Hybrid Electric Vehicle (HEV) topologies such as
series hybrid, parallel hybrid and series-parallel hybrid systems are presented in [4]-[6]. Of
these above mentioned topologies the series hybrid and parallel hybrid are the basic types of
HEV topologies primarily considered for PHEV application and they are discussed below:
1.1.1
Series Hybrid Electric Vehicle
Series HEV known as Extended Range Electric Vehicles is shown in Figure 1.2. An ICE is
generally run at an optimal efficiency point to drive the generator and charge the propulsion
batteries on-board the vehicle, as shown in Figure 1.2.
Figure 1.2
Typical layout of a series HEV drive train [4]
When the state of charge (SOC) of the battery is at a predetermined minimum, the ICE is
turned on to charge the battery. The ICE turns off again when the battery has reached a
desirable maximum SOC. It must be noted that, in a series HEV, there is no mechanical
connection between the ICE and the wheels. A series hybrid vehicle is more applicable in
city driving and can run solely on electricity until the battery needs to be recharged. For
shorter trips, these vehicles might not use gasoline at all.
3
1.1.2
Parallel Hybrid Electric Vehicle
Parallel or Blended HEVs has both the ICE and the traction motor mechanically connected to
the transmission. A schematic diagram of the parallel hybrid is shown in Figure 1.3.
Figure 1.3
Schematic of a parallel HEV drive train configuration [4]
The vehicle can be driven with the ICE, or the electric motor, or both at the same time and,
therefore, it is possible to choose the combination freely to feed the required amount of
torque at any given time. In parallel HEVs, there are many ways to configure the use of the
ICE and the traction motor. The most widely used strategy is to use the motor alone at low
speeds, since it is more efficient than the ICE, and then let the ICE work alone at higher
speeds. When only the ICE is in use, the traction motor can function as a generator and
charge the battery pack.
The fuel economy and AER of HEVs are highly dependent on the onboard ESS of the
vehicle. Energy-storage devices charge during low power demands and discharge during high
power demands, acting as catalysts to provide energy boost. Batteries are the primary energystorage devices in ground vehicles. Increasing the AER of vehicles by 15% almost doubles
the incremental cost of the ESS. This is due to the fact that the ESS of HEVs requires higher
peak power while preserving high energy density. Ultra capacitors (UCs) are the options with
higher power densities in comparison with batteries. A hybrid ESS composed of batteries,
4
UCs, and/or fuel cells (FCs) could be a more appropriate option for advanced hybrid
vehicular ESSs. The state-of-the-art energy-storage topologies for HEVs and plug-in HEVs
(PHEVs) along with battery, UC, and FC technologies are discussed and compared in [7].
1.2
Conductive Battery-Charging System Standard (SAE J1772TM) [8]
PHEV motor drive and energy storage technology is developing at a rapid rate in response to
expected market demand for PHEVs. Battery chargers are another key component required
for the emergence and acceptance of PHEVs. SAE J1772 defines conductive charging
methods and electrical interfaces required for EV/PHEV battery charging. Conductive
charging is a method for connecting the electric power supply network to the EV/PHEV for
the purpose of transferring energy to charge the battery and operate other vehicle electrical
systems, establishing a reliable equipment grounding path, and exchanging control
information between the EV/PHEV and the supply equipment [8].
Table 1.1
Charge method electrical ratings (North America) [8]
Charge Method
AC Level 1 (on-board)
AC Level 2 (on-board)
*
DC Level 1 (off-board)
DC Level 2 (off-board)
*
Not finalized
*
Nominal Supply
Voltage (Volts)
120 V AC, 1-phase
120 V AC, 1-phase
208 to 240 V AC, 1phase
200‐450 V DC
200‐450 V DC
Maximum Current
(Amps-continuous)
12 A
16A
≤ 80 A
Branch Circuit Breaker
rating (Amps)
15 A (minimum)
20 A
Per NEC 625
80 A (up to 36 kW)
200 A (up to 90 kW)
-
There are three basic functions (2 electrical and 1 mechanical) that must be performed to
allow charging of the EV/PHEV battery from the electric supply network. The first electrical
function is to convert the AC voltage to DC voltage which is commonly referred to as
rectification. The second electrical function is the control or regulation of the supply voltage
to a level that permits a managed charge rate based on the battery charge acceptance
characteristics – i.e., voltage, capacity, electrochemistry, and other parameters. The
5
mechanical function is the physical coupling or connecting of the EV/PHEV to the EVSE
(Electric Vehicle Supply Equipment) and is performed by the user. Thus a conductive
charging system consists of a battery charger and a coupler. The conductive system
architecture is suitable for use with electrical ratings as specified in Table 1.1.
1.2.1
AC Level 1 Charging
AC level 1 charging is a method of EV/PHEV charging that extends AC power from the
most common grounded electrical receptacle to an onboard charger using an appropriate cord
set, as shown in Figure 1.4 at the electrical ratings specified in Table 1.1. AC level 1 allows
connection to existing electrical receptacles in compliance with the National Electrical Code
- Article 625 [9].
Figure 1.4
AC Level 1 System Configuration [8]
The level 1 method uses a standard 120 VAC, 15 A (12 A useable) or 20 A (16 A useable)
branch circuit that is the lowest common voltage level found in both residential and
commercial buildings in the North America. Thus the maximum power supplied by level 1 is
6
1.92 kW and is deemed important due to the availability of 120 VAC outlets in an emergency
situation, even if it meant waiting several hours to obtain a charge [10].
1.2.2
AC Level 2 Charging
AC level 2 charging is a method of EV/PHEV charging that extends AC power from the
electric supply to an on-board charger from a dedicated EVSE as shown in Figure 1.5. The
electrical ratings are similar to large household appliances and specified in Table 1.1. AC
level 2 may be utilized at home, workplace, and public charging facilities; and the maximum
power supplied by level 2 is 19.2 kW (available from a 240 VAC and maximum 80 A outlet).
Figure 1.5
1.3
AC Level 2 System Configuration [8]
On-Board AC-DC Battery Charger
As presented in the previous section for PHEV’s, the accepted approach involves using an on
board battery charger for AC level 1 and level 2 charging systems.
7
Table 1.2 lists the various power levels of on-board battery chargers being considered for
level 1 and level 2 charging systems.
Table 1.2
Power levels of on-board battery chargers [11]
Charge
Method
Nominal AC Supply
Voltage (Volts)
Level 1
120
Level 2
240
Maximum AC Current
(Amps-continuous)
12
15
15
30
50
80
Charger DC Output
Power Rating (W)
1200
1650
3300
6600
12000
18000
The accepted charger power architecture includes an ac-dc converter [12]-[15] with power
factor correction (PFC) [16]-[19] followed by an isolated dc-dc converter as shown in Figure
1.6.
DC Link Bus
Capacitors
Universal
AC Input
Voltage
AC Input
Filter
AC-DC
PFC Boost
Converter
Isolated
DC-DC
Converter
DC Output
Filter
DC Output
for Battery
Charging
DSP/Micro Controller
Figure 1.6
Simplified block diagram of an ac-dc battery charger
The ac-dc plus PFC stage rectifies the input ac voltage, boosts it to a regulated intermediate
dc link bus (example 400 VDC) and also maintains unity power factor. Galvanic isolation is
required in the battery charger for meeting the user safety (UL2202) and regulatory
requirements [20]. Isolation is implemented in the dc-dc stage to take advantage of a smallsize, high-frequency transformer, which is commonly used in a dc-dc converter to step-up or
step-down the output dc voltage for charging batteries. One of the main advantages of this
two-stage approach is that the low frequency ac ripple can be easily rejected by the second
dc-dc stage, which is very favorable for charging lithium ion batteries.
8
1.4
DC-DC Converter for Battery Charging Application
The main purpose of the dc-dc converter stage in an on-board battery charger is listed below:
(1) Provide galvanic isolation between the primary ac circuits and secondary side PHEV
vehicle components to meet various safety and regulatory requirements.
(2) Step-up or step-down the intermediate PFC bus link voltage of 400V as required for
charging the PHEV battery pack.
(3) Regulate output voltage and current of the battery charger as required by the batterycharging algorithm.
A dc-dc converter operating at high switching frequency (> 20 kHz) reduces the size, weight,
and cost of the converter [21]. High-frequency (HF) switched dc-dc converters are basically
classified as hard-switched and soft-switched converters.
1.4.1
Hard-Switched Converter
Typical current, voltage and switching loss power waveforms during the turn-on and turn-off
transitions in a hard-switched converter are shown in Figure 1.7. The voltage and current is
simultaneously present across the switch during both switching intervals. This results in large
a power loss and thus requires large a heatsink. Therefore the switching frequency range is
limited, as it is directly proportional to switching losses. At lower switching frequencies, the
size of magnetic components and filters become large. Lossy RC snubbers are also needed to
protect the switch from large di/dt and dv/dt. Due to the parasitics (inductance and
capacitance) of the circuit, EMI (Electro Magnetic Interference) is also generated, which
needs additional filtering.
9
Turn-On
Transition
Turn-Off
Transition
vsw
Vs
isw
t
t
Turn-On
Power Loss
Figure 1.7
1.4.2
Turn-Off
Power Loss
Turn-on and turn-off transition in a hard-switched converter
Soft-Switched Converter
As seen in Figure 1.7, hard-switched converter switching losses occur during the turn-on and
turn-off transition of the semiconductor switch (MOSFET or IGBT), and these losses
increase with switching frequency. Soft-switching techniques can be used in dc-to-dc
converters to reduce switching losses without reducing the switching frequency. Softswitching techniques usually refer to zero voltage switching (ZVS) (Figure 1.8) and zero
current switching (ZCS) (Figure 1.9), which reduces the turn-on losses and turn-off losses
respectively. Another advantage is that EMI generated is significantly reduced, which eases
filter design and allows the converter to be switched at a higher frequency.
I switch
Vswitch
+
Vgate
Cs
I switch
Vgate
Figure 1.8
Vswitch
-
Zero Voltage Switching (ZVS)
10
Vgate
Vswitch
Iswitch
Figure 1.9
Zero Current Switching (ZCS)
As shown in Figure 1.8, ZVS can be achieved by forward biasing the anti-parallel diode of
the semiconductor switch prior to applying gating signal to turn-on the switch and similarly
ZCS can be achieved by reducing the current through the switch to zero prior to turning-off
the gating signal. If a converter operates with ZVS, then the turn-off losses can be easily
reduced by placing a lossless snubber (capacitor) directly across the switch. By doing this,
the switches are naturally protected from large di/dt at turn-on with the help of ZVS and from
large dv/dt with lossless snubber capacitor. Therefore ZVS operation is mainly considered in
this research.
1.5
Interleaving of DC-DC Converters
The dc-dc converter for this application has to be designed for the power levels as presented
in Table 1.2. Table 1.3, lists the major specifications to be considered while designing the dcdc converter stage for the power levels presented in Table 1.2.
Table 1.3
Major Specifications of dc-dc converter
Input Voltage Range
(VDC)
Output Voltage
Range (VDC)
380 to 420
150 to 450
Max. Output
Current (A)
4
5.5
11
22
40
60
Max. Output Power at 300
VDC output voltage (W)
1200
1650
3300
6600
12000
18000
11
Due to the high power requirement (mainly for 2 kW and higher power levels), an
interleaved, multi-cell configuration [22]-[25] that uses ‘n’ number of cells (each cell rated at
(maximum output power/n)) in parallel (both at the input and output) with each cell being
phase shifted by 360o/n could be adopted. Each cell shares equal power and the thermal
losses are distributed uniformly among the cells. Also, the input/output ripple frequency of
multi-cell configuration becomes ‘n’ times the input/output ripple frequency of each cell
[26].
1.6
Literature Review on ZVS Soft-Switched DC-to-DC Converters
There are three major types of HF transformer isolated soft-switching converter
configurations possible [27]: (a) Voltage fed resonant converters [28-35]; (b) current fed
resonant converters [28]; and (c) fixed-frequency resonant transition zero-voltage switching
(ZVS) PWM bridge converters [36]-[38]. The current fed resonant converters require high
frequency switches rated at five to six times the input voltage (reducing the efficiency) in the
present application and therefore they are not considered further. Voltage fed resonant
converters can be operated either in variable frequency mode or fixed frequency mode. But
the operation in variable frequency mode suffers from several disadvantages: wide variation
in switching frequency (considering the input and output voltage variation) making the
design of filters and control (feedback and protection) circuit difficult and complex.
Therefore, fixed frequency operation is adopted in this thesis.
From the above discussions, we are left with the following seven soft-switching converter
configurations for the PHEV battery charging application.
(1) Fixed-frequency series resonant converter (SRC) (Figure 1.10) [30].
(2) Fixed-frequency parallel resonant converter (PRC) (Figure 1.11) [31].
12
(3) Fixed-frequency series-parallel or LCC-type resonant converter (SPRC) (Figure 1.12)
[32].
(4) Fixed-frequency LCL series resonant converter (SRC) with a capacitive output filter
(Figure 1.13) [33], [34].
(5) Fixed-frequency LCL SRC with an inductive output filter (Figure 1.14) [35].
(6) Fixed-frequency phase-shifted ZVS PWM full-bridge converter with inductive output
filter (Figure 1.15) [36]-[38].
(7) Fixed-frequency phase-shifted ZVS PWM full-bridge converter with capacitive
output filter (Figure 1.16) [39]-[42].
1.6.1
Fixed-Frequency Series Resonant Converter (SRC)
Figure 1.10
Clamped Mode Series Resonant converter circuit [30]
A fixed-frequency clamped-mode series resonant converter (SRC) (Figure 1.10) is proposed
in [30]. This converter configuration can operate in the following switching conditions,
depending on the line and load condition: four switches operate with ZVS turn-on; four
switches operated with ZCS turn-off; two switches in one leg operate with zero-current turnoff and the other two switches operated with zero-voltage turn-on. The major problems with
this converter is that it offers a very narrow range of ZVS for varying line and load condition
in the present application, and with ZCS of the full-bridge switches, there is always an issue
of shoot-through due to slow reverse recovery of the MOSFETs anti parallel diodes.
13
1.6.2
Fixed-Frequency Parallel Resonant Converter (PRC)
Figure 1.11
Clamped-Mode Parallel Resonant converter [31]
A fixed-frequency clamped-mode parallel resonant converter (PRC) (Figure 1.11) is
proposed in [31]. The proposed converter offers ZVS from full load to no load, but the
inverter peak current does not decrease much with reduction in the load current, and there is
no dc blocking coupling capacitor in series to prevent saturation of the HF transformer.
1.6.3
Fixed-Frequency Series-Parallel or LCC Resonant Converter (SPRC)
Figure 1.12
Fixed-Frequency Series-Parallel Resonant converter
A fixed-frequency, series-parallel resonant converter (SPRC) (Figure 1.12) is proposed in
[32]. This converter also cannot maintain ZVS for all the primary switches for wide variation
in line and load condition in the present application.
1.6.4
Fixed-Frequency LCL Series Resonant Converter (SRC) with a Capacitive
Output Filter
Another fixed-frequency, LCL modified series resonant converter with capacitive output
filter (Figure 1.13) is described in [33],[34]. This converter offers ZVS for all the switches
14
for a wider change in load current variation. But one of the major issues with this converter is
the very high peak resonant tank current at lower output voltage and maximum high output
current.
Figure 1.13
1.6.5
Fixed Frequency LCL SRC with capacitive output filter [33],[34]
Fixed-Frequency LCL SRC with an Inductive Output Filter
Figure 1.14
Fixed Frequency LCL SRC with inductive output filter [35]
Fixed-frequency LCL, modified series resonant converter with inductive output filter (Figure
1.14) is discussed in [35]. This converter also offers ZVS for a wide change in load and
supply voltage variation. Moreover, the resonant current is clamped approximately to the
reflected load current. This converter, on the other hand, suffers from severe voltage
overshoot and ringing due to the interaction of the transformer leakage inductance with the
junction capacitance of the rectifier diode and loss of duty cycle on the secondary side of the
transformer.
15
1.6.6
Inductive Output Filter
Figure 1.15
Full-bridge Phase-shifted converter with inductive output filter [36]-[38]
All of the above-mentioned resonant converters suffer from high resonant peak stresses on
the circuit components and require components with higher current or voltage ratings. The
full-bridge phase-shifted converter with inductive output filter topology (Figure 1.15) [36][38] provides a much easier solution to the switching loss problem. Its control features are
similar to regular PWM converters and it uses parasitic elements (transformer leakage
inductance) to control the switching transition for ZVS. Also, the resonant peaks are absent
thus limiting the stresses on the converter components. This converter, on the other hand,
suffers from severe voltage overshoot and ringing due to the interaction of the transformer
leakage inductance with the junction capacitance of the rectifier diode, loss of duty-cycle on
the secondary side of the transformer and looses ZVS for wide variation in line and load
conditions [36],[38]. Another issue with phase-shifted gating scheme is that it is difficult to
achieve a 0% duty-cycle at lighter and no load due to delay mismatch in the duty-cycle
generation and gate-drive circuits. The rectifier’s ringing can be damped by using a clamp
circuit or by using clamp diodes and commutating inductor in the primary circuit [43]-[45].
The modified full-bridge, phase-shifted converter [43]-[45] reduces the switching losses in
rectifier diodes and offers ZVS over a wide range of line and load variation, provided the
16
transformer leakage inductance is very small and the required inductance for achieving ZVS
is realized by using an extra commutating inductor plus two clamp diodes and by increasing
the magnetizing current of the high frequency transformer. Thus the major issues of this
configuration are the increased circulating current in the primary MOSFETs along with
reverse recovery losses in the clamp diodes.
In [46]-[48] various configurations with additional passive and active auxiliary circuits to
overcome the basic issues in the phase-shifted converter mentioned above are presented.
None of these converters [46]-[48] solves all of the problems.
A novel, hybrid phase-shifted converter is presented in [49]. This configuration uses two
transformers and achieves ZVS for all the primary switches over the entire line and load
range but still suffers from loss of duty-cycle and high voltage ringing of the output rectifier
diodes.
Although various other solutions have been suggested for this converter [50]-[62], all of them
increase the component count and suffer from one or more disadvantages including limited
ZVS range, high-voltage ringing on the secondary side rectifier diodes, or loss of duty-cycle.
Wide ZVS range of operation is discussed in [51]-[54]. The high-voltage ringing on the
secondary-side rectifier diodes is addressed in [55]-[58]. The loss of duty-cycle is reviewed
in [59],[60]. In order to reduce the RMS current in the primary, a new leading-edge PWM
control scheme is presented in [61]. Various methods to increase light-load efficiency are
discussed in [62],[63].
A new complementary gating scheme for the full-bridge dc-dc PWM converter is presented
in [64]. This gating scheme requires an additional ZVT circuit to achieve ZVS for all the
switches for a wide variation in the load current.
17
A new full-bridge ZVS converter operating with trailing edge PWM gating scheme was
presented in [65]. This converter behaves like a traditional hard-switched topology, but rather
than driving the diagonal full-bridge switches simultaneously with PWM, the low-side, fullbridge switches are driven at a fixed 50 % duty cycle and the upper switches are pulse-width
modulated on the trailing edge as shown in Figure 1.16. As a result of this gating scheme, all
the switches operate with ZVS for a very wide range of load condition. This converter also
suffers with similar issues like the phase-shifted converter. The trailing-edge PWM gating
scheme can achieve 0% duty-cycle at lighter and no load without any issue since zero dutycycle can be realized by completely turning-off the upper switches.
Trailing-edge PWM
controlled (Q1 and Q2)
Vg1
Vg2
Vg3
Vg4
Fixed 50% duty cycle
Figure 1.16
Time (µs)
Trailing-edge PWM gating scheme
18
1.6.7
Capacitive Output Filter
Figure 1.17
Full-bridge Phase-shifted converter with capacitive output filter [39]-[42]
The full-bridge phase-shifted converter with capacitive output filter (Figure 1.17) inherently
minimizes diode rectifier ringing since the transformer leakage inductance is effectively
placed in series with the external resonant inductor [39]-[42]. The converter can be operated
in continuous conduction mode (CCM), boundary conduction mode (BCM), and
discontinuous conduction mode (DCM). When this converter is operated in BCM and DCM
all the secondary diodes turn-on and off with ZCS. Due to ZCS turn-on and turn-off of the
secondary diodes, there is no reverse recovery loss and the voltage across the diodes is
naturally clamped to the output voltage. Another advantage of this configuration is two
primary MOSFETs that turn-on with ZVS and the other two MOSFETs turn-on with zero
current over a wide range of load current. The major disadvantage is the turn-off current in
the primary switches is significantly high which causes high conduction and turn-off
switching losses. Another issue with phase-shift control gating scheme is that it is difficult to
achieve a 0% duty-cycle at lighter loads due to mismatch of delays in the duty-cycle
generation and gate-drive circuits.
A complementary gating scheme for the full-bridge dc-to-dc PWM converter with capacitive
output filter is presented in [66]. This gating scheme requires an additional ZVT circuit to
19
achieve ZVS for all the switches for a wide variation in the load current. The major
disadvantage of this converter is the asymmetry in the resonant inductor peak current which
makes it difficult to implement peak current mode control.
1.7
Thesis Motivation
As discussed in the previous section, the full-bridge dc-dc converter operating with trailingedge PWM gating is not completely explored in the literature. Thus in this thesis, the fullbridge dc-dc converter operating with trailing-edge PWM gating will be investigated in
detail. Both versions of the converter, with inductive and capacitive output filters are studied
in detail, as well.
The full-bridge ZVS converter with inductive output filter operating with trailing-edge PWM
gating scheme was presented in [65]. There is no detailed analysis and step-by-step design
procedure available in literature for this application. Therefore, a detailed mode analysis is
performed on this converter for the present application, and the results are presented in [67][69] along with detailed design procedure and experimental results for a 3.3 kW PHEV
battery charger.
The trailing-edge PWM gating scheme could be also applied to the full-bridge ZVS converter
with capacitive output filter. There is no detailed analysis and step-by-step design procedure
available in literature for this configuration. Therefore a detailed mode analysis is performed
on this converter for the present application and the results are presented in [70] and [71]
along with detailed design procedure and experimental results for a 1.65 kW PHEV battery
charger.
As presented in section 1.5, an interleaved, multi-cell configuration approach, for 2 kW and
higher power levels offers various advantages such as: each cell shares equal power, the
20
thermal losses are distributed uniformly among the cells, and the input/output ripple is four
times the switching frequency which reduces the filter size and cost. A 3.3 kW interleaved
(2-cell) full-bridge DC-DC converter with trailing-edge PWM gating scheme is analyzed and
designed, and a lab prototype was built and tested for the present application [72].
A 3.3 kW interleaved (2-cell) full-bridge DC-DC converter with capacitive filter and voltagedoubler rectifier operating with trailing-edge PWM gating scheme is also an attractive
solution for the present application. The output voltage-doubler rectifier reduces half the
number of secondary diodes (resulting in lower cost and overall converter size) as compared
to configuration proposed in [72]. There is no detailed analysis and step-by-step design
procedure available in literature for this configuration. A 3.3 kW interleaved (2-cell) fullbridge DC-DC converter with voltage-doubler rectifier with trailing-edge PWM gating
scheme is analyzed and designed, and a lab prototype was built and tested for the present
application [73].
1.8
Thesis Outline
The layout of the thesis is as follows:
Chapter 2- In this chapter the ZVS full-bridge dc-dc converter with inductive output filter
will be studied along with a detailed operating principle, steady-state analysis, design
consideration, and experimental results.
Chapter 3- In order to overcome some of the issues like rectifier diode ringing issue, loss of
duty cycle, and primary-side circulating current present in full-bridge dc-dc converter with
inductive output filter, a ZVS full-bridge dc-dc converter with capacitive output filter is
proposed. In this chapter the full-bridge dc-dc converter with capacitive output filter will be
21
studied along with a detailed operating principle, steady-state analysis, step-by-step design
procedure, simulation, and experimental results.
Chapter 4- In this chapter, in order to overcome issues with thermal management for high
power applications, an interleaved ZVS full-bridge dc-dc converter with capacitive output
filter will be presented along with a detailed operating principle, steady-state analysis, design
consideration, simulation, and experimental results.
Chapter 5- In order to minimize the number of components, reduce cost and power density
of the converter described in chapter 4, an interleaved ZVS full-bridge dc-dc converter with
capacitive output filter and voltage-doubler rectifier will be presented along with a detailed
operating principle, steady-state analysis, design consideration, simulation, and experimental
results.
Chapter 6- This chapter summarizes the contributions of the thesis and scope of future work.
22
Chapter 2: Full-Bridge DC-DC Converter with Inductive Output Filter
Operated with Trailing-Edge PWM Gating2
2.1
Introduction
This chapter presents a full-bridge dc-dc converter with inductive output filter operating with
trailing-edge PWM gating for use in the dc-dc converter stage of a PHEV onboard battery
charger.
As discussed in chapter 1, soft-switching techniques (i.e. ZVS and ZCS) can be used in dc-dc
converters to reduce switching losses without reducing the switching frequency. Operation at
high switching frequency aids in mainly reducing the size and weight. Additionally the use of
soft-switching increases the conversion efficiency of the converter. All the primary side
switches of the trailing-edge, PWM full-bridge converter turn-on with ZVS for a wide range
of load conditions. Another major advantage of this converter over a phase-shifted converter
is that it can achieve 0% duty-cycle at light and no-load conditions without any extra
circuitry since zero duty-cycle can be realized by completely turning-off the PWM controlled
upper switches. All the above discussed merits make it suitable for the trailing-edge, PWM
full-bridge converter to be used as the dc-dc converter stage of an onboard battery charger.
Therefore, this chapter presents a full-bridge dc-dc converter with inductive output filter
operating with trailing-edge PWM. Although this topology has been reported in [64], its
detailed operation, design and experimental results for an on-board battery charger
2
Content from this chapter has been published in [D.S. Gautam, Fariborz Musavi, Murray Edington, W. Eberle
and W.G. Dunford, "An Automotive On-Board 3.3 kW Battery Charger for PHEV application," Proceedings of
IEEE Vehicular Power and Propulsion Conference (VPPC 2011), Chicago, pp. 1-6, Sep. 2011], [D.S. Gautam,
Fariborz Musavi, Murray Edington, W. Eberle and W.G. Dunford, "An Automotive On-Board 3.3 kW Battery
Charger for PHEV application," IEEE Transactions on Vehicular Technology, vol. 61, no. 8, pp. 3466-3474,
Oct. 2012] and [D.S. Gautam, Fariborz Musavi, Murray Edington, W. Eberle and W.G. Dunford, "A Zero
Voltage Switching Full-Bridge DC-DC Converter for an On-Board PHEV Battery Charger," Proceedings of
IEEE Transportation Electrification Conference and Expo (ITEC 2012), Dearborn, pp. 1-6, Jun. 2012].
23
application are not available in the literature. The content of this chapter includes the
following. Section 2.2 explains the detailed operating principle. Section 2.3 gives the design
procedure for selecting various components and devices based on the analysis presented in
section 2.2. Based on this design method, a 3.3 kW, 200 kHz, dc-to-dc converter is designed
and built in the laboratory, and the experimental results are presented in Sections 2.4. Finally,
performance evaluation of this converter with various semiconductor combinations is
presented in Section 2.5.
2.2
Operating Principle
The circuit diagram of the full-bridge dc-dc converter with inductive output filter operating
with trailing-edge PWM gating scheme is shown in Figure 2.1.
Lo
Vin
Q1
CQ1
Vg1
Co1
Q3
Vg3
Figure 2.1
b
Q2
a
Vab
CQ3
CQ2
Vg2
Q4
Vg4
iLr
Lr
DR1 DR2
CQ4
Io
Dc
HF
Transformer
Isec
VRec_in VRectout
nt:1
DR3 DR4
Vo
iLo
Co2
Rc
HV
Battery
Cc
Trailing-edge PWM Full-bridge dc-dc converter with inductive output filter
MOSFETs Q1 – Q4 are the primary-side switches of the full-bridge, and as shown in the
circuit diagram, all the MOSFETs are also modeled with parasitic drain-to-source antiparallel
diodes, and capacitors CQ1 – CQ4. DR1 – DR4 are the secondary-side rectifier diodes. Primaryside resonant inductor Lr is a combination of the leakage inductance of the transformer
reflected to the primary side, and any external inductor connected in series with the
transformer. Lo, is the output filter inductor, and its value is very large as compared to that of
Lr. Co1, which is the input bulk filter capacitor and is usually also part of the output filter
24
capacitor of the preceding front-end PFC stage (not shown in Figure 2.1). Co2 is the output
filter capacitor and is very small in value as compared to Co1. Finally Rc, Cc and Dc, form the
RCD voltage clamp circuit to clamp the high voltage ringing across the diodes DR1 – DR4,
which is caused by interaction of the transformer leakage inductance with the diode parasitic
capacitance [36], [74].
Trailing-edge PWM
Vg1
Vg2
Vg3
Vg4
Fixed 50% duty cycle
Figure 2.2
Time (µs)
Trailing-edge PWM gating scheme
Fixed-frequency, trailing edge PWM gating control is achieved by driving the lower switches
(Q3 and Q4) at a fixed 50% duty cycle and the upper switches (Q1 and Q2) are pulse-width
modulated on the trailing edge as shown in Figure 2.2, which creates a potential difference,
Vab, across transformer primary winding. Due to this a voltage is induced in the secondary
winding of the transformer, which is then rectified by diodes DR1 – DR2, and finally, filtered
by Lo and Co2. Inductor Lr resonates with the parasitic capacitance CQ1 – CQ4 in order to
facilitate ZVS turn-on for MOSFETs. The operating waveforms are shown in Figure 2.3 for
an arbitrary pulse width ‘δ’.
To simplify the presentation of the operating principle, all components are assumed to be
ideal; input and output filter capacitor Co1 and Co2 is considered equivalent to a constant
25
voltage source (ripple free). All parasitic capacitances in the circuit, including winding and
heatsink capacitance, have been lumped together with the switch capacitances CQ1 – CQ4.
Diodes DR1 – DR4 are assumed to be ideal, hence the effect of high-voltage ringing, due to
resonance of the diode junction capacitance with the transformer leakage inductance, is not
discussed here. Also the magnetizing inductance of the transformer is considered to be large,
hence the effect of magnetizing current is neglected in the analysis. Figure 2.3 illustrates the
detailed operating waveforms of the trailing-edge PWM full-bridge converter with intervals
and devices conducting during each interval. The gating signals, Vg1 – Vg2, for all the
primary, switches Q1 – Q4, resonant inductor current iLr, the bridge voltage Vab, the output
bridge rectifiers voltage VRect_in and output inductor current iLo are also shown.
The state in which two diagonally opposite primary switches are conducting is called the
active state, and the state in which the primary current freewheels when two switches on the
same side of the power bus are conducting is called the passive state. The two sets of
switches (Q1, Q2 and Q3, Q4) operate under different conditions. As can be observed from
Figure 2.3, the converter moves from the active to the passive state whenever Q1 and Q2 turnoff.
26
Vg1
δ
Vg2
Vg3
Vg4
Vin
I’1=I1/nt
iLr
Vab
I5
I4
I’3=-I3/nt
I’1=-I1/nt
I’2=-I2/nt
Vin/nt
Passive State
Active State
Active Sate
Passive to
Active State
Active to
Passive State
VRec_in
Duty Cycle Loss
Resonant
Delay
I2
iLo
I3
Io
I1
Available Duty Cycle on
Secondary Side
I1
iQ2
iQ2
iDQ2+iCQ2
iQ3
iQ3
iDQ3+iCQ3
1
Intervals
2
3
11
45 6
7
8
9
10
12
DR2,DR3,Q1,Q4
Devices
Time
T0
Figure 2.3
DR2,DR3,CQ1,CQ3,Q4
DR1,DR4,Q2,Q3
DR2,DR3,Q4,DQ3
DR1~DR4,DQ2,DQ3
DR1~DR4,CQ2,CQ4,Q3
DR1~DR4,DQ2,DQ3
T1 T2
T3 T4 T5 T6
T7
t
Typical operating waveforms for an arbitrary pulse width ‘δ’ to illustrate the operation of
the trailing-edge PWM full-bridge converter
27
2.2.1
Interval 1 (T0 – T1)
Lo
Vin
Q1
CQ1
Vg1
Co1
Q3
b
Vg3
Figure 2.4
Q2
a
Vab
CQ3
CQ2
Vg2
iLr
Lr
DR1 DR2
Io
Dc
HF
Transformer
Co2
VRec_in
Q4
Vg4
CQ4
nt:1
DR3 DR4
Vo
iLo
Rc
HV
Battery
Cc
Equivalent circuit for Interval 1 (T0 – T1)
During this interval, switches Q1 and Q4 are on and Q2 and Q3 are off. Voltage across node
‘a’ and ‘b’ is vab=-Vin. On the secondary side, the rectifier diodes DR2 and DR3 are
conducting. This is a power-transfer mode (active power state) and the primary current flows
through Q1, transformer primary winding, resonant inductor Lr and finally, through Q4, as
illustrated in Figure 2.4.
The initial current in the output inductor iLo(0)=I1. The output inductor current iLo(t) using
initial condition iLo(0)=I1 is given by:
𝑉
( 𝑛𝑖𝑛 ) − 𝑉𝑜
𝑖𝐿𝑜 (𝑡) = 𝐼1 + [ 𝑡
] (𝑡 − 𝑇0 )
𝐿𝑜
2-1
The resonant inductor current iLr(t) is given by:
𝑖𝐿𝑟 (𝑡) = −
𝑖𝐿𝑜 (𝑡)
𝑛𝑡
2-2
𝑰
At the end of this interval iLr reaches the peak resonant inductor current, 𝑰′𝟐 = 𝒏𝟐 where I2 is
𝒕
the peak output inductor current as shown in Figure 2.3.
28
2.2.2
Lo
Vin
Q1
CQ1
Vg1
Co1
Q3
Vg3
Figure 2.5
b
Q2
a
Vab
CQ3
CQ2
Vg2
iLr
Lr
DR1 DR2
Io
Dc
HF
Transformer
Co2
VRec_in
Q4
Vg4
CQ4
nt:1
DR3 DR4
Vo
iLo
Rc
HV
Battery
Cc
Interval 1 terminates when switch Q1 turns off as determined by the PWM duty cycle ‘δ’.
This is a transition mode from active state to passive state. Since the current flowing through
the primary cannot be interrupted instantaneously, it finds an alternate path and flows
through the parasitic switch capacitances of CQ1 and CQ3, which discharges the node ‘b’ to 0
V by charging CQ1 and discharging CQ3, as shown in Figure 2.5.
During this interval, the resonant inductor current is assumed to be constant iLr = I’2. The
voltage across CQ1 and CQ3 is given by:
𝑣𝐶𝑄1 (𝑡) =
𝐼′2 (𝑡 − 𝑇1 )
(𝐶𝑄1 + 𝐶𝑄3 )
𝑣𝐶𝑄3 (𝑡) = 𝑉𝑖𝑛 − 𝑣𝐶𝑄1
2-3
2-4
During this switch transition, the energy stored in the output inductor (Lo) and the resonant
inductor (Lr) assist in charging and discharging the capacitances CQ1 and CQ3 respectively.
29
2.2.3
Lo
Vin
Q1
CQ1
Vg1
Co1
Q3
Vg3
Figure 2.6
b
Q2
a
Vab
CQ3
CQ2
Vg2
iLr
Lr
DR1 DR2
Io
Dc
HF
Transformer
Co2
VRec_in
Q4
Vg4
CQ4
nt:1
DR3 DR4
Vo
iLo
Rc
HV
Battery
Cc
At the end of interval 2, CQ3 has completely discharged vCQ3 = 0. During this interval, the
primary current freewheels (passive state) through the body diode of Q3, transformer primary
winding, resonant inductor Lr, and finally, through Q4, as illustrated in Figure 2.6. Since the
body diode of Q3 is conducting, switch Q3 is ready to be turned on under the ZVS condition.
In this topology, switches Q3 and Q4 (50% duty-cycle controlled switches) always achieve
ZVS with the help of the energy stored in the output inductor (Lo) for nearly the entire load
current (Io) range. On the secondary side, the rectifier diodes DR2 and DR3 are still
conducting.
The output inductor current iLo(t) is given by:
𝑖𝐿𝑜 (𝑡) = 𝐼2 − [
𝑉𝑜
] (𝑡 − 𝑇2 )
𝐿𝑜
2-5
The resonant inductor current iLr(t) is given by:
𝑖𝐿𝑜 (𝑡)
2-6
𝑛𝑡
At the end of this interval Q3 and Q4 toggle and 𝑖𝐿𝑜 reaches I3, as shown in Figure 2.3.
𝑖𝐿𝑟 (𝑡) = −
30
2.2.4
Lo
Vin
Q1
CQ1
Vg1
Co1
b
Q3
Figure 2.7
CQ2
Vg2
a
Vab
CQ3
Vg3
Q2
Lr
iLr
DR1 DR2
Io
Dc
HF
Transformer
Q4
nt:1
CQ4
DR3 DR4
HV
Battery
Co2
VRec_in
Vg4
Vo
iLo
Rc
Cc
During this interval, after gating signal Vg3 is applied, switch Q3 turns on with ZVS. After Q4
turns off, the current flowing in the resonant inductor resonates with parasitic switch
capacitances of CQ2 and CQ4, which charges node ‘a’ to Vin by charging CQ4 and discharging
CQ2, as shown in Figure 2.3 and 2.7. During this interval, all secondary diodes DR1 ~ DR4 are
free-wheeling and this commences the duty-cycle loss period, as shown in Figure 2.3 and 2.7.
The voltage across CQ2 and CQ4 is given by equation 2-7 and 2-8, where Z=√𝐶
𝐿𝑟
𝑄2 +𝐶𝑄4
and 𝜔𝑟 =
1
.
√𝐿𝑟 (𝐶𝑄2 +𝐶𝑄4 )
𝑣𝐶𝑄2 (𝑡) = 𝑉𝑖𝑛 − 𝑣𝐶𝑄4 (𝑡)
𝑣𝐶𝑄4 (𝑡) =
2-7
𝐼3
𝑍 sin{𝜔𝑟 (𝑡 − 𝑡3 )}
𝑛𝑡
The resonant inductor current iLr(t) using initial condition 𝑖𝐿𝑟 (0) = −
𝑖𝐿𝑟 (𝑡) = −
𝐼3
cos{𝜔𝑟 (𝑡 − 𝑡3 )}
𝑛𝑡
2-8
𝐼3
𝑛𝑡
is given by:
2-9
At the end of this interval, node ‘a’ reaches Vin and 𝑖𝐿𝑟 reaches I4 as shown in Figure 2.3.
31
Since ZVS transition of Q1 and Q2 is dependent on the output load current (Io) and energy
stored in the resonant inductor LR, the minimum load current required to achieve ZVS turnon for Q1 and Q2 is given by:
𝐼𝑜𝑚𝑖𝑛 =
2.2.5
𝑉𝑖𝑛 𝑛𝑡
𝑍
2-10
Lo
Vin
Q1
CQ1
Vg1
Co1
Q3
Vg3
Figure 2.8
Q2
b
a
Vab
CQ3
CQ2
Vg2
iLr
Lr
DR1 DR2
Io
Dc
HF
Transformer
Co2
VRec_in
Q4
Vg4
CQ4
nt:1
DR3 DR4
Vo
iLo
Rc
HV
Battery
Cc
During this interval, after CQ2 is fully discharged, the body diode of Q2 conducts. The
primary current freewheels (passive state) through the body diode of Q3, transformer primary
winding, resonant inductor Lr and finally through the body diode of Q2, as illustrated in
Figure 2.8. All secondary diodes DR1 ~ DR4 are still free-wheeling, and passive state
continues.
The resonant inductor current iLr(t) using initial condition 𝑖𝐿𝑟 (0) = 𝐼4 is given by:
𝑉𝑖𝑛
(𝑡 − 𝑡4 )
𝐿𝑟
reaches I5, as shown in Figure 2.3.
𝑖𝐿𝑟 (𝑡) = 𝐼4 +
At the end of this interval 𝑖𝐿𝑟
2-11
The period from T3 to T5 is the resonant delay time as shown in Figure 2.3. This is the delay
time after Q3 and Q4 toggles and before Q2 is turned on. In order to ensure that Q2 turns with
ZVS, the required resonant delay should be at least 1/4 of the period of the resonant
32
frequency of the circuit formed by the resonant inductor of the Lr and the parasitic
capacitances CQ2 and CQ4. This resonant transition may be estimated by:
𝜏=
2.2.6
𝜋
√𝐿 + (𝐶𝑄2 + 𝐶𝑄4 )
2 𝑟
2-12
Lo
Vin
Q1
CQ1
Vg1
Co1
Q3
Vg3
Figure 2.9
b
Q2
a
Vab
CQ3
CQ2
Vg2
iLr
Lr
DR1 DR2
Io
Dc
HF
Transformer
Co2
VRec_in
Q4
Vg4
CQ4
nt:1
DR3 DR4
Vo
iLo
Rc
HV
Battery
Cc
During this interval, gating signal is applied to switch Q2, and it turns on with ZVS, as shown
in Figure 2.9, and polarity of the primary current changes. In addition, all secondary diodes
DR1 ~ DR4 are still free-wheeling, and passive state continues.
The resonant inductor current iLr(t) using initial condition 𝑖𝐿𝑟 (0) = 𝐼5 is given by:
𝑖𝐿𝑟 (𝑡) = 𝐼5 +
𝑉𝑖𝑛
(𝑡 − 𝑡5 )
𝐿𝑟
2-13
At the end of this interval, the current through the primary of the transformer and resonant
inductor Lr reaches I’1 and equals the current through the output inductor Lo, as shown in
Figure 2.3. The secondary rectifier diodes DR2 and DR3 completely turns off, and the entire
load current flows through DR1 and DR4 only, thus ending the duty-cycle loss period, and the
power transfer mode commences again.
Operation of the intervals 7 to 12 can be explained in a similar way to intervals 1 to 6.
33
2.3
Design Procedure
This section provides the details for designing and selection of various components of the
trailing-edge PWM full-bridge converter as discussed in the previous section. Based on the
design procedure, a 3.3 kW dc-dc converter stage is designed to meet the specification of a
level-2 charger as discussed in Table 1.2. The detailed specifications for designing the dc-dc
converter are given in Table 2.1.
Table 2.1 Design specification of the Trailing-edge PWM Full-bridge dc-dc converter
2.3.1
Parameters
Value[Units]
Input DC Voltage (from PFC stage)
380 to 420 [V]
Output DC Voltage Range
200 to 450 [V]
Maximum Output DC Current
Maximum Output Power (at 300V
output voltage)
Output Voltage Ripple
11 [A]
< 2 [Vp-p]
Efficiency
Up to 96 [%]
3.3 [kW]
Selection of Switching Frequency (fs)
An experimental efficiency comparison for the trailing-edge PWM full-bridge converter is
provided in Figure 2.10 at half-load power, (i.e. 1.65 kW) for switching frequencies between
66 kHz and 250 kHz. At 66 kHz, the converter has maximum overall efficiency. Since the
converter components, including the resonant inductor, transformer, and output inductor
were optimized for 66 kHz operation, the efficiency is lower at 150, 200 and 250 kHz.
However, since the difference in losses at full load is limited to 2.3 %, a 200 kHz switching
frequency was selected for the dc-to-dc stage. And finally, the magnetic components were
redesigned for the final selected switching frequency of 200 kHz.
34
Figure 2.10
Comparison of measured efficiency as a function of output power for different switching
frequencies at Vo = 300V and Po = 1.65 kW
2.3.2
Selection of Transformer Turns Ratio (nt)
The transformer turns ratio nt is calculated using equation 2-14, where DCloss includes deadtime and duty-cycle loss.
𝑛𝑡 =
𝑉𝑖𝑛 (1 − 𝐷𝐶𝐿𝑜𝑠𝑠 )
𝑉𝑜
2-14
The transformer turns ratio is determined to be 0.75 for an input and output voltage of 400 V,
and DCLoss is assumed to be 0.25. An EE55 shape ferrite core (using material R from Mag
Inc.) transformer was designed using turns ratio of 12 (number of primary turns):16 (number
of secondary turns). Two 18 AWG (65 strands of 36 AWG wire) twisted Litz wires were
used for the primary and secondary windings.
2.3.3
Selection of Output Filter Inductor (Lo)
The output filter inductor Lo is calculated to be 400 µH using equation 2-15, where ΔIo is the
peak-to-peak output inductor ripple current is assumed to be 1 A (10 % of maximum output
current).
35
𝐿𝑜 =
𝑉
( 𝑛𝑖𝑛 − 𝑉𝑜 )(1 − 𝐷𝐿𝑜𝑠𝑠 )
𝑡
2-15
∆𝐼𝑜 2𝑓𝑠
A 400 µH inductor was designed using a 125 µ permeability toroidal core (part number:
44738 from Mag Inc.) and by winding 38 turns of 15 AWG copper wire.
2.3.4
Selection of Resonant Inductor (Lr)
The resonant inductor Lr is calculated to be 8 µH using equation 2-16.
𝐿𝑟 =
𝑛𝑡 𝑉𝑖𝑛 (𝐷𝐿𝑜𝑠𝑠 )
4 ∆𝐼𝑜 𝑓𝑠
2-16
A 6µH resonant inductor was selected, which is smaller as compared to the value calculated
using equation 2-16. By using a 6 µH inductor, ZVS can be achieved for Q1 and Q2 from load
current of Io = 11 A down to 5.5 A. Below 5.5 A, Q1 and Q2 will have turn-on switching
losses, but the total losses at 5.5 A are 16 W, which are considerably lower than the 20 W of
total loss with ZVS at 11 A load. The heatsink around the primary MOSFETs is designed to
extract 20W from each primary device, which is sufficient to handle the light-load losses
when the MOSFETs lose ZVS. Finally, a lower resonant inductor value reduces the dutycycle loss and helps achieve higher full-load efficiency by minimizing the circulating current
conduction loss. A 6 µH inductor was realized by connecting two 2 µH each external
inductors in series (total 4 µH) and an additional 2 µH was obtained using the transformer
leakage inductance. Each 2 µH external inductor was designed using a 14 µ permeability
toroidal core (part number: 55123A2 from Mag Inc.) and by winding 13 turns of 15 AWG
Type 2 Litz wire (3x43 strands of 36 AWG wire).
36
2.3.5
Selection of MOSFETs (Q1 – Q4)
Using the analysis presented in section 2.2 (equation 2-1 to 2-13), the RMS current through
switches Q1, Q2 and Q3, Q4 is given by equation 2-17 and 2-18 respectively:
1 𝑇7
𝐼𝑄12(𝑟𝑚𝑠) = √ ∫ 𝑖𝐿𝑅 (𝑡)2 𝑑𝑡
𝑇 𝑇3
2-17
1 𝑇9
= √ ∫ 𝑖𝐿𝑅 (𝑡)2 𝑑𝑡
𝑇 𝑇1
2-18
𝐼𝑄34(𝑟𝑚𝑠)
RMS current through the primary switches was calculated to be 8 A and 10 A using equation
2-17 and 2-18 for full-load condition (Vin = 400 V, Vo = 300 V and Io = 11 A).
A 600V, 83 mΩ Rdson (switch ON state resistance), 46 A, 450 pF Cds (parasitic capacitance)
MOSFET (part number: SPW47N60CFD from Infineon) with a fast body diode was selected
for the four primary switches.
2.3.6
Selection of Rectifier Diodes (DR1 – DR4)
The average current through the output rectifier diodes DR1 to DR4, IDR(ave) is given by:
𝐼𝐷𝑅(𝑎𝑣𝑒) =
𝐼𝑜
2
2-19
Average current through the rectifier diodes was calculated to be 5.5 A using equation 2-19
for Io = 11 A.
A 600 V, 12 A silicon carbide (SiC) schottky diode (part number: IDH12S60C from Infineon)
was selected for the four rectifier diodes.
37
2.3.7
Selection of Trailing-edge PWM Controller and MOSFET Gate Driver
For implementing the trailing edge PWM gating scheme, ISL6753 PWM controller (from
Intersil) was used and for driving the MOSFETs Q1 – Q4, IR2110 gate driver (from
International Rectifier) was selected.
2.3.8
Selection of Output filter capacitor (Co2)
The RMS current through the output filter capacitor Co2 is given by equation 2-20 and its
capacitance value is determined using equation 2-21.
𝑇7
1
𝐼𝐶𝑜2(𝑟𝑚𝑠) = √
∫ (𝑖𝐿𝑜 (𝑡) − 𝐼𝑜 )2 𝑑𝑡 = 0.22 [𝐴]
𝑇7 − 𝑇1 𝑇1
𝐶02 =
𝐼𝐶𝑜2(𝑟𝑚𝑠)
= 0.2 [𝜇𝐹]
4𝜋𝑓𝑠 𝑉𝑟𝑖𝑝𝑝𝑙𝑒
2-20
2-21
A 33 µF, 500V electrolytic capacitor (part number ECST501ELL330MLN from NipponChemi-Con) was selected for the output filter capacitor.
38
2.4
Experimental Results
Based on the design presented in section 2.3, a 3.3 kW laboratory prototype was built to the
design specifications of Table 2.1, as shown in Figure 2.11.
Figure 2.11
Prototype unit of trailing-edge PWM full-bridge converter with inductive output filter
Experimentally measured efficiency curves at Vo = 200, 300, 400 and 450 V output over the
entire power range with Vin = 400 V are provided in Figure 2.12.
Figure 2.12 Measured efficiency versus output power at different output voltages with Vin = 400 V
39
It should be noted that the converter achieves a peak efficiency of 96 % at Vo = 400 V, Io =
8.25 A and maximum output power of 3.3 kW. At maximum output current Io = 11 A, Vo =
300 V and output power of 3.3 kW the converter achieves an efficiency of 94.9 %. It should
be also noted that below half output power 1.65 kW the efficiency reduces drastically. This is
due to switches Q1 and Q2 losing ZVS and loss in the RCD clamp components dominating at
lighter load.
The experimental waveforms of output voltage and current are shown in Figure 2.13 for Vo =
400 V and Io = 8 A. As seen in Figure 2.13, both output voltage and current are nearly free
from low-frequency (120 Hz) ripple. This is one of the important requirements for batterycharging applications to maintain good health of the batteries.
Output Voltage
Ch1 = Vo 100V/div.
Output Current
Ch4 = Io 2A/div.
Figure 2.13
Experimental waveforms of output voltage and current Ch1= Vo 100V/div. Ch4= Io 2A/div.
Figures 2.14, 2.15 and 2.16 demonstrate ZVS turn-on of primary switch Q1, high-voltage
ringing across the rectifier diodes DR1 – DR4 and also loss of duty-cycle. As seen in these
figures, switch Q3 achieves ZVS from 150 W to 3.3 kW loads since the drain-to-source
voltage across Q3, VDSQ3 drops to 0V prior to the gate voltage Vg3 that is applied to turn-on
Q3. Thus as discussed in section 2.2.3, these results show that switch Q3 turns on with ZVS
40
over the entire load range since this transition is assisted by the energy stored in the large
output filter inductor Lo.
Transformer
Q3 Drain to Source
Gate Voltage
secondary current
Voltage
Ch1 = Vg3 5V/div. Ch4 = VDSQ3 100V/div. Ch3 = Isec 2A/div.
Figure 2.14
Rectified Voltage
Ch4 = Vrectout 100V/div.
Experimental waveforms obtained for (Ch1) Q3 gating signal, Vg3 (Ch2) Q3 drain to source
voltage, VDSQ3 (Ch3) Transformer secondary current, Isec (Ch4) Rectifier output voltage, Vrectout at lightload (150 W) with Vin = 400 V and Vo = 300 V
Gate Voltage
Ch1 = Vg3 5V/div.
Q3 Drain to Source
Transformer
Voltage
secondary current
Ch4 = VDSQ3 100V/div. Ch3 = Isec 2A/div.
Rectified
Voltage
Ch4 = Vrectout
100V/div.
Duty cycle
loss
Figure 2.15
Experimental waveforms of Figure 2.14 repeated for half-load (1.65 kW) with Vin = 400 V
and Vo = 300 V
41
Transformer
Q3 Drain to Source
Gate Voltage
secondary current
Voltage
Ch1 = Vg3 5V/div. Ch4 = VDSQ3 100V/div. Ch3 = Isec 5A/div.
Rectified Voltage
Ch4 = Vrectout 100V/div.
Duty cycle
loss
Figure 2.16
Experimental waveforms of Figure 2.14 repeated for full-load (3.3 kW) with Vin = 400 V
and Vo = 300 V
As discussed in section 1.6.6, this converter suffers from excessive high-voltage ringing due
to the interaction of transformer leakage inductance with the junction capacitance of the
rectifier diode. Figures 2.14 to 2.16 show the high-voltage ringing across the output of the
rectifier diodes, Vrectout. This high-voltage ringing is clamped to 532.5 V with the help of a
RCD clamp circuit [74].
The duty-cycle loss period was discussed in section 2.2.4 and 2.2.6. During this period a
significant portion of the primary duty cycle is lost due to the freewheeling of the secondary
diodes, which results in a higher transformer turns ratio, nt as shown in Equation 2.14. As
shown in Figures 2.15 and 2.16, the duty-cycle becomes very significant at half-load and
full-load conditions. As seen in the Figures 2.15 and 2.16 almost 300 ns (13 %) duty-cycle
period is lost at half-load and 1.2 µs (25 %) duty-cycle period is lost at full-load respectively.
Figure 2.17 and 2.18 show the anti-parallel diodes of Q1 and Q3 conduct current (shown in
the grey-shaded area) prior to conducting current through the drain-to-source channel of the
MOSFET. It can be also observed that before the current flows through the channel of the
42
MOSFET, the voltage across the drain-to-source of the MOSFET Q1 and Q3 drops to 0 V,
thus enabling them to turn-on with ZVS.
Q1 Drain to Source
Voltage During Turn ON
Ch1 = VDSQ1 100V/div.
ZVS
Q1 anti-parallel
diode conduction
Q1 Drain to Source
Current During
Turn ON
Ch2 = IDSQ1 5A/div.
Figure 2.17
Experimental waveforms of MOSFET Q1 voltage and current during Turn-ON at Vo = 300
V and Io = 11 A
Q3 Drain to Source
ZVS
Q3 anti-parallel
diode conduction
Q3 Drain to Source
Current During
Turn ON
Ch2 = IDSQ1 5A/div.
Figure 2.18
V and Io = 11 A
Figure 2.19 shows that at light load (300 W) the energy stored in the resonant inductor Lr is
not sufficient to completely charge and discharge CQ3 and CQ1 completely. The anti-parallel
43
diode of Q1 does not conduct current at all. Thus, after the resonant delay time has elapsed
and the gating signal Vg1 is applied, Q1 turns on without ZVS. It was also found that during
this transition a total of 5 W dissipated as turn-on switching loss occurred.
Q1 Drain to Source
Loss of ZVS
Q1 Drain to Source
Current During
Turn ON
Ch2 = IDSQ1 1A/div. CQ1 Parasitic
capacitance
conduction
Figure 2.19
V and Io = 1 A
2.5
Performance Evaluation
Infineon MOSFET and SiC diode
Infineon MOSFET and Hyperfast diode
Fairchild MOSFET and SiC diode
Fairchild MOSFET and Hyperfast diode
95.00
Efficiency (%)
94.50
94.00
93.50
93.00
92.50
Primary MOSFETs and Secondary Diodes
Figure 2.20 Measured Efficiency comparison with different combination of primary MOSFETs and
secondary diodes at Vo = 300 V and Io = 11 A
44
Figure 2.20 illustrates a performance evaluation of the trailing-edge PWM full-bridge dc-dc
converter with inductive output filter for various semiconductor combinations. The various
semiconductor devices used were: Infineon MOSFET (SPW47N60CFD), Fairchild MOSFET
(FCH47N60F), Infineon Silicon Carbide diode (IDH12S60C), and Fairchild hyperfast diode
(ISL9R1560). As shown in Figure 2.20, full-load efficiency was measured to compare their
performances. As shown in the comparison, Infineon’s SPW47N60CFD MOSFET and
IDH12S60C silicon carbide diode performed the best, as it had the highest efficiency of 94.6
%. It should be also noted that the efficiency of Fairchild’s FCH47N60F and Infineon’s SiC
diode combination was almost equally efficient 94.53 %. Also irrespective of the MOSFET
type, the SiC diode outperformed the hyperfast diode. It can be also observed that the
combination of Fairchild’s MOSFET and diode was the least efficient. One of the main
reasons was that the Fairchild MOSFETs were dissipating more power when operated with
the hyperfast diode. Thus, it can be concluded that an SiC diode is an attractive solution for
this topology, since higher efficiency is achieved as compared to the hyperfast diode.
2.6
Conclusions
A new full-bridge dc-dc converter with inductive output filter operating with trailing-edge
PWM gating has been presented in this chapter for the dc-dc stage in PHEV battery charger.
The proposed converter has been presented with detailed analysis, design and experimental
results. It has been shown through experimental results that this converter achieves high
efficiency, the output voltage and currents waveforms are free from 120 Hz AC ripple, and
all the primary MOSFETs achieve ZVS from full-load to half-load condition. It is also shown
that performance with SiC rectifier diode is more superior to hyperfast diode.
45
Some of the drawbacks of the converter such as, duty-cycle loss, high voltage rectifier
ringing and circulating currents in primary side switches were also discussed.
In the next chapter, a full-bridge dc-dc converter with capacitive output filter operating with
trailing-edge PWM is presented, which overcomes all the above mentioned drawbacks such
as, duty-cycle loss, high voltage rectifier ringing and circulating currents in primary-side
switches of the converter presented in this chapter.
46
Chapter 3: Full-Bridge DC-DC Converter with Capacitive Output Filter
Operated with Trailing-Edge PWM Gating3
3.1
Introduction
This chapter presents a full-bridge dc-dc converter with capacitive output filter operating
with trailing-edge PWM gating scheme, as discussed in chapter 1 for use in the dc-dc
converter stage of a PHEV onboard battery charger.
In chapter 2, the trailing-edge PWM full-bridge dc-dc converter with inductive output filter
was presented. As discussed in the last chapter, this converter achieves ZVS for all the
switches with the help of energy stored in the resonant inductor Lr and output filter inductor
Lo. At lighter loads when sufficient energy is not available in Lr, the PWM controlled
switches Q1 and Q2 lose ZVS. It was also shown through experimental results that the 3.3
kW prototype converter achieved 96 % efficiency. As explained in section 2.2, the duty-cycle
loss and high-voltage ringing issues were also demonstrated with experimental results.
To overcome these issues a new topology, the full-bridge dc-dc converter with capacitive
output filter operating with trailing-edge PWM gating is proposed in this chapter. It is noted
that there is no detailed analysis and step-by step-design procedure available in the literature
for this configuration. This proposed converter eliminates all the above-mentioned issues
present in the converter with inductive filter and helps significantly to reduce the size and
cost of the converter.
3
Content from this chapter has been published in: [D.S. Gautam, Fariborz Musavi, Murray Edington, W. Eberle
and W.G. Dunford, "A Zero Voltage Switching Full-bridge DC-DC Converter with Capacitive Output Filter for
a Plug-in-Hybrid Electric Vehicle Battery Charger," Proceedings of IEEE Applied Power Electronics
Conference and Exposition (APEC 2012), Orlando, pp. 1381-1386, Feb. 2012] and [D.S. Gautam, Fariborz
Musavi, Murray Edington, W. Eberle and W.G. Dunford, "A Zero Voltage Switching Full-bridge DC-DC
Converter with Capacitive Output Filter for a Plug-in-Hybrid Electric Vehicle Battery Charger," IEEE
Transactions on Power Electronics, vol. 28, no. 12, pp. 5728-5735, Dec. 2013].
47
The layout of the chapter is as follows. Section 3.2 explains the detailed operating principle.
Section 3.3 gives the design procedure for selecting various components and devices based
on the analysis presented in section 3.2. Based on this design method, a 1.65 kW, 100 kHz,
dc-to-dc converter is designed. PSIM simulation and experimental results are presented in
Sections 3.4. Finally, performance evaluation of this converter with various semiconductor
combinations is presented in Section 3.5.
3.2
Operating Principle
The circuit diagram of the full-bridge dc-dc converter with capacitive output filter operating
with trailing-edge PWM gating scheme is shown in Figure 3.1.
Vin
Vo
Q1
CQ1
Vg1
iLr
Vab
Q3
Vg3
CQ2
Vg2
a
Co1
Figure 3.1
Q2
CQ3
Q4
Vg4
Lr
DR1 DR2
HF
Transformer
Isec
b
CQ4
VRec_in
iRect
iCo2
Co2
Io
HV
Battery
nt:1
DR3 DR4
Trailing-edge PWM Full-bridge dc-dc converter with capacitive output filter
MOSFETs Q1 – Q4 are the primary-side switches of the full-bridge, and as shown in the
circuit diagram, all the MOSFETs are also modeled with parasitic drain to source antiparallel
diodes and capacitors CQ1 – CQ4. DR1 – DR4 are the secondary-side rectifier diodes. The
primary-side resonant inductor Lr is a combination of the leakage inductance of the
transformer reflected to the primary side and any external inductor connected in series with
the transformer. Co1 is the input bulk filter capacitor and is usually also part of the output
filter capacitor of the preceding front-end PFC stage (not shown in Figure 3.1). Co2 is the
output filter capacitor and is very small in value as compared to Co1.
48
As presented in section 2.2 and Figure 2.2, fixed-frequency, trailing edge PWM gating
control is achieved by driving the lower switches (Q3 and Q4) at a fixed 50 % duty cycle and
the upper switches (Q1 and Q2) are pulse-width modulated on the trailing-edge, which creates
a potential difference, Vab across transformer primary winding. Due to this, a voltage is
induced in the secondary winding of the transformer, which is rectified by diodes DR1 – DR2
and finally filtered by Co2. ZVS and ZCS turn-on for the primary MOSFETs are achieved due
to energy stored in inductor Lr during commutation of the MOSFETs Q1 – Q4.
The resonant inductor current (iLr) of the proposed converter can operate in either
discontinuous conduction mode (DCM), boundary conduction mode (BCM), or continuous
conduction mode (CCM). The detailed circuit operation in all three modes is discussed
below:
This converter has six operating intervals for DCM, BCM or CCM. The operating intervals
are determined by the on/off states of the four primary switches. Detailed operating
waveforms are provided for DCM in Figure 3.2, for BCM in Figure 3.3 and for CCM in
Figure 3.4. In the analysis that follows, all components are assumed to be ideal; input and
output filter capacitor Co1 and Co2 is considered equivalent to constant voltage source (ripple
free). All parasitic capacitances in the circuit including winding and heatsink capacitance
have been lumped together with the switch capacitances CQ1 – CQ4. Also the magnetizing
inductance of the transformer is considered to be large and hence the effect of magnetizing
current is neglected in the analysis. The output rectifiers are considered ideal and the external
resonant inductor also includes the transformer leakage inductance.
49
Vg1
Vg2
Vg3
Vg4
+400V
IP1
vab
iLr
-400V
iQ3
iQ2
vo
vDR1 &
vDR4
T
TP
iDR1 &
iDR4
Intervals
Devices
1
3
Q1,Q4, DQ3,Q4,DR1,
DR1,DR4
DR4
T0
Figure 3.2
2
T1
4
5
6
Q2,Q3,DR2, DQ4,Q3,DR2
,DR3
DR3
T2 T3
T4
T5 T6
Time (µs)
Typical operating waveforms to illustrate the operation of the trailing-edge PWM full-bridge
converter in DCM mode
50
Vg1
Vg2
Vg3
Vg4
+400V
IP1
iLR
IP2
vab
iLr
-400V
iQ3
iQ2
T
vo
vDR1 &
vDR4
TP
iDR1 &
iDR4
Intervals
Devices
T0
Figure 3.3
1
2
3
Q1,Q4, DQ3,Q4,
DR1,DR4 DR1,DR4
T1
4
5
Q2,Q3,DR2,DR3
DQ4,Q3,
DR2,DR3
DQ3,Q4,DR1,DR4
T2 T3
T4
6
DQ4,Q3,DR2,
DR3
T5 T6
Time (µs)
converter in BCM mode
51
Vg1
Vg2
Vg3
Vg4
IP1
IP2
iLR
vab
iLr
+400V
-400V
iQ3
iQ2
vDR1 &
vDR4
vo
T
TP
iDR1 &
iDR4
Intervals
1
Devices
Q1,Q4, DR1,DR4
2 3
4
5 6
Q2,Q3, DR2,DR3
DQ1,DQ4, DR2,DR3
DQ3,DQ2, DR1,DR4
DQ3,Q4, DR1,DR4
DQ4,Q3,DR2,DR3
T0
Figure 3.4
T1
T2 T3
T4
T5 T6
Time (µs)
converter in CCM mode
52
3.2.1
Vin
Vo
Q1
Q2
DR1
Lr
DR2
Io
a
Co1
Q3
Figure 3.5
HV
Battery
Co2
b
Q4
DR3
DR4
Equivalent circuit for Interval 1 (T0-T1) for DCM, BCM and CCM
Referring to Figure 3.2 - 3.4, during Interval 1 (T0-T1), switches Q1 and Q4 are on and Q2 and
Q3 are off. This is a power transfer interval, and the primary current flows through Q1,
resonant inductor (Lr), transformer primary, and Q4, as illustrated in Figure 3.5. The rate of
rise of the current (di/dt) through Lr is proportional to the difference between the input
voltage Vin and the output voltage Vo. During this mode power flows to the output through
rectifier diodes DR1 and DR4 and also energy is stored in Lr. The resonant inductor current,
𝑖𝐿𝑟 (𝑡), using initial condition 𝑖𝐿𝑟 (0) = 0 is given by:
𝑉
(𝑉𝑖𝑛 − 𝑛𝑜 )
𝑖𝐿𝑟 (𝑡) =
(𝑡 − 𝑇𝑜 )
𝐿𝑟
3.2.2
3-1
Case (a): Operating in DCM
Vin
Vo
Q1
Q2
Lr
DR1 DR2
Io
a
Co1
Q3
Figure 3.6
Co2
b
Q4
HV
Battery
DR3 DR4
Equivalent circuit for Interval 2 (T1 – T2) for DCM, BCM and CCM and Interval 3 (T2 – T3)
for BCM
53
Referring to Figure 3.2, interval 2 begins after switch Q1 turns off, as determined by the
PWM duty cycle. Since the current flowing in the primary side cannot be interrupted
instantaneously, it finds an alternate path and flows through the parasitic switch capacitances
of Q3 and Q1, which discharges the node ‘a’ to 0V and then forward biases the body diode
D3. During this switch transition, the energy stored in the resonant inductor (Lr) assists in
transferring energy from the lower to upper bridge MOSFET capacitance. Therefore switches
Q3 and Q4 always achieve ZVS with the help of the energy stored in the resonant inductor
(Lr) for nearly the entire load current (Io) range. During this interval the energy stored in Lr is
transferred to the output. The primary resonant inductor (Lr) maintains the current, which
circulates around the path of body diode of Q3, resonant inductor (Lr), transformer primary
and Q4, as illustrated in Figure 3.6. The rate of the downslope of the current through Lr is
proportionate to the output voltage Vo. At T2 the energy stored in Lr is transferred to the
output, the current becomes zero, and the rectifier diodes DR1 and DR4 turn-off. The resonant
inductor current, 𝑖𝐿𝑟 (𝑡), using initial condition 𝑖𝐿𝑟 (0) = 𝐼𝑃1 is given by:
𝑖𝐿𝑟 (𝑡) = 𝐼𝑃1 −
𝑉𝑜
(𝑡 − 𝑇1 )
𝑛𝐿𝑟
3-2
Case (b): Operating in BCM and CCM
Referring to Figures 3.3 and 3.4, the only difference in BCM or CCM as compared to DCM
during Interval 2 is that the current through the resonant inductor doesn’t reach zero at T2,and
the rectifier diodes DR1 and DR4 are on. At the end of this interval, 𝑖𝐿𝑟 (𝑡) = 𝐼𝑃2 . Figure 3.6
illustrates the equivalent circuit for this interval.
54
3.2.3
Case (a): Operating in DCM
Referring to Figure 3.2, during this interval no power is transferred to the secondary.
Accordingly, this interval is a passive interval. In this interval, the parasitic capacitances of
the rectifier diodes resonate with Lr. This resonance appears across the rectifier diodes DR1
and DR4 as illustrated in Figure 3.2. For this interval, current in the resonant inductor remains
zero (𝑖𝐿𝑟 = 0).
Case (b): Operating in BCM
During this interval, the resonant inductor current continues to circulate around the path of
DQ3, resonant inductor (Lr), transformer primary, and Q4, as illustrated in Figure 3.3 and 3.6.
The rate of the downslope of the current through Lr is proportionate to the output voltage Vo.
At T3,the entire energy stored in Lr is transferred to the output, the current becomes zero, and
the rectifier diodes DR1 and DR4 turn-off. The resonant inductor current, 𝑖𝐿𝑟 (𝑡), using initial
condition 𝑖𝐿𝑟 (0) = 𝐼𝑃2 is given by (3).
𝑉𝑜
(𝑡 − 𝑇2 )
𝑛𝐿𝑅
3-3
Case (c): Operating in CCM
Vin
Vo
Q1
Q2
Lr
DR1 DR2
Io
a
Co1
Q3
Figure 3.7
Co2
b
Q4
HV
Battery
DR3 DR4
Equivalent circuit for Interval 3 (T2 – T3) for CCM
55
Referring to Figure 3.4 and Figure 3.7, in CCM at T2, Q3 and Q4 toggle. The timing of this
toggle is dependent on the resonant delay that occurs prior to Q2 turning-on. When Q3 and Q4
toggle, the primary resonant inductor current that was flowing through Q4 finds an alternate
path by charging/discharging the parasitic capacitances of switches Q4 and Q2 until the body
diode of Q2 is forward biased. If the resonant delay is set properly, switch Q2 can be turned
on with ZVS. At T3, the entire energy stored in Lr is transferred to the output, the current
becomes zero, and the rectifier diodes DR1 and DR4 turn-off. The resonant inductor current,
𝑖𝐿𝑟 (𝑡), using initial condition 𝑖𝐿𝑟 (0) = 𝐼𝑃2 is given by (4).
𝑉
(𝑉𝑖𝑛 + 𝑛𝑜 )
(𝑡 − 𝑇2 )
𝐿𝑟
3.2.4
3-4
Interval 4 (T3 – T4) through Interval 6 (T5 – T6)
Intervals 4 to 6 are the negative equivalent of Intervals 1 to 3 as shown in Figures 3.8 to 3.10.
Vin
Vo
Q1
Q2
Lr
DR1 DR2
Io
a
Co1
Q3
Figure 3.8
Co2
b
Q4
DR3 DR4
Equivalent circuit for Interval 4 (T3 – T4) for DCM, BCM and CCM
Vin
Vo
Q1
Q2
Lr
DR1 DR2
a
Co1
Q3
Figure 3.9
HV
Battery
Co2
b
Q4
HV
Battery
DR3 DR4
Equivalent circuit for Interval 5 (T4 – T5) DCM, BCM and CCM and Interval 6 (T5 – T6) for
BCM
56
Vin
Vo
Q1
Q2
Lr
DR1 DR2
Io
a
Co1
Q3
Figure 3.10
3.3
Co2
b
Q4
HV
Battery
DR3 DR4
Equivalent circuit for Interval 6 (T5 – T6) for CCM
Design Procedure
This section provides the details for designing and selecting of various components of the
trailing-edge, PWM full-bridge converter with capacitive output filter, as discussed in the
previous section. Based on the design procedure, a 1.65 kW dc-dc converter stage is designed
to meet the specification of a level-1 charger, as discussed in Table 1.2. The detailed
specifications for designing the dc-dc converter are given in Table 3.1.
Table 3.1
Design specification of the Trailing-edge PWM Full-bridge dc-dc converter with capacitive
filter
3.3.1
Parameters
Value[Units]
380 to 420 [V]
200 to 450 [V]
output voltage)
5.5 [A]
< 4 [Vp-p]
Efficiency
Up to 96 [%]
1.65 [kW]
Selection of Operating Mode
As discussed in the previous section, this converter can operate in DCM, BCM, or CCM.
When the converter is operated in DCM, or BCM, the 50% fixed duty-cycle controlled
switches (Q3 and Q4) can achieve both ZVS turn-on and ZCS turn-off, and also the PWM
controlled switches (Q1 and Q2) can achieve ZCS turn-on. In addition, the secondary-side
57
rectifier diodes can achieve ZCS, which significantly reduces the reverse recovery losses due
to the low di/dt. As an additional benefit, the voltage across the diodes is clamped to the
output voltage, enabling the use of lower breakdown voltage diodes and eliminating the use
of lossy RCD voltage clamps, which are typically required in traditional CCM dc-dc
converters with inductive output filters. When operated in BCM, the converter retains the
advantages of DCM but also has relatively low RMS currents, decreasing conduction loss.
Operation in CCM results in the lowest RMS currents, and ZVS can be achieved for all
switches, but the high di/dt results in large reverse-recovery losses in the secondary-side
rectifier diodes and high voltage ringing. Moreover, to operate this converter in CCM, it
requires a larger resonant inductor which also increases the transformer turns ratio, thus,
increases stress on the primary side switches. Thus, this converter should be designed to
operate in DCM, or BCM.
3.3.2
Selection of Switching Frequency (fs)
An experimental efficiency comparison for the trailing-edge, PWM full-bridge converter is
provided in Figure 3.11 at 1.65 kW for switching frequencies of 100 kHz and 200 kHz. At
100 kHz the converter has maximum overall efficiency. Even though the converter
components, including the resonant inductor and transformer were optimized for 200 kHz
operation, the efficiency at 200 kHz was 2 % lower than at 100 kHz. Thus, a 100 kHz
switching frequency was selected. And finally, the magnetic components were redesigned for
the selected switching frequency of 100 kHz.
58
Efficiency (%)
95
90
100 kHz
85
200 kHz
80
75
206
413
825
1238
1650
DC-DC converter Output Power (W)
Figure 3.11
Comparison of measured efficiency as a function of output power for different switching
frequencies at Vo = 300V and Po = 1.65 kW
3.3.3
The transformer turns ratio nt is calculated using equation 3-5 where Dmax is the maximum
duty-cycle.
𝑛𝑡 =
𝑉𝑜𝑚𝑎𝑥
𝐷𝑚𝑎𝑥 𝑉𝑖𝑛
3-5
The transformer turns ratio is determined to be 1.17 for Vin = 400 V, Vomax = 450 V at
maximum duty cycle of Dmax = 0.96. An EE55 shape ferrite core (using material R from Mag
Inc.) transformer was designed using turns ratio of 12(number of primary turns):14(number
of secondary turns). Two 18 AWG (65 strands of 36 AWG wire) twisted Litz wires were
used for primary and secondary winding.
3.3.4
The converter DC gain in DCM (MDCM) is given by equation 3-6, where n is the transformer
turns ratio; D is the duty cycle; k is the normalized time constant of the converter; Lr is the
resonant inductor, which also includes the leakage inductance of the transformer; Ro is the
load resistance; and T is the switching period.
59
𝑀𝐷𝐶𝑀 =
𝑉𝑜
=
𝑉𝑖𝑛
2𝑛𝑡
1 + √1 +
3-6
4𝑘
𝐷2
The normalized time constant of the converter is given by:
4𝑛𝑡 2 𝐿𝑟
𝑘=
𝑅𝑜 𝑇
3-7
The converter DC gain in BCM is given by:
𝑀𝐵𝐶𝑀 =
𝑉𝑜
= 𝐷𝑛𝑡
𝑉𝑖𝑛
3-8
Using equations 3-5 to 3-8 the design curves are plotted for Gain versus Duty cycle for
various values of k in DCM and BCM as shown in Figure 3.12.
1.25
Vo = 450 V
DCM Gain
k = 0.01
1
BCM Gain
Vo = 300 V
Gain
0.75
DCM Gain
k = 0.33
DCM Gain
k=1
0.5
Vo = 150 V
Design
Operating
Point
0.25
0
Figure 3.12
0
0.25
0.5
Duty Cycle (D)
0.75
1
Design Curve obtained for Gain versus Duty cycle for various values of k in DCM and
BCM
60
To operate the converter in BCM at maximum output current of Io = 5.5 A and Vo = 300 V
(Pomax = 1.65 kW), k = 0.33 is selected as shown in Figure 3.2. Finally, using equation 3-7
and k = 0.33, the resonant inductor Lr = 33 µH is selected.
The 33 µH inductor was designed using a RM12 ferrite core (Material: N97 from Epcos)
with an air gap of 2.1 mm and by winding 18 turns of 19 AWG Type 2 Litz wire (5x46
strands of 42 AWG wire).
3.3.5
Selection of MOSFETs (Q1 – Q4)
The RMS current through the switches Q1 and Q2, IQ12(rms) is given by:
1 𝑇1
𝐼𝑄12(𝑟𝑚𝑠) = √ ∫ 𝑖𝐿𝑟 (𝑡)2 𝑑𝑡
𝑇 𝑇0
3-9
The RMS current through the switches Q3 and Q4, IQ34(rms) is given by:
𝑇3
1 𝑇1
𝐼𝑄34(𝑟𝑚𝑠) = √ [∫ 𝑖𝐿𝑟 (𝑡)2 𝑑𝑡 + ∫ 𝑖𝐿𝑟 (𝑡)2 𝑑𝑡]
𝑇 𝑇0
𝑇1
3-10
The average current through the anti-parallel diodes of switches Q3 and Q4, IDQ34(ave) is given
by:
𝐼𝐷𝑄34(𝑎𝑣𝑒)
1 𝑇3
= ∫ 𝑖𝐿𝑟 (𝑡)𝑑𝑡 = 1.17 [𝐴]
𝑇 𝑇1
3-11
RMS current through the primary switches was calculated to be 4.35 A and 5.42 A using
equation 3-9 and 3-10 for full-load condition (Vin = 400 V, Vo = 300 V and Io = 5.5 A).
A 600V, 190 mΩ Rdson (switch ON state resistance), 20 A, MOSFET (part number:
FCB20N60F from Fairchild) with a fast body diode was selected for the four primary
switches.
61
3.3.6
𝐼𝐷𝑅(𝑎𝑣𝑒) =
𝐼𝑜
2
3-12
Average current through the rectifier diodes was calculated to be 2.75 A using equation 3-12
for Io = 5.5 A.
A 600 V, 8 A hyperfast diode (part number: ISL9R0860 from Fairchild) was selected for the
four rectifier diodes.
3.3.7
Selection of Output filter capacitor (Co2)
The RMS current through the output filter capacitor Co2 is given by equation 3-13 and its
capacitance value is determined using equation 3-14.
=√
1 𝑇𝑃
∫ (𝑖𝑅𝐸𝐶 (𝑡) − 𝐼𝑜 )2 𝑑𝑡 = 3.4 [𝐴]
𝑇𝑃 0
3-13
= 5.4 [𝜇𝐹]
3-14
A 10 µF, 630V film capacitor (part number B32676G6106 from Epcos) was selected for the
𝐶02 =
output filter capacitor.
3.3.8
Selection of Trailing-edge PWM Controller and MOSFET Gate Driver
For implementing the trailing edge PWM gating scheme, ISL6753 PWM controller (from
Intersil) was selected and for driving the MOSFETs Q1 – Q4, IR2110 gate driver (from
International Rectifier) was selected.
62
3.4
Simulation and Experimental Results
The performance of the converter designed in Section 3.3 was evaluated using PSIM
simulation software. Simulations were run for full- and light-load conditions. Circuit
parameters, including component stresses, obtained from theoretical analysis and simulation
are listed in Tables 3.2 at Vin = 400V and Io = 5.5A and 0.7A. As can be observed, there is a
close match between the theoretical prediction and simulation results.
Table 3.2
Comparison of various parameters obtained from simulation and analysis at 5.5 A and 0.7 A
load current and 400 V input voltage
Parameters
Output voltage, Vo (V)
Analysis
Output current, Io (A)
Duty cycle, D (%)
Simulation
300
Analysis
5.5
Simulation
300
0.7
62
63.3
21.9
19.4
4.35
4.5
0.91
0.9
0
0
0
0
Q3, Q4 RMS current IQ34(rms) (A)
5.42
5.3
1.13
1.2
DQ3, DQ4 average current IDQ34(ave) (A)
1.17
1
0.16
0.19
Peak current through Lr, ILrp (A)
13.44
13
4.73
4.65
RMS current through Lr, ILrr (A)
Average current through DR1 – DR4, IDR(ave)
(A)
RMS current through Co, ICo2(rms) (A)
7.65
7.5
1.6
1.65
2.75
2.75
0.35
0.35
3.4
3.24
1.2
1.2
Q1, Q2 RMS current IQ12(rms) (A)
DQ1, DQ2 average current IDQ12(ave) (A)
63
A 1.65 kW experimental prototype was built to verify the operation of the proposed
converter. A photo of the prototype is provided in Figure 3.13.
Resonant
Inductor
Transformer
Output
Capacitor
Primary
MOSFETs
Output
Bridge
Rectifiers
Control
Boards
Figure 3.13
Experimental prototype of 1.65 kW ZVS full-bridge dc-dc converter with capacitive output
filter
Efficiency (%)
entire power range with Vin = 400 V are provided in Figure 3.14.
96
95
94
93
92
91
90
89
88
87
86
Vo = 150V
Vo = 200V
Vo = 300V
Vo = 400 V
0
500
1000
1500
2000
Output Power (W)
Figure 3.14
Experimental measurement of efficiency of the proposed converter as a function of output
power at 400 V input and different output voltages
64
It should be noted that the converter achieves a peak efficiency of 95.7 % at Vo = 400 V, Io =
3 A and output power of 1.2 kW. At maximum output current Io = 5.5 A, Vo = 300 V and
output power of 1.65 kW, the converter achieves an efficiency of 94.9 %. It should be also
noted that below 25 % of output power 1.65 kW, the efficiency reduces drastically. This is
due to turn-on and turn-off switching losses of Q1 and Q2 dominating at lighter loads.
Output Voltage
(Vo)
Output Current
(Io)
Figure 3.15
Experimental waveforms of output voltage and current Ch1= Vo 100 V/div. Ch4= Io 2 A/div.
The experimental waveforms of output voltage and current are shown in Figure 3.15 for Vo =
300 V and Io = 5.5 A. As seen in Figure 3.15, both output voltage and current are nearly free
from low-frequency (120 Hz) ripple. This is one of the important requirements for batterycharging applications.
Figure 3.16, 3.17, and 3.18 provide the experimental waveforms for MOSFET Q3 voltage
and resonant inductor Lr current. Figure 3.16 and 3.17 shows DCM operation at 10 % and 50
% load condition, and Figure 3.18 shows BCM operation at full load. It is noted that the
current in MOSFET Q3 is analyzed using the measured resonant inductor current iLr. The
anti-parallel diode of Q3 conduct current (shown in grey shaded area) prior to conducting
current through the drain-to-source channel of the MOSFET. It can be also observed that
65
before the current flows through channel of the MOSFET, the voltage across the drain to
source of the MOSFET Q3 drops to 0 V, thus enabling it to turn-on with ZVS. As noted, the
current through Q3 reduces to 0 A naturally prior to turning-off, thus enabling it to turn-off
with ZCS.
Drain-Source
Voltage VDS-Q3
ZVS Turn-on
of Q3
Q3 anti-parallel
diode conduction
Figure 3.16
Resonant
Current (ILr)
Gating Signal
VGS-Q3
ZCS Turn-off
of Q3
Experimental waveforms of the MOSFET Q3 voltage and resonant inductor Lr current at
Vin = 400 V, Vo = 300 V, Po = 200 W and fs = 100 kHz. Ch1=VDS-Q3 200 V/div. Ch2= iLr 5 A/div. Ch3= VGSQ3
10 V/div. Time scale=1.16 µs/div.
Drain-Source
Voltage VDS-Q3
ZVS Turn-on
of Q3
Q3 anti-parallel
diode conduction
Resonant
Current (ILr)
Figure 3.17
Gating Signal
VGS-Q3
ZCS Turn-off
of Q3
Experimental waveforms of Figure 3.16 repeated for half-load (800 W) with Vin = 400 V and
Vo = 300 V
66
Drain-Source
Voltage VDS-Q3
ZVS Turn-on
of Q3
Gating Signal
VGS-Q3
Q3 anti-parallel
diode conduction
Resonant
Current (ILr)
Figure 3.18
ZCS Turn-off
of Q3
Experimental waveforms of Figure 3.16 repeated for full-load (1.65 kW) with Vin = 400 V
and Vo = 300 V
Figure 3.19 and 3.20 shows the voltage across and current through rectifier diode DR3 in
DCM and BCM, respectively. As seen, the voltage across the diode is clamped to the output
voltage, at Vo = 300V, and the di/dt through the diode is low enough to minimize the losses
due to reverse-recovery issues inherent with hyperfast diodes.
Rectifier
Diode Voltage
VDR3
Rectifier
Diode Current
IDR3
ZCS Turn-off
of DR3
Figure 3.19
Proposed converter experimental waveforms of the diode DR3 voltage and current at Vin =
400 V, Vo = 300 V, Po = 200 W and fs = 100 kHz. Ch1=VDR3 200 V/div. Ch2= IDR3 5 A/div. Time scale=900
ns/div.
67
Rectifier
Diode Voltage
VDR3
Rectifier
Diode Current
IDR3
ZCS Turn-off
of DR3
Figure 3.20
Experimental waveforms of the diode DR3 voltage and current at Vin = 400 V, Vo = 300 V, Po
= 1650 W and fs = 100 kHz. Ch1=VDR3 100 V/div. Ch2= IDR3 5 A/div. Time scale=900 ns/div.
Also observed, the current through DR3 reduces to 0 A naturally prior to turning-off, thus
enabling it to turn-off with ZCS.
3.5
Figure 3.21 provides an efficiency comparison including a benchmark ZVS full-bridge DCDC converter with inductive output filter and two versions of the proposed converter (with
capacitive output filter) using: ISL9R0860 (Hyperfast diodes) and IDH06S60C Silicon
Carbide (SiC) secondary rectifier diodes. The benchmark converter circuit is illustrated in
Figure 3.22, and a list of its components is provided in Table 3.3.
The overall efficiency of the proposed converter, particularly at light-load conditions, is
much higher than the benchmark converter. The benchmark converter has lower efficiency
due to losses in the secondary-side RCD clamp circuit. The performance of the proposed
converter with hyperfast diodes is very similar to that with SiC diodes. Therefore, this
converter permits use of inexpensive hyperfast diodes, which are typically one quarter of the
cost of SiC diodes.
68
96
94
92
Efficiency (%)
90
88
Vo = 300V SiC Diode
Vo = 300V Hyperfast Diode
Vo = 300V Benchmark Converter
86
84
82
80
78
0
500
1000
1500
2000
Output Power (W)
Figure 3.21
Efficiency comparison for the proposed converter as a function of output power at 400 V
input and 300V output voltage for different rectifier diodes and benchmark converter
Lo
Vin
Q1
Q2
a
Co1
Table 3.3
DR1 DR2
Io
Dc
Co2
b
Q3
Figure 3.22
LLK
Vo
Q4
DR3 DR4
Rc
HV
Battery
Cc
Schematic of the benchmark ZVS full-bridge converter with inductive output filter
Components Used In the Benchmark Converter
1
Parameters
Q1-Q4
Value [Units]
FCB20N60F [each]
2
DR1-DR4
IDH06S60C [each]
3
Transformer turns ratio
1.22
4
Transformer Leakage Inductance
1.6 [µH]
5
Output Inductor
600 [µH]
6
DC-DC Switching Frequency
70 [kHz]
69
3.6
Conclusions
A new full-bridge dc-dc converter with capacitive output filter operating with trailing-edge
PWM gating has been presented in this chapter for the dc-dc stage in a PHEV battery
charger. The proposed converter has been presented with detailed analysis, design and
experimental results. It has been shown through analysis and experimental results that this
converter overcomes all the major issues such as, duty-cycle loss, high voltage rectifier
ringing and circulating currents in primary side switches present in the converter with
inductive output filter, as presented in the previous chapter. This converter also achieves high
efficiency and the output voltage and currents waveforms are free from 120 Hz AC ripple.
All the primary MOSFETs and secondary rectifier operate with soft-switching. This
converter also permits use of inexpensive hyperfast diode, since its performance is very
similar to that of SiC rectifier diode.
For higher power application (> 2 kW), this converter could suffer from high peak current
stress in the primary MOSFETs, which may cause thermal management issues and
compromise the reliability of the converter. In order to overcome these issues, an interleaved,
multi-cell, full-bridge dc-dc converter with capacitive output filter operating with trailingedge PWM is presented in the next chapter. This configuration not only reduces the stress on
the devices, but also aids in reducing the size of the input and output filter components.
70
Chapter 4: An Interleaved Full-Bridge DC-DC Converter with Capacitive
Output Filter Operated with Trailing-Edge PWM Gating4
4.1
Introduction
This chapter presents an interleaved full-bridge dc-dc converter with capacitive output filter
operating with trailing-edge PWM gating scheme, as discussed in chapter 1, for use in the dcdc converter stage of a PHEV onboard battery charger.
In chapter 3, the trailing-edge, PWM full-bridge dc-dc converter with capacitive output filter
was presented. As discussed in the last chapter, this converter overcomes the main issues of
duty-cycle loss, high-voltage ringing on the rectifier diodes and circulating currents in the
primary side, which are present in the full-bridge converter with inductive output filter, as
discussed in chapter 2. It was also shown in chapter 3 that the full-bridge converter with
capacitive filter significantly improved the light-load efficiency and also permitted the use of
inexpensive hyperfast rectifier diodes, while reducing the size of the converter by using
fewer components.
As discussed in section 1.5 of chapter 1, an interleaved, multi-cell configuration that uses ‘n’
number of cells in parallel (both at the input and output) with each cell being phase-shifted
by 360o/n for high power application is an interesting approach.
Due to interleaving, each cell shares equal power, and the thermal losses are distributed
uniformly among the cells, and also, the input/output ripple frequency of multi-cell
configuration becomes ‘2n’ times the switching frequency of each cell, which reduces the
4
and W.G. Dunford, "An Interleaved Zero Voltage Switching Full-Bridge DC-DC Converter with Capacitive
Output Filter for a Plug-in-Hybrid Electric Vehicle Battery Charger," Proceedings of IEEE Energy Conversion
Congress and Exposition (ECCE 2012), Raleigh, pp. 2827-2832, September 2012].
71
filter size and cost. Since there is no detailed analysis and step-by-step design procedure
available in the literature for this configuration, this chapter presents a 2-cell, interleaved,
full-bridge dc-dc converter with capacitive output filter operating with trailing-edge PWM
gating scheme.
The layout of the chapter is as follows. Section 4.2 explains the operating principle; section
4.3 gives the design procedure for selecting various components and devices. Based on this
design method, a 3.3 kW, 100 kHz, dc-to-dc converter is designed. PSIM simulation and
experimental results are presented in Sections 4.4. Finally, performance evaluation of this
converter with a benchmark converter is presented in Section 4.5.
4.2
Operating Principle
The proposed interleaved, 2-cell, full-bridge dc-dc converter topology is illustrated in Figure
4.1. As shown in Figure 4.1, each cell A and B is the basic full-bridge converter as described
in the previous chapter with the inputs and outputs of each cell connected in parallel. In order
to obtain higher output power more cells could be connected in parallel in a similar fashion.
Vin
Q1
Vg1
Co1
iLr
Vab
Q3
Q4
Q1
Q4
Vg4
iCo2
VRec_in
Co2
Io
HV
Battery
DR3 DR4
Cell-B
Lr
DR1 DR2
iRect
HF
Transformer
iLr
Vab
Figure 4.1
Isec
1:nt
Vg2
a
DR1 DR2
b
Q2
Vg1
Vo
iRect
HF
Transformer
Vg4
Vg3
Vg3
Lr
Vg2
a
Q3
Cell-A
Q2
Isec
b
VRec_in
1:nt
DR3 DR4
A 2-cell interleaved trailing-edge PWM full-bridge converter with capacitive output filter
72
Vg1A
Vg2A
Vg3A
Vg4A
Vg1B
Vg2B
Vg3B
Vg4B
+400V
iLrA
vabA
-400V
iLrB
+400V
vabB
-400V
iin
iCo2
Time (µs)
Figure 4.2
Typical operating waveforms to illustrate the operation of the trailing-edge PWM 2-cell,
interleaved, full-bridge converter in DCM mode
73
Vg1A
Vg2A
Vg3A
Vg4A
Vg1B
Vg2B
Vg3B
Vg4B
+400V
iLrA
vabA
-400V
iLrB
+400V
vabB
-400V
iin
iCo2
Time (µs)
Figure 4.3
Typical operating waveforms to illustrate the operation of the trailing-edge PWM 2-cell,
interleaved, full-bridge converter in BCM mode
74
Figure 4.2 and 4.3 shows the operating waveforms of the converter when operated in
discontinuous conduction (DCM) and boundary conduction modes (BCM) respectively. The
operating principle of the converter individual cells in DCM and BCM modes is same as
presented in section 3.2 of chapter 3 and is not discussed here. From Figure 4.2 and 4.3, it
can be clearly seen that the input (iin) and output capacitor ripple (iCo2) current frequency is
four times the switching frequency.
Although the proposed converter can operate in DCM, BCM, or continuous conduction mode
(CCM), only the DCM and BCM modes are desirable for the present application, as
explained in section 3.3.1. Operation in CCM results in the lowest RMS currents, and ZVS
can be achieved for all switches, but the high di/dt results in large reverse-recovery losses in
the secondary-side rectifier diodes and high-voltage ringing. Moreover, to operate this
converter in CCM requires a larger resonant inductor which also increases the transformer
turns ratio and increases stress on the primary-side switches. Hence, this converter should be
designed to operate in DCM, or BCM.
4.3
Design Procedure
This section provides the details for designing and selection of various components of the
trailing-edge PWM, 2-cell, interleaved, full-bridge converter, as discussed in the previous
section. Based on the design procedure, a 3.3 kW dc-dc converter stage is designed to meet
the specification of a level-2 charger as discussed in Table 1.2. The detailed specifications for
designing the dc-dc converter are given in Table 4.1.
In order to design a 2-cell interleaved dc-dc converter, it can be treated as two separate ZVS
dc-dc converters; each operating at half of the load power rating. With this approach,
75
selection of operating modes, switching frequency (fs), and all equations for selecting the
transformer turns ratio (nt), resonant inductor (Lr), MOSFETs (Q1 - Q4), and rectifier diodes
(DR1 – DR4) in the full-bridge dc-dc converter with capacitive output filter (as discussed in
section 3.3) remains valid, since the stresses are unchanged with the only exception being the
reduced ripple current through the input and output filter capacitors.
Table 4.1
4.3.1
Design specification of the Trailing-edge PWM Full-bridge dc-dc converter
Parameters
Value[Units]
380 to 420 [V]
200 to 450 [V]
output voltage)
11 [A]
< 4 [Vp-p]
Efficiency
Up to 96 [%]
3.3 [kW]
HF Transformer Design
An ER16x25x49 shape ferrite core (using material TP4D from TDG Cores) transformer was
designed using turns ratio of 12(number of primary turns):14(number of secondary turns) to
achieve a turns ratio nt of 1.17, as calculated in section 3.3.3 using Equation 3-5. Two 18
AWG (400 strands of 44 AWG wire) twisted Litz wires were used for primary winding and
two 20 AWG (165 strands of 42 AWG wire) twisted Litz wires were used for secondary
winding.
4.3.2
Selection of Output Filter Capacitor (Co2)
The worst case scenario of RMS current through the output filter capacitor Co2 is given in
equation 4-1, and its capacitance value is determined using equation 4-2.
76
𝐼𝐶𝑜2(𝑟𝑚𝑠) = √
1 𝑇𝑃
∫ (𝑖𝑅𝐸𝐶 (𝑡) − 𝐼𝑜 )2 𝑑𝑡 = 3.4 [𝐴]
𝑇𝑃 0
𝐶02 =
= 5.4 [𝜇𝐹]
4-1
4-2
A 10 µF, 630V, film capacitor (part number: B32676G6106 from Epcos) was selected for the
output filter capacitor. It should be noted here that the capacitor used is the same as the one
used in chapter 3, but for 2x output power due to interleaving.
The various components selected for the circuit are listed in Table 4.2.
Table 4.2
Components Selection
Parameters
Q1A-Q4A and Q1B-Q4B
Value [Units]
FCB20N60F [each]
DR1A-DR4A and DR1B-DR4B
ISL9R0860 [each]
LrA and LrB
Output Capacitor
33 [µH]
1.17
10 [µF]
77
4.4
The 3.3 kW 2-cell interleaved dc-dc converter designed in the previous section was simulated
using PSIM software for Vo = 300 V and load current Io = 11 and 1 A. Typical HF
waveforms obtained using PSIM simulation for the converter with an input voltage Vin = 400
V at full load and 10% load are shown in Figure 4.4 and 4.5, respectively.
Figure 4.4
Simulation results of resonant inductor LrA and LrB with current through the output filter
capacitor Co2 at Vin = 400 V and Vo = 300 V and Io = 1 A
Figure 4.5
Simulation results of Figure 4.4 repeated at at Vin = 400 V and Vo = 300 V and Io = 11 A
78
As seen in Figure 4.4 and 4.5, at lighter load, both the cells operate in DCM and in BCM at
full-load condition, respectively. Also both the resonant inductors equally share the currents,
and the frequency of the ripple current in the output filter capacitor is four times the
switching frequency.
A 2-cell 3.3 kW experimental prototype was built to verify the operation of the proposed
Output
Capacitor
Transformer
A
Full-bridge
MOSFETs A
Resonant
Inductor A
Figure 4.6
Rectifier
Diodes B
Control
Board
Full-bridge
MOSFETs B
Transformer B
Resonant
Inductor B
Experimental prototype of 3.3 kW, 2-cell, interleaved, full-bridge dc-dc converter with
capacitive output filter
79
Gatedrive signals
HV
Battery
Primary current
sense
Gatedrive
signals
Primary
current
sense
Ri
Ci
Rv
Cv
Peak I Mode
PWM #1
Iloop
Clamp
Peak I Mode
PWM #2
External Clock
Synchronising
Circuit
Figure 4.7
Vloop
Vcmd
Icmd
An inner-loop, current-sharing control scheme
The feedback control scheme for the proposed converter configuration is shown in Figure
4.7. An inner-loop, current-sharing control scheme is used to achieve current sharing among
both the cells. Inner-loop current-sharing is inherently peak current-mode control. The output
of the current/voltage compensator serves as the current-sharing bus and provides the output
current reference for both the cells. For interleaving, an external clock synchronizing circuit
is used to phase-shift cell B by 180° with respect to cell A.
entire power range with Vin = 400 V are provided in Figure 3.14. It should be noted that the
converter achieves a peak efficiency of 95.7 % at Vo = 400 V, Io = 6 A and output power of
2.4 kW. At maximum output current Io = 11 A, Vo = 300 V, and output power of 3.3 kW the
converter achieves an efficiency of 95 %.
80
Efficiency (%)
96
95
94
93
92
91
90
89
88
87
86
Vo = 150V
Vo = 200V
Vo = 300V
Vo = 400 V
0
1000
2000
3000
4000
Output Power (W)
Figure 4.8
Experimental measured efficiency of the proposed converter as a function of output power at
400 V input and different output voltages
It should be also noted that below 25 % of the output power of 1.65 kW the efficiency
reduces drastically. This is due to turn-on and turn-off switching losses of Q1 and Q2
dominating at lighter loads. The light-load efficiency can be significantly improved by
completely turning-off a cell below 50% of rated load power.
Experimental waveforms of the dc-dc converter in DCM and BCM mode are provided in
Figure 4.9 and 4.10. It is noted that the MOSFET Q3B turns on with ZVS and turns off with
ZCS, and the current through the transformer secondary winding also has a very low di/dt. It
is also noted that both the cells equally share the load current, which aids in distributing
thermal losses between the two cells and thus helps in improving efficiency.
81
Transformer B Sec.
Winding Current
Transformer A Sec.
Winding Current
Drain-Source
Voltage VDS-Q3B
Gating Signal
VGS-Q3B
ZVS Turn-on of Q3
Figure 4.9
Experimental waveforms of the MOSFET Q3B voltage and transformer secondary winding
current at Vin = 400 V, Vo = 300 V, Po = 300 W and fs = 100 kHz. Ch1=VDS-Q3B 200 V/div. Ch2= VGS-Q3 10
V/div. Ch3= Tx. B Sec. current 2 A/div. Ch4= Tx. A Sec. current 2 A/div.Time scale=2 µs/div.
Transformer B Sec.
Winding Current
Drain-Source
Voltage VDS-Q3B
Transformer A Sec.
Winding Current
Gating Signal
VGS-Q3B
ZVS Turn-on of Q3
Figure 4.10
Experimental waveforms of Figure 4.10 repeated for Vin = 400 V, Vo = 300 V, Po = 3300 W
and fs = 100 kHz. Ch1=VDS-Q3B 200 V/div. Ch2= VGS-Q3 10 V/div. Ch3= Tx. B Sec. current 10 A/div. Ch4=
Tx. A Sec. current 10 A/div.Time scale=2 µs/div.
82
4.5
An efficiency comparison of the proposed converter with the benchmark interleaved ZVS
full-bridge dc-dc converter with inductive output filter is provided in Figure 4.11. The
benchmark converter was also operated with trailing-edge PWM gating. The benchmark
converter circuit is illustrated in Figure 4.12, and the list of components used is provided in
Table 4.3. The overall efficiency of the proposed converter, particularly at light load
conditions, is much higher than the benchmark counterpart. The benchmark converter has
lower efficiency due to losses in the secondary-side RCD clamp circuit.
96
94
92
Efficiency (%)
90
88
86
Vo = 300V Proposed Converter
84
Vo = 300V Benchmark Converter
82
80
78
0
1000
2000
3000
4000
Output Power (W)
Figure 4.11
input and 300V output voltage and benchmark converter
83
LOA
Vin
Q2A
Q1A
Co1
LLKA
a
Vo
Io
DCA
DR1A DR2A
Co2
b
CCA
RCA
Q4A
Q3A
HV
Battery
DR3A DR4A
Q2B
Q1B
a
LLKB
LOB
DCB
DR1B DR2B
b
Q3B
CCB
RCB
Q4B
DR3B DR4B
Figure 4.12
Benchmark 2-Cell Interleaved PWM ZVS full-bridge converter topology with inductive
output filter
Table 4.3
Components Used In the Benchmark Converter
Parameters
Q1A-Q4A and Q1B-Q4B
Value [Units]
FCB20N60F [each]
ISL9R0860 [each]
1.6 [µH]
Output Inductor
600 [µH]
DC-DC Switching Frequency
4.6
1.22
70 kHz]
Conclusions
An interleaved, 2-cell, full-bridge dc-dc converter with capacitive output filter operating with
trailing-edge PWM gating has been presented in this chapter. The proposed converter has
been analyzed in BCM and DCM modes. A 2-cell, 3.3 kW dc-dc converter laboratory
prototype was build based on the step-by-step procedure presented in the chapter. It has been
shown that both the cells share the total output power equally, thereby equally sharing the
power losses between the two cells. It was also shown that by interleaving the ripple
84
frequency in the input and output filter capacitors are doubled which aids in reducing the size
of the filter components.
In order to reduce the number of rectifier diodes by half (resulting in lower cost and overall
lower converter size), a new topology, an interleaved, multi-cell, full-bridge dc-dc converter
with voltage-doubler rectifier and capacitive output filter operating with trailing-edge PWM
gating is presented in the next chapter.
85
Chapter 5: An Interleaved, Full-Bridge DC-DC Converter with VoltageDoubler Rectifier and Capacitive Output Filter Operated with TrailingEdge PWM Gating5
5.1
Introduction
This chapter presents an interleaved, full-bridge dc-dc converter with voltage doubler
rectifier and capacitive output filter operating with trailing-edge PWM gating scheme, as
discussed in chapter 1, for use in the dc-dc converter stage of a PHEV on-board battery
charger.
In chapter 3, the trailing-edge PWM full-bridge dc-dc converter with capacitive output filter
was presented. It was shown in chapter 3 that the full-bridge converter with capacitive filter
significantly improved the light-load efficiency and also permitted the use of inexpensive
hyperfast rectifier diodes. This reduced the size of the converter by using fewer components.
For higher power application, an interleaved, multi-cell configuration that uses two cells in
parallel (both at the input and output) was presented in chapter 4. It was shown in chapter 4,
that due to interleaving, each cell shares equal power and the thermal losses are distributed
uniformly among the cells. The input/output ripple frequency becomes 4 times the switching
frequency of each cell which reduces the filter size and cost.
In order to further reduce the size and cost of the converter configuration presented in chapter
4, a multi-cell, interleaved, full-bridge DC-DC converter with capacitive filter and voltagedoubler rectifier operated with trailing-edge PWM gating scheme is an attractive solution for
5
and W.G. Dunford, "An Isolated Interleaved DC-DC Converter with Voltage Doubler Rectifier for PHEV
Battery Charger, "Proceedings of IEEE Applied Power Electronics Conference and Exposition (APEC 2013),
Long Beach, pp. 3067-3072, Mar. 2013].
86
the present application. The output voltage-doubler rectifier reduces half the number of
secondary diodes (resulting in lower cost and overall lower converter size) as compared to
the topology presented in the previous chapter.
Since there is no detailed analysis and step-by-step design procedure available in the
literature for this configuration, this chapter presents a 2-cell, interleaved, full-bridge dc-dc
converter with voltage-doubler rectifier and capacitive output filter operating with trailingedge PWM gating scheme.
The layout of the chapter is as follows. Section 5.2 first explains the operating principle of
the basic full-bridge dc-dc converter with voltage-doubler rectifier and later presents the
proposed interleaved converter with voltage-doubler rectifier; section 5.3 gives the design
procedure for selecting various components and devices. Based on this design method, a 3.3
kW, 100 kHz, dc-to-dc converter is designed. PSIM simulation and experimental results are
presented in Sections 5.4. Finally, performance evaluation of this converter with benchmark
converters is presented in Section 5.5.
5.2
Operating Principle
Vin
Q1
Vg1
Cin
iLr
Vab
Q3
Figure 5.1
Lr
Vg2
a
Vg3
Vo
Q2
Q4
Vg4
DR1
HF
Transformer
iCo1
Co1
Isec
Io
HV
Battery
b
1:nt
DR2
Co2
iCo2
Trailing-edge PWM Full-bridge dc-dc converter with voltage-doubler rectifier and
capacitive output filter
The circuit diagram of the full-bridge dc-dc converter with voltage-doubler rectifier and
capacitive output filter operating with trailing-edge PWM gating scheme is shown in Figure
87
5.1. Detailed operating waveforms are provided for DCM in Figure 5.2 and for BCM in
Figure 5.3.
Vg1
Vg2
Vg3
Vg4
+400V
IP1
vab
iLr
-400V
vo
vDR1
T
TP
iDR1
iCo1
Intervals
Devices
1
3
Q1,Q4,
DQ3,Q4,DR1
DR1
T0
Figure 5.2
2
T1
4
5
6
Q2,Q3,DR2 DQ4,Q3,DR2
T2 T3
T4
T5 T6
Time (µs)
converter with voltage doubler-rectifier in DCM mode
88
Vg1
Vg2
Vg3
Vg4
+400V
IP1
iLR
IP2
vab
iLr
-400V
T
vo
vDR1
TP
iDR1
TP
iCo1
Intervals
Devices
T0
Figure 5.3
1
2
3
Q1,Q4, DQ3,Q4,
DR1
DR1
T1
4
5
Q2,Q3,DR2
DQ4,Q3,
DR2
DQ3,Q4,DR1
T2 T3
T4
6
T5 T6
DQ4,Q3,DR2
Time (µs)
converter with voltage-doubler rectifier in BCM mode
89
The operation of this converter can be explained in the same way as the converter presented
in chapter 3; the only exception being the converter in Figure 5.1 has only two rectifier
diodes (DR1 and DR2) and two voltage divider output filter capacitors (Co1 and Co2).
Figure 5.4 and 5.5 shows the operation of the circuit for interval 1, 2 and 3 during DCM and
BCM operating modes.
Vin
Q1
Vg1
Cin
iLr
Vab
Q3
Figure 5.4
Lr
Vg2
a
Vg3
Vo
Q2
Q4
DR1
HF
Transformer
iCo1
Co1
Isec
Io
HV
Battery
b
1:nt
Vg4
DR3
Co2
iCo2
Equivalent circuit for Interval 1 for DCM and BCM
During Interval 1, switches Q1 and Q4 are on and Q2 and Q3 are off. This is a power transfer
interval, and the primary current flows through Q1, resonant inductor (Lr), transformer
primary, and Q4, as illustrated in Figure 5.4. The rate of rise of the current (di/dt) through Lr
is proportionate to the difference between the input voltage Vin and the output voltage Vo.
On the secondary-side, the current flows out of the secondary winding of the transformer
through diode DR1, filter capacitor Co1, and back to the winding of the transformer. Capacitor
Co1 and Co2 also supplies the load current Io to the HV battery.
Referring to Figure 5.5, interval 2 begins after switch Q1 turns off, as determined by the
PWM duty cycle. Since the current flowing in the primary side cannot be interrupted
instantaneously, it finds an alternate path and flows through the parasitic switch capacitances
of Q3 and Q1, which discharges the node ‘a’ to 0V and then forward biases the body diode
DQ3. During this switch transition, the energy stored in the resonant inductor (Lr) assists in
transferring energy from the lower to upper bridge MOSFET capacitance. Therefore switches
90
Q3 and Q4 always achieve ZVS with the help of the energy stored in the resonant inductor
(Lr) for nearly the entire load current (Io) range.
Vin
Q1
Vg1
Cin
iLr
Vab
Q3
Figure 5.5
Lr
Vg2
a
Vg3
Vo
Q2
Q4
DR1
HF
Transformer
iCo1
Co1
Isec
HV
Battery
b
Vg4
Io
1:nt
DR2
Co2
iCo2
Equivalent circuit for Interval 2 for DCM and Interval 2 and 3 for BCM
During interval 2 the energy stored in Lr is transferred to the output. The primary resonant
inductor (Lr) maintains the current, which circulates around the path of body diode of Q3,
resonant inductor (Lr), transformer primary, and Q4, as illustrated in Figure 5.5. The rate of
the downslope of the current through Lr is proportionate to the output voltage Vo. The only
difference in DCM is that at T2 the energy stored in Lr is transferred to the output, the current
becomes zero, and the rectifier diodes DR1 and DR4 turn-off as illustrated in Figure 5.2. In
BCM, the current through the resonant inductor doesn’t reach zero at T2, and the rectifier
diode DR1 is still on.
Referring to Figure 5.2, in DCM, during interval 3 no power is transferred to the secondary.
During this interval, the parasitic capacitances of the rectifier diodes resonate with LR as
illustrated in Figure 5.2, and the current in the resonant inductor remains zero (𝑖𝐿𝑟 = 0).
Referring to Figure 5.3 and 5.5 in BCM, during interval 3 the resonant inductor current
continues to circulate around the path of DQ3, resonant inductor (Lr), transformer primary,
and Q4. The rate of the downslope of the current through Lr is proportionate to the output
voltage Vo. At T3, the entire energy stored in Lr is transferred to the output, the current
becomes zero, and the rectifier diode DR1 turns off.
91
The proposed interleaved, 2-cell, full-bridge dc-dc converter with voltage-doubler rectifier
topology is illustrated in Figure 5.6. As shown in Figure 5.6, each cell A and B is the basic
full-bridge converter with voltage-doubler rectifier as shown in Figure 5.1, where both the
inputs and outputs of each cell are connected in parallel.
Vin
Q1
Vg1
Cin
iLr
Vab
Q3
Q4
Q1
Figure 5.6
Isec
DR3
Q4
iCo1
I
Co1 o
HV
Battery
DR4
Co2
iCo2
Cell-B
Lr
HF
Transformer
iLr
Vab
DR2
1:nt
Vg2
a
DR1
b
Q2
Vg1
Vo
HF
Transformer
Vg4
Vg3
Q3
Lr
Vg2
a
Vg3
Cell-A
Q2
Isec
b
1:nt
Vg4
A 2-cell, interleaved, trailing-edge PWM, full-bridge converter with voltage-doubler rectifier
and capacitive output filter
Figure 5.7 and 5.8 shows the operating waveforms of the converter when operated in
discontinuous conduction (DCM) and boundary conduction modes (BCM) respectively. The
operating principle of the converter individual cells in DCM and BCM modes is same as
presented in beginning of this section 5.2 of this chapter. From Figure 5.7 and 5.8, it can be
clearly seen that the input current (iin) frequency is four times of the switching frequency, but
the output capacitor ripple (iCo2) is the same as the switching frequency. This interleaving
configuration doesn’t offer any benefits for output filter capacitor size reduction.
92
Vg1A
Vg2A
Vg3A
Vg4A
Vg1B
Vg2B
Vg3B
Vg4B
+400V
iLrA
vabA
-400V
+400V
iLrB
vabB
-400V
iin
iCo2
Time (µs)
Figure 5.7
Typical operating waveforms to illustrate the operation of the trailing-edge PWM, 2-cell,
interleaved, full-bridge converter with voltage-doubler rectifier in DCM mode
93
Vg1A
Vg2A
Vg3A
Vg4A
Vg1B
Vg2B
Vg3B
Vg4B
+400V
iLrA
vabA
-400V
+400V
iLrB
vabB
-400V
iin
iCo2
Time (µs)
Figure 5.8
Typical operating waveforms to illustrate the operation of the trailing-edge PWM, 2-cell,
interleaved, full-bridge converter with voltage-doubler rectifier in BCM mode
94
Although the proposed converter can operate in DCM, BCM, or continuous conduction mode
(CCM), only the DCM and BCM modes are desirable for the present application, as
explained in section 3.3.1. Operation in CCM results in the lowest RMS currents and ZVS
can be achieved for all switches, but the high di/dt results in large reverse-recovery losses in
the secondary side rectifier diodes and high-voltage ringing. Moreover, to operate this
converter in CCM, requires a larger resonant inductor, which also increases the transformer
turns ratio and increases stress on the primary-side switches. Thus, this converter should be
designed to operate in DCM, or BCM.
5.3
Design Procedure
This section provides the details for designing and selection of various components for the
trailing-edge PWM 2-cell interleaved full-bridge converter with voltage doubler rectifier. As
discussed in the previous section based on the design procedure, a 3.3 kW dc-dc converter
stage is designed to meet the specification of a level-2, charger as discussed in Table 1.2 of
chapter 1. The detailed specifications for designing the dc-dc converter are given in Table
5.1.
In order to design a 3.3 kW 2-cell interleaved dc-dc converter, it can be treated as two
separate ZVS dc-dc converters with each operating at half of the load power rating (1.65
kW). With this approach, selection of operating modes, switching frequency (fs), and all
equations for selecting the MOSFETs (Q1 - Q4) in the full-bridge dc-dc converter with
capacitive output filter (as discussed in section 3.3 of chapter 3) remains valid. The procedure
to select the transformer turns ratio (nt), resonant inductor (Lr), rectifier diodes (DR1 – DR2),
and output filter capacitor (Co1 – Co2) is presented here.
95
Table 5.1
Design specification of the Trailing-edge PWM Full-bridge dc-dc converter
Parameters
Value[Units]
380 to 420 [V]
200 to 450 [V]
output voltage)
Efficiency
5.3.1
11 [A]
3.3 [kW]
< 4 [Vp-p]
Up to 96 [%]
The transformer turns ratio nt is calculated using equation 5-1 where Dmax is the maximum
duty-cycle.
𝑛𝑡 =
𝑉𝑜𝑚𝑎𝑥
2𝐷𝑚𝑎𝑥 𝑉𝑖𝑛
5-1
The transformer turns ratio is determined to be 0.6 for Vin = 400 V, Vomax = 450 V at
maximum duty cycle of Dmax = 0.96. An ER16x25x49 shape ferrite core (using material
TP4D from TDG Cores) transformer was designed using turns ratio of 10(number of primary
turns):6(number of secondary turns). Two 19 AWG (360 strands of 44 AWG wire) twisted
Litz wires were used for primary winding and three 18 AWG (400 strands of 44 AWG wire)
Litz wires were used for secondary winding.
5.3.2
The converter DC gain in DCM (MDCM) is given by equation 5-2 and 5-3, where nt is the
transformer turns ratio; D is the duty cycle; k is the normalized time constant of the
converter; Lr is the resonant inductor, which also includes the leakage inductance of the
transformer; Ro is the load resistance; and T is the switching period.
96
𝑀𝐷𝐶𝑀 =
𝑉𝑜
=
𝑉𝑖𝑛
4𝑛𝑡
1 + √1 +
5-2
16𝑘
𝐷2
The normalized time constant of the converter is given by:
4𝑛𝑡 2 𝐿𝑟
𝑘=
𝑅𝑜 𝑇
5-3
The converter DC gain in BCM is given by:
𝑀𝐵𝐶𝑀 =
𝑉𝑜
= 2𝐷𝑛𝑡
𝑉𝑖𝑛
5-4
Using equations 5-1 to 5-4 the design curves are plotted for Gain versus Duty cycle for
various values of k in DCM and BCM, as shown in Figure 3.12.
1.25
Vo = 450 V
1
DCM Gain
k = 0.0025
BCM Gain
Vo = 300 V
Gain
0.75
DCM Gain
k = 0.089
DCM Gain
k = 0.25
0.5
Vo = 150 V
Design
Operating
Point
0.25
0
Figure 5.9
0
0.25
0.5
Duty Cycle (D)
0.75
1
Design Curve obtained for Gain versus Duty cycle for various values of k in DCM and BCM
97
To operate an individual converter cell in BCM at maximum output current of Io = 5.5A and
Vo = 300V (Pomax = 1.65kW), k = 0.089 is selected as shown in Figure 5.9. Finally, using
Equation 5-3 and k = 0.089, the resonant inductor Lr = 33 µH is selected.
The 33 µH inductor was designed using a RM12 ferrite core (Material: N97 from Epcos)
with an air gap of 2.1 mm and by winding 18 turns of 19 AWG Type 2 Litz wire (5x46
strands of 42 AWG wire).
5.3.3
𝐼𝐷𝑅(𝑎𝑣𝑒) = 𝐼𝑜
5-5
Average current through the rectifier diodes was calculated to be 5.5 A using equation 5-5 for
Io = 5.5 A for individual converter cell. A 600 V, 15 A hyperfast diode (part number:
ISL9R1560 from Fairchild) was selected for the four rectifier diodes.
98
5.3.4
Selection of Output Filter Capacitors (Co1 and Co2)
The worst case ripple current through the output filter capacitors Co1 and Co2 is 9A rms, and
the maximum voltage across the capacitor is 225 V DC. Four 2.2 µF, 250 V ceramic
capacitors from Murata (part number: KC355WD72E225M) are connected in parallel to
obtain 8.8 µF. The ripple current rating for this capacitor (part number:
KC355WD72E225M) is 5A rms at 100 kHz. A photo showing paralleled capacitors for
realizing output filter capacitor Co1 and Co2 is shown in Figure 5.10.
Figure 5.10
Output filter capacitor C01 and C02
The various components selected for the circuit are listed in Table 5.2.
Table 5.2
Components Selection
Parameters
Q1A-Q4A and Q1B-Q4B
Value [Units]
FCB20N60F [each]
ISL9R0860 [each]
LrA and LrB
32 [µH]
1.17
0.7 [µH]
Output Capacitor
8.8 [µF]
99
5.4
The 3.3 kW 2-cell interleaved dc-dc converter with voltage-doubler rectifier design described
in the previous section was simulated using PSIM software for Vo = 300 V and load current Io
= 1 and 11 A. Typical HF waveforms obtained using PSIM simulation for the converter with
an input voltage Vin = 400 V at full load and 10% load are shown in Figure 5.11 and 5.12,
respectively.
Figure 5.11
Simulation results of resonant inductor LrA and LrB with current through the output filter
capacitors Co1 and Co2 at Vin = 400 V and Vo = 300 V and Io = 1 A
Figure 5.12
Simulation results of Figure 5.11 repeated at Vin = 400 V and Vo = 300 V and Io = 11 A
100
As seen in Figure 5.11 and 5.12, at lighter load, both the cells operate in DCM and in BCM
at full-load condition, respectively. Also, both the resonant inductors equally share the
currents, and the frequency of the ripple current in the output filter capacitor is same as the
switching frequency. Figure 5.13 and 5.14 shows the simulation results of voltage across and
current through rectifier diode DR2A and DR2B in DCM and BCM, respectively. As seen, the
voltage across the diode is clamped to the output voltage, at Vo = 300 V, and the di/dt
through the diode is low enough to minimize the losses due to reverse-recovery issues
inherent with hyperfast diodes.
Figure 5.13
VDR2A
I(DR2A)*20
VDR2B
I(DR2B)*20
Simulation results of voltage across and current through output rectifier diodes DR2A and
DR2B at Vin = 400 V and Vo = 300 V and Io = 1 A
Figure 5.14
VDR2A
I(DR2A)*20
VDR2B
I(DR2B)*20
Simulation results of Figure 5.13 repeated at Vin = 400 V and Vo = 300 V and Io = 11 A
101
A 2-cell, 3.3 kW experimental prototype was built to verify the operation of the proposed
Figure 5.15
Experimental prototype of 3.3 kW 2-cell interleaved full-bridge dc-dc converter with
voltage-doubler rectifier and capacitive output filter
Gatedrive signals
HV
Battery
Primary current
sense
Gatedrive
signals
Primary
current
sense
Ri
Ci
Rv
Peak I Mode
PWM #1
Iloop
Peak I Mode
PWM #2
External Clock
Synchronising
Circuit
Figure 5.16
Clamp
Vloop
Cv
Vcmd
Icmd
An inner-loop current-sharing control scheme
102
The feedback control scheme for the proposed converter configuration is shown in Figure
5.16, and, as can be seen, its implementation is similar to one presented in section 4.4 of the
previous chapter.
entire power range with Vin = 400 V are provided in Figure 5.17. It should be noted that the
converter achieves a peak efficiency of 96 % at Vo = 400 V, Io = 7 A and output power of 2.8
kW. At maximum output current Io = 11 A, Vo = 300 V and output power of 3.3 kW, the
converter achieves an efficiency of 95.2 %.
98
96
Efficiency (%)
94
92
90
Vo = 400V
Vo = 300V
Vo = 200V
Vo = 150V
88
86
84
82
80
0
Figure 5.17
1000
2000
Output Power (W)
3000
Experimental measured efficiency of the proposed converter as a function of output power
at 400 V input and different output voltages
It should be also noted that below 25 % of the output power of 1.65 kW the efficiency
reduces drastically; this is due to turn-on and turn-off switching losses of Q1 and Q2
dominates at lighter load. The light-load efficiency can be significantly improved by
completely turning-off a cell below 50% of rated load power.
103
Experimental waveforms of the dc-dc converter in DCM and BCM mode are provided in
Figure 5.18 and 5.19. It is noted that the MOSFET Q3B turns on with ZVS and turns off with
ZCS, and the current through the resonant inductor also has a very low di/dt. It is also noted
that both the cells equally share the load current, which aids in distributing thermal losses
between the two cells and helps in improving efficiency.
Resonant Inductor Resonant Inductor
LRA Current
LRB Current
Drain-Source
Voltage VDS-Q3B
ZCS Turn-off of Q3
Q3 anti-parallel
diode conduction
Gating Signal
VGS-Q3B
ZVS Turn-on of Q3
Figure 5.18
Experimental waveforms of current through resonant inductor LRA and LRB and
MOSFET Q3B voltage at Vin = 400 V and Vo = 300 V, Po = 300 W and fs = 100 kHz. Ch1=VDS-Q3B 200
V/div. Ch2= VGS-Q3 10 V/div. Ch3= Resonant inductor LrA current 5 A/div. Ch4= Resonant inductor LrB 5
A/div. Time scale=2 µs/div.
Figure 5.20 and 5.21 show the experimental results of voltage across rectifier diode DR2B and
current through transformer B secondary winding in DCM and BCM, respectively. The
voltage across the diode is clamped to the output voltage, at Vo = 300V, and the di/dt through
the secondary winding (which is same as the rectifier diode current), is low enough to
minimize any issues due to reverse recovery inherent with hyperfast diodes. The current
through the secondary winding reduces to 0 A naturally prior to turning-off, enabling the
diodes to turn-off with ZCS.
104
Resonant Inductor Resonant Inductor
LRB Current
LRA Current
ZCS Turn-off of Q3
Q3 anti-parallel
diode conduction
Drain-Source
Voltage VDS-Q3B
Gating Signal
VGS-Q3B
ZVS Turn-on of Q3
Figure 5.19
Experimental waveforms of Figure 5.18 repeated for Vin = 400 V and Vo = 300 V, Po = 3300
W and fs = 100 kHz. Ch1=VDS-Q3B 200 V/div. Ch2= VGS-Q3 10 V/div. Ch3= Resonant inductor LrA current
10 A/div. Ch4= Resonant inductor LrB 10 A/div. Time scale=2 µs/div.
Resonant Inductor
LRB Current
Rectifier
Diode
Voltage
DR2B
Transformer B
Sec. Current
Figure 5.20
ZCS Turn-off of DR2B
Experimental waveforms of current through resonant inductor LrB and transformer B
secondary winding and voltage across diode DR2B at Vin = 400 V and Vo = 300 V, Po = 300 W and fs = 100
kHz. Ch1= VDR2B 100 V/div. Ch3= Tx. B Sec. winding current 5 A/div. Ch4= L RB current 5 A/div. Time
scale=2 µs/div.
105
Resonant Inductor
LRB Current
Transformer B
Sec. Current
Rectifier Diode
Voltage DR2B
ZCS Turn-off of DR2B
Figure 5.21
Experimental waveforms of Figure 5.20 repeated for Vin = 400 V and Vo = 300 V, Po = 3300
W and fs = 100 kHz. Ch1= VDR2B 100 V/div. Ch3= Tx. B Sec. winding current 20 A/div. Ch4= LRB current
10 A/div. Time scale=2 µs/div.
5.5
96
94
92
Efficiency (%)
90
88
Vo = 300V Proposed interleaved converter with 4
diode voltage doubler rectifier and capacitive filter
86
84
82
Vo = 300V Benchmark interleaved converter with
inductive filter
80
Vo = 300V Benchmark Interleaved converter with 4
diode bridge rectifier and capacitive filter
78
0
Figure 5.22
500
1000
1500
2000
Output Power (W)
2500
3000
3500
input and 300V output voltage and benchmark converter
106
An efficiency comparison of the proposed converter with the benchmark interleaved ZVS
full-bridge DC-DC converter with capacitive output filter of Figure 4.1 and inductive output
filter of Figure 4.12 is provided in Figure 5.22. The benchmark converters were also operated
with trailing-edge PWM gating scheme.
The overall efficiency of the proposed converter is quite similar to the bench-mark converter
with capacitive output filter. The main advantage of the proposed converter with voltagedoubler rectifier is that it requires only half the number of secondary rectifier diodes as
compared to the benchmark converters.
much higher than the benchmark counterpart with inductive output filter. The benchmark
converter has lower efficiency due to losses in the secondary side RCD clamp circuit.
Thus the proposed interleaved dc-dc converter with voltage-doubler rectifier and capacitive
output filter stands out as the best choice in terms of physical size, weight, cost and
efficiency for power levels greater than 2 kW.
5.6
Conclusions
An interleaved, 2-cell, full-bridge dc-dc converter with voltage-doubler rectifier and
capacitive output filter operating with trailing-edge PWM gating has been presented in this
chapter. The proposed converter has been analyzed in BCM and DCM modes. A 2-cell, 3.3
kW dc-dc converter laboratory prototype was build based on the step-by-step procedure
presented in the chapter. It has been shown that both the cells share the total output power
equally, thereby equally sharing the power losses between the two cells. The main advantage
of the proposed converter is that, it requires only half the number of secondary rectifier
diodes as compared to the converter presented in the previous chapter.
107
much higher than the benchmark converter with inductive as well as capacitive output filter.
Thus this proposed converter with voltage-doubler rectifier stands out as the best choice in
terms of physical size, weight, cost and efficiency for power levels greater than 2 kW.
108
Chapter 6: Conclusion and Future Work
6.1
Introduction
With the emergence of PHEV’s and EV’s, one of the main concerns is the increased stress on
the existing utility grid infrastructure from the charging of high power battery packs. Another
concern for consumers is the increasing utility costs, making it extremely necessary for onboard battery chargers to operate with high efficiency. Other key requirements for chargers
include small size, low weight and low cost. As discussed in Chapter 1, the accepted power
architecture for a battery charger includes an ac-dc converter with power factor correction
(PFC) followed by an isolated dc-dc converter.
To meet the above mentioned requirements, four different isolated dc-dc converter topologies
have been proposed in this thesis. Thus this chapter summarizes four different contributions
of the thesis in section 6.2 and section 6.3 presents the scope of future work to be carried out.
6.2
Summary of Contributions
6.2.1
DC-DC Converter with Inductive Filter Operated with Trailing-Edge PWM
Gating
The first contribution is an isolated full-bridge dc-dc converter with inductive output filter
operated with trailing-edge PWM gating scheme for level-2 on-board battery charging
application. This converter was analyzed for all the operating intervals and based on the
analysis a 3.3 kW dc-dc converter prototype was also designed. The proposed converter
achieves a full-load efficiency of 96 % at an output of 400 V and 8.25 A and 94.9% at 300 V
and 11 A. It is shown that both output voltage and current are nearly free from low-frequency
(120 Hz) ripple. This is one of the important requirements for battery charging application. It
is also shown that all the primary side switches achieve ZVS resulting in lower losses, which
109
simplifies heatsink design. Another major advantage of this converter over traditional phaseshifted converter is that, it can achieve 0 % duty-cycle at lighter and no load conditions by
completely turning-off the PWM controlled switches. Some of the drawback like duty-cycle
loss, high voltage rectifier diode ringing and circulating current on the primary side of the
converter were also discussed.
6.2.2
DC-DC Converter with Capacitive Filter Operated with Trailing-Edge PWM
Gating
The second contribution is an isolated full-bridge dc-dc converter with capacitive output
filter operated with trailing-edge PWM gating scheme. This converter overcomes all the
issues inherent to converter with inductive filter such as duty-cycle loss, high voltage rectifier
diode ringing and circulating current on the primary side of the converter. This converter was
analyzed for all the operating modes and based on the analysis a 1.65 kW dc-dc converter
was also designed to operate in BCM mode at full-load. The proposed converter achieves a
peak efficiency of 95.7 % at an output of 400 V and 3 A and 94.9 % at 300 V and 5.5 A. It is
shown that both output voltage and current are nearly free from low frequency (120 Hz)
ripple. It is also shown that two primary side switches achieve ZVS at turn-on and ZCS at
turn-off and other two switches achieve ZCS turn-on resulting in lower losses. All the four
rectifier diodes achieve ZCS at turn-off enabling use of inexpensive hyperfast diodes since
reverse recovery is no longer an issue with this topology. As compared to the converter with
inductive filter, this converter doesn’t require lossy RCD voltage clamp circuit, since this
converter doesn’t suffer from high voltage ringing issue. The voltage across the rectifier
diode is naturally clamped to the output voltage. It is also shown that light-load efficiency of
this converter is significantly higher than its inductive filter counterpart. Finally, this
110
converter can also achieve 0 % duty-cycle at lighter and no-load conditions by completely
turning-off the PWM controlled switches.
6.2.3
Interleaved DC-DC Converter with Capacitive Filter Operated with Trailing-
Edge PWM Gating
As a third contribution a multi-cell, interleaved, isolated full-bridge dc-dc converter with
capacitive output filter operated with trailing-edge PWM gating scheme is presented for
level-2 (> 2 kW) where thermal management is a challenge with a single cell processing all
the power required for battery charging application. To illustrate this concept a 3.3 kW 2-cell
interleaved dc-dc converter with each cell operating at a maximum 1.65 kW of output power
was designed. It is shown that both the cells shared the total output power equally thus
illustrating power losses are also shared and distributed equally between the two cells, which
increases the reliability of the converter. It is also shown that by interleaving the ripple
frequency in the input and output filter capacitors are doubled which aids in reducing the size
of the filter components.
6.2.4
Interleaved DC-DC Converter with Capacitive Filter and Voltage Doubler
Rectifier
In the fourth contribution a multi-cell, interleaved, isolated full-bridge dc-dc converter with
capacitive output filter and voltage-doubler rectifier is presented. Voltage-doubler rectifier
configuration reduces the size and cost of the overall converter by using only four diodes for
2-cell configuration without impacting the thermal management concerns. To illustrate this
concept, a 3.3 kW 2-cell interleaved dc-dc converter with each cell operating at a maximum
1.65 kW of output power was designed. The proposed converter achieves a peak efficiency
of 96 % at an output of 400 V and 7 A and 95.2 % at 300 V and 11 A.
111
6.2.5
Comparison of Proposed Topologies
Table 6.1 summarizes the performance of all the proposed topologies based on the analysis
and experimental results presented in chapters 2 to 5.
Based on the comparison presented in the Table 6.1, the interleaved dc-dc converter with
voltage-doubler rectifier and capacitive output filter stands out as the best choice in terms of
physical size, weight, cost and efficiency for power levels greater than 2 kW. For power
levels under 2 kW, a single cell full-bridge dc-dc converter with capacitive filter would be a
preferred choice.
Table 6.1
Performance comparison of the proposed dc-dc converter topologies
Topology
Power Rating
EMI / Noise
Output
capacitor ripple
Magnetic Size
Efficiency
Power Density
(W/in3)
Cost (W/$)
Weight
Reliability
Full-bridge dcdc converter
with inductive
filter
< 3.3 kW
Poor
Full-bridge dcdc converter
with capacitive
filter
< 2 kW
Fair
Interleaved
Interleaved
full-bridge dc
full-bridge dc
converter with converter with
capacitive filter voltage doubler
> 2 kW
> 2 kW
Good
Good
Medium
High
Low
High
Large
Poor
Medium
Fair
Medium
Fair
Medium
Best
232
221
266
338
82
Heavy
Low
82
Medium
Medium
96
Medium
High
96
Light
Very High
112
6.3
Suggestions for Future Work
This sub-section outlines the possible future work for the thesis topics.
6.3.1
Full-Bridge DC-DC Converter with Clamp Diodes to Reduce Rectifier Ringing
Issues
In order to reduce the high voltage ringing issue in full-bridge dc-dc converter with inductive
output filter, clamp diodes could be implemented on the primary-side in between the external
resonant inductor Lr and transformer winding thus reducing voltage stress on the rectifier
diodes and reducing the size of the RCD clamp circuit. The circuit diagram of this
configuration is shown in Figure 6.1 below.
Lo
Clamp Diodes
Vin
Q1
Q2
Lr
DR1 DR2
Vo
Io
Dc
a
Co1
Q3
Figure 6.1
Co2
b
Q4
DR3 DR4
Rc
HV
Battery
Cc
Trailing-edge PWM Full-bridge dc-dc converter with inductive output filter and clamp
diodes
6.3.2
Full-bridge DC-DC Converter with Lossless Snubber
In order to reduce turn-off losses in the primary-side PWM controlled switches of the fullbridge dc-dc converter with capacitive output filter (presented in Chapter 3), some form of
active or passive lossless snubber could be implemented to increase the efficiency of the
converter.
113
6.3.3
Feedback Control Analysis for the Interleaved DC-DC Converter with
Capacitive Output Filter
A detailed feedback control analysis is required to understand the dynamics of the
interleaved dc-dc converter with capacitive output filter as shown in Figure 4.7 for battery
charging application.
114
Bibliography
[1]
M. Ehsani, Y. Gao, S. E. Gay, and A. Emadi, Modern electric, hybrid electric, and fuel
cell vehicles : fundamentals, theory, and design, CRC, 2005.
[2]
M. Shams-Zahraei and A.Z. Kouzani, “A study on plug-in hybrid electric vehicles,” in
Proceedings of TENCON IEEE Region 10 Conference, 2009, pp. 1 – 5.
[3]
M. Ferdowsi, “Plug-in Hybrid Vehicles - A Vision for the Future,” in Proceedings of
Vehicle Power and Propulsion Conference, 2007, pp. 457 – 462.
[4]
A. Emadi, K. Rajashekara, S. S. Willaimson, and S. M. Lukic, “Topological overview
of hybrid electric and fuel cell vehicular power systems architectures and
configurations,” IEEE Trans. Veh. Technol., vol. 54, no. 3, pp. 763–770, May 2005.
[5]
C. C. Chan, “The state of the art of electric, hybrid, and fuel cell vehicles,” Proc. IEEE,
vol. 95, no. 4, pp. 704–718, Apr. 2007.
[6]
C. C. Chan, A. Bouscayrol and K. Chen, “Electric, Hybrid, and Fuel-Cell Vehicles:
Architectures and Modeling,” IEEE Trans. Veh. Technol., vol. 59, no. 2, pp. 589-598,
Feb. 2010.
[7]
A. Khaligh and Z. Li, “Battery, ultracapacitor, fuel cell, and hybrid energy storage
systems for electric, hybrid electric, fuel cell, and plug-in hybrid electric vehicles: State
of the art,” IEEE Trans. Veh. Technol., vol. 59, no. 6, pp. 2806–2814, Jul. 2010.
[8]
SAE J1772, “SAE Electric Vehicle and Plug in Hybrid Electric Vehicle Conductive
Charge Coupler,” Rev. Feb. 2012, http://standards.sae.org/j1772_201202/
[9]
National Electrical Code, “Electric Vehicle Charging System,” Article 625.
[10] K. Morrow, D. Karner and J. Frankfort, “Plug-in Hybrid Electric Vehicle Charging
Infrastructure Review,” U.S. Department of Energy - Vehicle Technologies Program,
2008.
[11] J. Spangler, “Charging Standards for HEV and Plug-in EVs,” Presented in industry
session of Applied Power Electronics Exposition and Conference, 2012.
[12] M. M. Morcos, C. R. Mersman, G. G. Sugavanam, and N. G. Dillman, “Battery
chargers for electric vehicles,” IEEE Power engineering Review, pp. 8-11, November
2000.
115
[13] Y.J. Lee, A. Khaligh and A. Emadi, “Advanced integrated bidirectional AC-DC and
DC-DC converter for plug-in hybrid electric vehicles,” IEEE Trans. Veh. Technol., vol.
58, pp. 3970 - 3980, Oct. 2009.
[14] J. S. Kim, G. Y. Choe, H. M. Jung, B. K. Lee, Y. J. Cho, and K. B.Han, “Design and
implementation of a high-efficiency on- board battery charger for electric vehicles with
frequency control strategy,” in Proc. of IEEE Veh. Power Propulsion Conf., 2010, pp.
1–6.
[15] H. Bai, Y. Zhang, C. Semanson, C. Luo, C.C. Mi, “Modelling, design and optimisation
of a battery charger for plug-in hybrid electric vehicles,” IET Electrical Systems in
Transportation, vol. 1, no. 1, pp. 3 - 10, Mar. 2011.
[16] Shi Lisheng, M. Meintz, and M. Ferdowsi, “Single-phase bidirectional AC-DC
converters for plug-in hybrid electric vehicle applications” in Proceedings of Vehicle
Power and Propulsion Conference, 2008, pp. 1 - 5.
[17] B. Singh, B.N. Singh, A. Chandra, K. Al-Haddad, A. Pandey and D.P. Kothari, “A
review of single-phase improved power quality AC-DC converters,” IEEE
Transactions on Industrial Electronics, vol. 50, pp. 962 - 981 2003.
[18] F. Musavi, W. Eberle and W.G. Dunford, “Efficiency evaluation of single-phase
solutions for AC-DC PFC boost converters for plug-in-hybrid electric vehicle battery
chargers,” in Proceedings of IEEE Vehicle Power and Propulsion Conference, 2010,
pp. 1 – 6.
[19] F. Musavi, M. Edington, W. Eberle and W.G. Dunford, “Evaluation and Efficiency
Comparison of Front End AC-DC Plug-in Hybrid Charger Topologies”, IEEE
Transactions on Smart Grid, vol. 3, no. 1, pp. 413 - 421, Mar. 2012.
[20] UL 2202 - http://ulstandardsinfonet.ul.com/scopes/scopes.asp?fn=2202.html
[21] N. Mohan, T. M. Undeland and W. P. Robbins, "Power electronics converter,
application and design," 3rd ed., John Wiley & Sons, Inc., 2003.
[22] D.J. Shortt, W.T. Michael, R.L. Avert, and R.E. Palma, “A 600 W four stage phaseshifted parallel DC to DC converter,”, IEEE Power Electronics Specialists Conf., 1985,
pp. 136-143.
116
[23] V. Nguyen, J. Dhyanchand, and P. Thollot, “A multiphase topology series-resonant
DC-DC converter,” in Proceedings of Power Conversion International, 1985, pp. 4553.
[24] S. V. Dhople, A. Davoudi, A. D. Dominguez-Garcia, and P. L. Chapman, “A unified
approach to reliability assessment of multiphase DC–DC converters in photovoltaic
energy conversion systems,” IEEE Trans. on Power Electronics, vol. 27, no. 2, pp. 739
- 751, Feb. 2012
[25] Bong-Chul Kim, Ki-Bum Park and Gun-Woo Moon, “Analysis and design of twophase interleaved LLC resonant converter considering load sharing" Proceedings of
IEEE Energy Conversion Congress and Exposition, 2009.
[26] Jinsong Zhu and Annabelle Pratt, “Capacitor Ripple Current in an Interleaved PFC
Converter," IEEE Transactions on Power Electronics, vol. 24, pp. 1506 - 1514, 2009.
[27] D.S. Gautam, “Soft-Switched DC-to-DC Converters for Power Conditioning of
Electrolyser in a Renewable Energy System,” M.A.Sc. thesis, University of Victoria,
2006.
[28] R.L. Steigerwald, “High-Frequency Resonant Transistor DC-DC Converters,” IEEE
Trans. on Industrial Electronics, vol. 31, no. 2, May 1984, pp. 181-191.
[29] R.L. Steigerwald, “A Comparison of Half-Bridge Resonant Converter Topologies,”
IEEE Trans. on Power Electronics, vol. 3, no. 2, April 1988, pp. 174-182.
[30] J.A. Sabate, and F.C. Lee, “Off-line application of the fixed-frequency clamped-mode
series resonant converter,” IEEE Trans. on Power Electronics, vol. 1, no. 1, January
1991, pp. 39-47.
[31] F.S. Tsai, J. Sabate, and F.C. Lee, “Constant-frequency, zero-voltage-switched,
clamped-mode parallel-resonant converter,” IEEE International Telecommunications
Energy Conference, 1989, paper # 16.4, pp. 1-7.
[32] A.K.S. Bhat, “Fixed frequency PWM series-parallel resonant converter,” IEEE Industry
Applications Society Annual Meeting, vol. 1, October 1989, pp. 1115-1121.
[33] A.K.S. Bhat, "Analysis and design of a fixed-frequency LCL-type series-resonant
converter with capacitive output filter", IEE Proceedings: Circuits, Devices and
Systems, vol. 144, no. 2, April 1997, pp. 97-103.
117
[34] A.K.S. Bhat, "Analysis and design of LCL-type resonant converter", IEEE Trans. on
Industrial Electronics, vol. 41, no. 1, pp. 118-124, Feb. 1994.
[35] A.K.S. Bhat, "Analysis and design of a fixed frequency LCL-type series resonant
converter," IEEE Trans. on Aerospace and Electronic Systems, vol. 31, no. 1, Jan.
1995, 125-137.
[36] L. H. Mweene, C.A. Wright and M.F. Schlecht, "A 1 kW, 500 kHz front-end converter
for a distributed power supply system," IEEE Applied Power Electronics Conference,
1989, pp. 423-432.
[37] D.B. Dalal, “A 500 kHz multi-output converter with zero voltage switching”, IEEE
Applied Power Electronics Conf., 1990, pp. 265-274.
[38] J. A. Sabate, V. Vlatkovic, R.B. Ridley, F.C. Lee and B.H. Cho, "Design considerations
for high voltage, high power, full-bridge ZVS PWM converters," IEEE Applied Power
Electronics Conf., 1990, pp. 275-284.
[39] I.D. Jitaru, "A 3 kW soft switching DC-DC converter" presented at the IEEE Applied
Power Electronics Conference and Exposition, APEC, 2000.
[40] Chen Zhao, Xinke Wu, Wei Yao and Zhaoming Qian, "Synchronous Rectified SoftSwitched Phase-Shift Full-Bridge Converter with Primary Energy Storage Inductor"
presented at the IEEE Applied Power Electronics Conference and Exposition, APEC,
2008.
[41] Chen Zhao, Xinke Wu, Wei Yao, Zhaoming Qian, "Optimum Design Considerations
for Soft- Switched Phase-shift Full-bridge Converter with Primary-side Energy Storage
Inductor" presented at the IEEE Power Electronics Specialist Conference, PESC, 2008.
[42] Sriram Jala, “High Frequency Transformer Isolated Soft-Switched Hybrid Phase
Modulated DC-to-DC Converters,” M.A.Sc Thesis, Dept. of ECE, University of
Victoria, 2009.
[43] Redl, R.; Sokal, N. O.; Balogh, L.; “A Novel Soft-switching Full-bridge DC/DC
Converter: Analysis, Design consideration and Experimental results at 1.5 kW, 100
kHz.” IEEE Trans. on Power Electronics, vol. 6, no. 3, July 1991, pp. 408-418.
[44] Redl, R.; Edwards, D. W.; Balogh, L.; “Switch Transitions in the Soft-Switching FullBridge
PWM
Phase-Shift
DC/DC
Converter: Analysis
and
Improvements”
International Telecommunication Energy Conf., 1993, pp. 350-357.
118
[45] Redl, R.; Edwards, D. W.; Balogh, L.; “Optimum ZVS Full-Bridge DC/DC Converter
with PWM Phase-Shift Control: Analysis, Design Considerations, and Experimental
Result”, IEEE Applied Power Electronics Conf., 1994, pp. 159-165.
[46] M. Pahlevaninezhad, J. Drobnik, P.K. Jain and Alireza Bakhshai, “A load adaptive
control approach for a zero voltage switching DC/DC converter used for electric
vehicles,” IEEE Trans. on Industrial Electronics, vol. 59, no. 2, pp. 920-933, Feb.
2012.
[47] P.K. Jain, W. Kang, H. Soin, and Xi Youhao, “Analysis and design considerations of a
load and line independent zero voltage switching full bridge DC/DC converter
topology,” IEEE Trans. on Power Electronics, vol. 17, no. 5, pp. 649-657, Sep. 2002.
[48] G. Moschopoulos and P.K. Jain, “ZVS PWM Full-Bridge Converters With Dual
Auxiliary Circuits” in Proceedings of IEEE International Telecommunications Energy
Conference, 2000, pp. 574-581.
[49] R. Ayyanar and N. Mohan, “Novel Soft-Switching DC-to-DC Converter with full ZVSRange and Reduced Filter Requirement – Part I: Regulated-Output Applications” IEEE
Trans. on Power Electronics, vol. 16, no. 2, March 2001, pp. 184-192.
[50] Y. Jang and M.M. Jovanovic, “A new family of full-bridge ZVS converter” IEEE
Trans. on Power Electronics, vol. 19, no. 3, May 2004, pp. 701-708.
[51] G-B Koo, G-W Moon, and M-J Youn, "Analysis and design of phase shift full bridge
converter with series-connected two transformers," IEEE Trans. on Power Electronics,
vol. 12, no. 2, pp. 411 - 419, Mar. 2004.
[52] A.J. Mason and P.K. Jain “New Phase Shift Modulated ZVS Full-Bridge DC/DC
Converter with Minimized Auxiliary Current for Medium Power Fuel Cell
Application” IEEE Power Electronics Specialists Conf., 2005, pp. 244-249.
[53] A.J. Mason, D.J. Tschirhart and P.K. Jain, "New ZVS phase shift modulated full-bridge
converter topologies with adaptive energy storage for SOFC application," IEEE Trans.
on Power Electronics, vol. 23, no. 1, pp. 332-342, Jan. 2008.
[54] M. Borage, S. Tiwari, S. Bhardwaj, and S. Kotaiah, "A full-bridge DC–DC converter
with zero-voltage-switching over the entire conversion range," IEEE Trans. on Power
Electronics, vol. 23, no. 4, pp. 1743-1750, Jul. 2008.
119
[55] X. Wu, J. Zhang, X. Xie, and Z. Qian, "Analysis and optimal design considerations for
an improved full bridge ZVS DC-DC converter with high efficiency," IEEE Trans. on
Power Electronics, vol. 21, no. 5, pp. 1225-1234, Sept. 2006.
[56] X. Wu, X. Xie, J. Zhang, R. Zhao, and Z. Qian, "Soft switched full bridge DC–DC
converter with reduced circulating loss and filter requirement," IEEE Trans. on Power
Electronics, vol. 22, no. 5, pp. 1949-1955, Sept. 2007.
[57] Wu Chen, X. Ruan and R. Zhang, "A novel zero-voltage-switching PWM full bridge
converter," IEEE Trans. on Power Electronics, vol. 23, no. 2, pp. 793-801, Mar. 2008.
[58] Wu Chen, X. Ruan, Q. Chen, and J. Ge, "Zero-voltage-switching PWM full-bridge
converter employing auxiliary transformer to reset the clamping diode current," IEEE
Trans. on Power Electronics, vol. 25, no. 5, pp. 1149-1162, May 2010.
[59] Y. Jang, and M.M. Jovanovic, "A new PWM ZVS full-bridge converter," IEEE Trans.
on Power Electronics, vol. 22, no. 3, pp. 987 - 994, May 2007.
[60] I. Cho, K. Cho, J. Kim and G. Moon, “A new phase-shifted full-bridge converter with
maximum duty operation for server power system,” IEEE Transactions on Power
Electronics, vol. 26, no. 12, pp. 3491-3500, Dec. 2011.
[61] M. Ordonez, and J.E. Quaiceo, “Soft-switching techniques for efficiency gains in fullbridge fuel cell power conversion,” IEEE Transactions on Power Electronics, vol. 26,
no. 2, pp. 482-492, Feb. 2011.
[62] Y. Jang and Jovanovic, M.M., "Light-load efficiency optimization method," IEEE
Transactions on Power Electronics, vol. 25, no. 1, pp. 67-74, Jan. 2010.
[63] Bo-Yuan Chen, and Y-S. Lai, "Switching control technique of phase-shift-controlled
full-bridge converter to improve efficiency under light-load and standby conditions
without additional auxiliary components," IEEE Trans. on Power Electronics, vol. 25,
no. 4, pp. 1001 - 1012, Apr. 2010.
[64] A. K. S. Bhat, and F. Luo, “A new gating scheme controlled soft-switching DC-to-DC
bridge converter,” IEEE Power Electronics and Drive Systems Conf. Record, 2003, pp.
8-15.
[65] L. Hitchcock, "Full bridge power converter circuit," US Patent 4860189, 1989.
120
[66] F.S. Hamdad and A.K.S Bhat, “A novel pulse width control scheme for fixed-frequency
zero-voltage-switching DC-to-DC PWM bridge converter,” IEEE Trans. on Industrial
Electronics, vol. 48, no. 1, pp. 101-110, Feb. 2001.
Automotive On-Board 3.3 kW Battery Charger for PHEV application," Proceedings of
IEEE Vehicular Power and Propulsion Conference (VPPC 2011), Chicago, pp. 1-6,
Sep. 2011.
Zero Voltage Switching Full-Bridge DC-DC Converter for an On-Board PHEV Battery
Charger," Proceedings of IEEE Transportation Electrification Conference and Expo
(ITEC 2012), Dearborn, pp. 1-6, Jun. 2012
Automotive On-Board 3.3 kW Battery Charger for PHEV application," IEEE
Transactions on Vehicular Technology, vol. 61, no. 8, pp. 3466-3474, Oct. 2012.
[70] D.S. Gautam, W. Eberle and W.G. Dunford, "A Zero Voltage Switching Full-bridge
DC-DC Converter with Capacitive Output Filter for a Plug-in-Hybrid Electric Vehicle
Battery Charger," Proceedings of IEEE Applied Power Electronics Conference and
Exposition (APEC 2012), Orlando, pp. 1381-1386, 2012.
for a Plug-in-Hybrid Electric Vehicle Battery Charger," IEEE Transactions on Power
Electronics, vol. 28, no. 12, pp. 5728-5735, Dec. 2013
Interleaved Zero Voltage Switching Full-Bridge DC-DC Converter with Capacitive
Output Filter for a Plug-in-Hybrid Electric Vehicle Battery Charger," Proceedings of
IEEE Energy Conversion Congress and Exposition (ECCE 2012), Raleigh, pp. 28272832, September 2012.
[73] D.S. Gautam, Fariborz Musavi, Murray Edington, W. Eberle and W.G. Dunford, " An
Isolated Interleaved DC-DC Converter with Voltage Doubler Rectifier for PHEV
Battery Charger, "Proceedings of IEEE Applied Power Electronics Conference and
Exposition (APEC 2013), Long Beach, pp. 3067-3072, Mar. 2013.
121
[74] Song-Yi Lin and Chern-Lin Chen, “Analysis and design for RCD clamped snubber
used in output rectifier of phase-shift full-bridge ZVS converters,” IEEE Trans. on
Industrial Electronics, vol. 45, no. 2, pp. 358-359, Apr. 1998.
122

INVESTIGATION OF HIGH-PERFORMANCE DC-DC CONVERTERS FOR PLUG-

Transcription

Similar documents

TopConvert - Komax Group

WF-8900REP Series Brochure

illustrated guide to upgrading y magnetek/parallax 6300 series pow

RS SERIES REDUNDANT SWITCHOVER UNITS Electronic Products

Bidirectional Power Train DC

View - IJEI

AN HF TO VHF RECEIVE CONVERTER By G0CWA DEC. 2012

Root Cause Analysis (RCA): Receipe for disaster

Rotary Phase Converter Data Sheet