FOCUS ON TUBES

Transcription

FOCUS ON TUBES

CAPACITOR REPORT: A VISIT TO MUNDORF
M AY
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Tube, Solid State,
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sound solutions
By Walt Jung
Sources 101: Audio Current Regulator
Tests for High Performance
Part 2: Precise High Current/Voltage Operation
Measurement tests can help reveal which configuration is
best for your power supply application.
I
will conduct many additional measurements here. Within this phase,
the focus is on current regulators
that operate at higher voltages, at
higher currents, and do so with a higher
degree of precision. This implies higher
initial accuracy, as well as good temperature stability, for all circuits discussed
hereafter, with the exception of those
MOSFET based.
LM317 CURRENT SOURCE/SINK
One of the easiest ways to make a quite
good audio current source is to simply
connect an LM317 IC with a current
set resistor (Fig. 10A, left). This circuit,
which is simplicity personified, cannot
be reduced further in functionality. Details of the LM317 operation are described in References 7 and 8 (highly
recommended reading). The wide avail-
ability of this useful part in a variety of
packages at low cost makes it attractive.
The LM317 is a floating three-terminal regulator, meaning it can be applied
quite flexibly, and no pin inherently
needs to be grounded. When operated
in a current mode, the internal 1.25V
reference voltage appears between the
OUT and ADJ pins, so a simple resistor
Rset programs the current into a load.
In this case a fixed 20Ω value sets up a
62mA load current. The 1.25V is held to
±50mV, and is stable over temperature.
Thus, an LM317-based current source
will be one of the more predictable and
stable types for DC. Of course, at such
higher currents power dissipation will
be an issue, so you should use a TO220 package part at these current levels,
along with the appropriate heatsink.
It may not be obvious at first, but the
FIGURE 10A: Basic LM317 current source (left) and sink (right).
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JungPart2-2779-2.indd 8
LM317 can function as both a current
source (as in the left case) and as a current sink, shown at the right. In either
case, the IC and its Rset resistor are
treated as a two-terminal circuit, which
is applied between the source and the
load. The LM317 current sink is implemented with similar connections shown
at the right, with the load connected to
the IC’s IN pin, and using a negative
power supply. Note that in such cases a
small AC bypass capacitor may be necessary at this pin, ~1µF.
The LM317 working in this current
output mode will require about 2.5V
across the IC, plus the 1.25V, for a total
of nearly 4V to make it operate. The IC
also needs a 10mA minimum of output
current for regulation. Practically speaking, this means that Rset should never
be any higher than about 125Ω.
FIGURE 10B: Performance of the LM317 as a 62mA current
source shows 110dB or more rejection below 10kHz, but rapid
deterioration at higher frequencies.
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3/26/2007 9:59:43 AM
LM337
CURRENT SOURCE
FIGURE 11A:
Basic LM337
current source.
Once biased properly, the IC operates
reasonably well, as shown in Fig. 10B,
an AC rejection performance plot of the
output. Here the low frequency (LF)
rejection is about 110dB, equivalent to
an impedance of 316kΩ. There is, however, noticeable deterioration at higher
frequencies.
This is one aspect of the LM317’s
performance that would be desirable
to improve, because the rejection at
200kHz is only about 60dB, meaning
potentially increased sensitivity to high
frequency (HF) intermodulation. A couple of the following circuits address this
aspect of the LM317’s operation.
formance (Fig. 11B). While the LM337
rejection is good below a few hundred
Hz, it degrades steadily above this, to
the point where the rejection is less than
30dB above 100kHz. This is an example
of the type of rejection not sought for
higher performance audio circuits!
A detail worth noting at this point: If
complementary source and sink circuits
are needed for an application, it is actu-
A companion device to the LM317
positive regulator IC is the LM337, designed to operate from negative sources.
It also has a 1.25V reference voltage and
can be configured to regulate current
(Fig. 11A). The LM337 uses a similar set
resistor (Rset) to set up an output current Iout, but it also requires an output
capacitor for frequency compensation, C1. A typical
value for this capacitor is shown.
While the
LM317
and
LM337 have
complementar y
functionality, they
achieve radically
different degrees
of rejection versus
frequency as operated in a current
mode. This is best
FIGURE 11B: Performance of the LM337 as a 62mA current source
appreciated by the
shows 110dB or more rejection—but only at the lowest frequencies.
LM337’s AC per-
audioXpress May 2007
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3/22/2007 4:17:28 PM
ally better performance-wise to use a
pair of LM317s as in Fig. 10A left and
right, than it would be to use an LM317
and an LM337.
Caveats: A further special point on threeterminal regulator types is to simply be cautious about replacement or “improved” 317type regulators, especially those designed for
low dropout. As a byproduct of their design for low DC dropout voltage, these
regulator types can have much worse
AC rejection characteristics vis-à-vis the
original. For example, two low dropout
versions of the 317 were tested for rejection in a current regulator mode similar
to Fig. 10A, and had responses more like
that of Fig. 11B than the more desirable
LM317 response of Fig. 10B. So, this is
definitely a case of caveat emptor!
DEPLETION MODE MOSFET
CURRENT SOURCE/SINKS
uum-tube-based audio projects where
high voltage capability is required. Examples can be found via Reference 12.
In application, a basic current source
using either part can be accomplished
( Fig. 12A ). This circuit is exactly the
same as with a JFET device, save the
addition of the gate-stopper resistor R1,
and the important fact that the applied
voltage can go up to 450V. And, like the
JFET counterpart current regulator, this
circuit is two-terminal, and so can be
used either as a source (shown here), or
as a sink, where the load is in the drain
lead and negative voltage is applied to
the bottom of Rset and R1. The tests described here used an 18V power supply.
For a load current of 30mA, I found
that the two resistor values noted for
Rset were appropriate. This underscores
a basic point: These depletion mode
MOSFETs aren’t precision devices like
the LM317 and other ICs with their
fixed reference voltage(s). Rather, the
gate bias for these MOSFETs sample to
sample will vary, just as it does for other
JFET and MOSFET parts. Nevertheless, this circuit still has the utility of
extreme simplicity, and Rset is simply
chosen to get the required current.
Operated within the test circuit of Fig.
12A, the two sample parts produced the
data of Fig. 12B. Both devices show LF
rejections of around 110dB (~316kΩ),
with a gradual degradation beginning
in the 5–10kHz range. The DN2540 is
measurably better in terms of AC rejection at the higher frequencies. This is
apparently due to the lower parasitic
Power MOSFETs are both extremely
popular and widely available, and for
many years have seen widespread use in
audio amplifiers. Typically, these have
been the original format, which is that
of enhancement mode devices. This means
simply that they require an applied gate
voltage to conduct.
More recently, depletion mode MOSFETs have become available, which enables easier use of such parts in audio
power supplies. Like the small signal
JFETs, a depletion mode MOSFET is
fully on with 0V bias, and is controlled to
lower degrees of conduction with the applied bias voltage. Thus far the depletion
mode MOSFETs that have appeared
are N-channel
parts. Two TO-220
packaged examples
are the DN2540
from Supertex and
the IXCP 10M45
from Ixys. See References 10 and 11
for further information.
These TO-220
devices can operate at voltages up
to 450V, and at
currents from the
low mA range up
to about 100mA.
FIGURE 12A: Basic depletion mode
They are already
MOSFET current source.
being found in vac10
audioXpress 5/07
capacitance of the DN2540 versus the
IXCP 10M45, but I cannot precisely
confirm this (the latter isn’t specified for
capacitance).
Nevertheless, these general patterns
of AC rejection seemed to be typical for
the two devices, and were observed with
tests of other samples. The DN2540 is
preferred for operation in this circuit,
not only because of the better AC rejection at high frequencies, but because the
Idss of this part is 150mA, making it
more widely applicable.
CASCODE LM317
CURRENT SOURCES
These higher current regulators, like the
low-level circuits described in Part 1, can
also be enhanced for AC performance
by means of cascoding. As the DC current carried by the regulator is increased,
the rejection performance inevitably degrades, making the value of an effective
cascode circuit more and more important toward good results.
A circuit that can be used to cascode
the operation of an LM317 is shown
in Fig. 13A. This is similar to the basic
regulator of Fig. 10A , with an additional regulator added—stage U2. The
U1 LM317 operates just as previously,
producing an output current as noted,
which is proportional to 1.25V and inversely proportional to Rset. The input
drive for U1 is derived from cascode IC
U2, which floats atop U1’s output, 2.5V
higher by virtue of resistors R1 and R2.
C1 and R3 provide necessary stabilization for the cascode.
FIGURE 12B: Performance of two depletion mode MOSFET
30mA current sources shows ~110dB rejection below 5-10kHz,
then deterioration as frequency increases.
3/22/2007 4:17:29 PM
I tested the Fig. 13A circuit at a current level of 62mA, to be consistent with
the basic LM317 operation of Fig. 10A.
The results are shown in Fig. 13B for
both the basic and cascode modes of
operation. Note that the addition of the
cascode reduces the noise down to a
level approaching the setup residual at
all but the very highest frequencies. Although not shown here, for lower levels of current operation (i.e., ~15mA),
FIGURE 13A:
Cascode LM317
current source.
this cascode scheme showed even lower
noise levels.
Some caveats for the Fig. 13A circuit:
Although the AC rejection properties
of this relatively simple circuit could be
considered exemplary in some regards,
I cannot recommend it unconditionally
for several important reasons. One, it has
a rather high dropout voltage, requiring
~6.5V across it—just to function! This
is due primarily to the basic characteris-
tics of the LM317, and can’t be easily reduced. Anticipating potential questions
here, using low dropout 317 regulators
isn’t any real help, either. I tried this, and
it does reduce the dropout—but at the
expense of rejection.
A second caveat is that the basic
LM317 dropout voltage is actually
specified as 3V for currents up to 1.5A.
Datasheet graphs show it to be typically
~1.7V at a current of 200mA at 25°C. So
FIGURE 13B: Performance of the LM317/LM317 as a cascode
62mA current source shows much greater rejection than in
basic mode, at all frequencies.
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3/22/2007 4:17:31 PM
the scheme here won’t really work well at
high currents and/or low temperatures.
But, there is still much latitude for
use at much lower currents and typical
temperatures from 25° C and up. Here
operation of U1 is at a fixed input/output voltage of 2.5V, and because this is
still somewhat of a gray area, only load
currents of <100mA are suggested. Finally, and perhaps most important, this
setup can and will oscillate under certain conditions, so be wary. All cascodetype schemes using additional high gain,
wide bandwidth parts have this potential
and should be rigorously checked. Input
bypassing should be used, with a film
capacitor such as C2 close to U2, and
the C1/R3 network always used.
Fortunately, for all of the cascode
schemes tested for this series, only a
couple of them showed oscillation tendencies, this one included. The absence
of oscillation for an LM317 regulator
can be checked by the presence of a stable 1.25V ±50mV output (or, the exact
target DC current for this or other precision regulators). If a scope is used, the
output should be clean on a scale of a
few mV.
For some more carefully selected operating conditions, a cascode LM317
arrangement can be implemented using
an LM317 as the control IC and a depletion mode MOSFET as the cascode
part. This variation (Fig. 13C) can use
either the DN2540 or the 10M45 as the
cascode device M1. Note that this circuit
FIGURE 13C:
Cascode LM317 +
MOSFET current
source.
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will simply not work with a conventional
MOSFET! For the two M1 device types,
it has the advantage of workability at
very high voltages, up to 450V, making
it quite attractive as a simple and precise
current source for tube circuits.
This circuit also has some caveats,
including the general ones for the 317.
For the LM317 to properly function as
a regulator, the input/output voltage, labeled here as V317, must meet the LM317
device dropout limits. In this circuit V317
is the Vgs of M1, and this should be
2.5V or more. Both the devices listed
for M1 typically meet this requirement
at lower currents of 10–20mA, and the
DN2540 holds up even higher. And,
don’t forget the RC stabilization network, R2/C1.
AC rejection performance of this circuit operating at 16mA is shown in Fig.
13D, and for either of the cascode devices it is nearly ideal. Only a tiny deviation
above the noise level at the very highest
frequencies can be noted. This exceptional performance makes this a very attractive circuit for such lower currents.
At the higher current of 38mA (Fig.
13E), the 10M45 begins to approach the
sample device Idss. Therefore, V317 is
lower than the minimum required for effective LM317 operation, and as a result,
the data for the 10M45 shows noticeable
deterioration vis-à-vis lower currents. By
contrast, the DN2540, a higher current
device, still shows excellent rejection for
these conditions.
A power caveat: While you should always be aware of power dissipation limits for any of these circuits, this boundary can quickly sneak up on you within
tube circuits—even at relatively low current levels. For example, a 10mA current
in M1 of Fig. 13C with 150V across
it implies an M1 dissipation of 1.5W,
which will definitely require a heatsink.
Don’t operate under the assumption that
a datasheet rating of 1W at 25° C for a
TO-220 will guarantee a safe and long
life of the part, if it sees 1W of constant
power while the room is 25° C. Internally, the part will be much hotter, and it is
highly likely a hefty heatsink is in order
for a truly reliable design. See Reference
13 for further heatsink information.
TLV431 CURRENT SINK
The TLV431 is a three-terminal IC designed to be used as a programmable
shunt regulator, from 1.24 to 6V14. It has
an uncommitted feedback path, meaning that external active parts can be used
with it to extend the basic current and
voltage range. As you will see, this part
operates as a current regulator referred to
the negative rail, thus it is most suited to
make current sinks.
The TLV431 reference voltage of
1.24V has a tolerance of ±18mV (1.5%),
but A and B suffix parts tighten this to
12mV (1%) and 6mV (0.5%), respectively. The TLV431 is related to the very
popular TL431, which offers similar
functionality at a reference voltage of
FIGURE 13D: Performance of the LM317 + MOSFET cascode 16mA current source shows excellent rejection compared to basic mode at all frequencies.
3/22/2007 4:17:32 PM
2.5V. Because the TLV431’s lower voltage of 1.24V is more desirable for a current regulator (it means lower dropout),
I chose it for this test. But note that
the same principles applied here for the
TLV431 also work for the TL431, except
for the higher reference voltage of 2.5V.
Figure 14A is a basic TLV431 current sink that you can use over a range
of voltages up to 40V, and currents up
to several tens of mA. The final voltage/
current rating for this circuit is a function of the transistor type used for Q1
and the heatsinking. Typically the load
would be applied between the OUT1
and OUT2 terminals. Note that the
OUT1 terminal need not be common
to the +18V supply as shown; it can (and
often will) be a higher voltage.
The U1 IC, a TLV431, regulates with
a 1.24V developed between the R and A
terminals as noted, so the Rset resistance
determines the current flowing into Q1Q2 and the external load. The feedback
path is via terminal K and the baseemitter path of Q1–Q2. Z1 performs as
load impedance for IC U1, and can be
one of three options, all of which should
provide for a current of 100µA, mini- serving the total current in Rload1.
mum. The simplest option is a 100kΩ
The current in Rload1 has two comresistor (A); next most simple a current ponents, the output current flowing in
source such as the J507 (B); and finally, Rset-Q1/Q2, and Iz1, the bias current
for highest performance from the cir- of U1, which flows in Z1-U1. When the
cuit as a whole, functioning as a current current in Rload1 is monitored, both of
source, a J202 operating at ~280µA and these currents are, in fact, being meacascoded with a 2N5486 (similar to Fig. sured. It would be desirable that only Iz1
8A, right option, Part one).
be dominant, because this would mean
Because this circuit is more aptly
used as a current
sink, the measure
of how it performs would best
be told by a sense
resistor placed at
OUT1-OUT2.
But, as noted, the
test setup here
measures current
in Rload1, which
is tied to ground.
Interestingly, however, you can still
FIGURE 13E: Performance of the LM317 + MOSFET cascode
infer some degree
38mA current source still shows excellent rejection for the
of performance of
DN2540, but deterioration for the 10M45S.
the circuit by ob-
13
3/22/2007 4:17:35 PM
that the Rset-Q1/Q2 current path is
noise free.
To a great extent this is indeed true,
and is reflected by a related change in
Iout rejection, because Z1 is varied. This
is shown in Fig. 14B for various Z1 conditions. Note that for a finite resistance
value for Z1, the net impedance is shown
by the Vout (100k) plot (as was true
for the calibration plots of Part one of
this article). And, as Z1 takes on higher
impedance characteristics, such as with
the Vout ( J507) plot, this condition is
reflected in a higher impedance display
(i.e., more rejection). The greatest rejection at the lower frequencies is provided
by the cascoded J202 setup, while the
J507 provides the most rejection with a
single component used for Z1.
So, while this test method doesn’t directly measure just the current flowing
in the collector of Q1/Q2, it still suggests some aspects of relative quality—a
good thing, nevertheless. The bottom
line is that you can use the circuit as either a current sink, in which case Z1 can
likely be the simple 100kΩ resistor, or,
alternately, as a current source, whereby
the higher impedance choices for Z1
are suggested, such as the J507 or the
cascoded J202.
You might ask what the need is for
this type of current sink, when previous
examples have provided good performance at these currents. The answer
lies in the overall flexibility of Fig. 14A.
FIGURE 14A:
TLV431 current
sink (source).
14
audioXpress 5/07
Operated as a current sink, and with Q1
properly selected for power and voltage
handling, this circuit can handle currents of amperes and voltages as high
as the Q1 device rating. Although data
isn’t shown for this example, with a D44
series power transistor for Q1, output
currents of 350mA have been witnessed.
This is all available with relative simplicity—Z1 a 100k resistor (Z1) and Rset
chosen for the current desired. Or, with
Q1/Q2 2SC2362XS, the OUT1 terminal can operate up to 150V, at low currents, with proper heatsinking.
LM4041 CURRENT SOURCE
The LM4041-ADJ is a three-terminal
IC designed to be used as a programmable shunt regulator, from 1.233 to 10V15.
Like the counterpart TLV431 series, it
also has an uncommitted feedback path.
And, as with the TLV431, this means
external active parts can be used with it
to extend the basic current and voltage
range.
A key difference in applicability is
that the LM4041-ADJ operates with
a positive rail common, as opposed to
the TLV431, which uses a negative rail
common scheme. The two devices can be
viewed as complements, performing similar tasks. The basic LM4041-ADJ reference voltage is 1.233V, and the available
grades of C and D for this version have
initial tolerances of ±0.5% and ±1%, respectively, for Vout = 5V.
Inasmuch as the operation of the
LM4041-ADJ is with the positive rail
common, you can easily use it to make
current sources operating over a wide
range. An example is shown in Fig. 15A,
which is a mirror image of the TLV431
circuit of Fig. 13A.
In this current source circuit, the output current is measured in Rload1, which
is in series with the Q1-Q2 collectors.
There is no error current from the internal amp of the LM4041, thus the rejection characteristics measured at Rload1
are indeed what you get. This is shown
in Fig. 15B, for conditions as shown and
a current of 38mA. The LF rejection is
approaching 130dB (3.16MΩ), which,
while good, is still well above the noise
level. However, the rejection deteriorates
above 1kHz.
Cascoding of Q1-Q2 in this circuit
did not improve the performance to any
great degree, only 2-3dB. At lower current levels of a few mA, the rejection
improved to just above the residual noise
level. From this, you would conclude
that this particular circuit is better used
at the lower current levels.
TLV431 BOOSTED CURRENT
SOURCE/SINK
As noted, the TLV431 circuits are better
suited to use as current sinks, as opposed
to sources. But, with some key changes,
you can use a TLV431 current regulator either as a source or sink, and/or at
FIGURE 14B: Performance of the TLV431 as a 38mA current
source depends upon Z1, but is excellent with a high-Z for Z1.
See text on current sink operation.
3/22/2007 4:17:37 PM
high voltages. One scheme to do this is
shown in Fig. 16A.
This circuit is like Fig. 14A, except Q1
uses a standard connection (non-Darlington), and the current source portion
represented by Z1 of Fig. 14A is replaced
by a high current or high voltage equivalent. This has the effect of regulating the
current in R1, making the error current
flowing from Rset and the TLV431-A
pin constant. Therefore, this circuit, operating as a whole, can be used either as
a source or as a sink.
With an LM317 for U2, R1 establishes a current of ~800µA, providing drive
to Q1 for currents of 50mA or more.
Q1 is bootstrapped by the LM317 at
the collector and sees less than 2V C-E.
It thus does not dissipate high power at
38mA of output or even higher currents.
The LM317 will need the heatsink in
this circuit long before Q1!
For operating voltages higher than the
40V LM317 rating, you can also use a
FIGURE 15B: Performance of the LM4041 as a 38mA current source is good, but falls short of excellent, particularly
at the higher frequencies.
FIGURE 15A:
LM4041 current
source.
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depletion mode MOSFET by substituting an M1 device at the points marked X
and Y. R1 can remain the same, and the
current limit for this mode will of necessity be much less than 40mA. But, the
voltage limitation becomes that of the
M1 device used, or 450V as shown. Take
care to use a proper heatsink for M1!
Performance in terms of AC rejection
is shown in Fig. 16B for all three cascoding options, operating at 38mA. Overall,
the best performance is achieved with
the LM317, where the errors are only
slightly more than residual noise, except
for the very highest frequencies. The two
MOSFET parts are nearly as good at LF,
but deteriorate more rapidly above 1kHz.
Of the two MOSFETs, the DN2540 is
favored due to lower noise at all frequencies, plus its ability to handle more current. To get higher output currents, Q1
can be operated as parallel devices driven
from R1-bottom end, with 10Ω current
sharing resistors in the emitters.
CONCLUSIONS, CAVEATS, AND
RECOMMENDATIONS
This concludes the testing portions of
this series. A future article will explore
some example applications of current
sources and sinks within audio circuits,
and discuss some general power supply
system noise reduction techniques.
Some general caveats are appropriate
here, beyond those specifically stated. I
believe the tests are valid for the conditions cited, and in general can be used to
differentiate among the various circuits.
Of course, there is an infinite set of different load, voltage, and current operating conditions that you may require. So,
you should not expect to duplicate any
measurements exactly for other conditions. But, in general the observations
should hold up—cascodes work better, JFETs need proper voltages to work
best, and so on.
To summarize, here are some principles to keep in mind:
• Select single JFET parts from families with lowest Vgs and thus highest
rejection. An example would be the
J201/2 series.
• Alternately, select from a specified
JFET current regulator device family,
such as the J507 series.
• Always operate current regulator circuits with sufficient voltage headroom
to maximize rejection.
• Above 4-5mA of current, consider cascode type circuits. At several tens of
mA, this should be considered mandatory for good performance.
• For any current regulator circuit, minimize
capacitance in whatever active devices are
used. This will enhance high frequency
noise rejection and minimize the possibility of high frequency intermod.
If I were asked to recommend which
of the many current regulators described
here to use, I’d try to keep it as simple as
possible. The maximum bang-for-the-
FIGURE 16A:
Boosted TLV431
current
source/sink.
16
audioXpress 5/07
buck is the cascode LM317 + MOSFET of Fig. 13C, assuming your current
requirement is 40mA or less. This one
worked great for me within a 12mA
current feed for a 24V shunt regulator.
For higher currents, the Fig. 16A circuit
is both flexible as a source or sink and
capable of much higher currents when
Q1 is appropriately selected. For low
currents of just a few mA, single and/or
cascoded JFETs are likely best (Fig. 8A).
Or, you could select the reference diode
circuit of Fig. 6A.
SOME HOMEWORK ASSIGNMENTS
One manuscript reviewer asked about
very high output currents, i.e., several
amps. My general answer is that yes, this
should be possible with minor revisions.
I said, “You could use Fig. 14A with
a conventional N-channel MOSFET
replacing Q1/Q2. The 1-2V Vgs of a
MOSFET will bias the K pin of U1
roughly 1-2V above the R pin (but don’t
forget a 100Ω snubber in the MOSFET
gate circuit). This should work OK for
ampere outputs. Pick the FET for the
required current, voltage, power, and,
preferably, lowest C. I’m sure you have
a favorite here. One possibility might
be the Fairchild FQP4N20L, a TO-220
part, available from Mouser. I think I’ll
put this idea in at the end of the Part 2,
as a reader ‘Homework’ assignment.”
So there you have one assignment
for some fun experiments. Let us know
what you find out with this MOSFET-
FIGURE 16B: Performance of the boosted TL431 as a 38mA
current source or sink ranges from good to excellent, dependent upon the cascode device chosen.
3/22/2007 4:17:41 PM
boosted current source idea!
Another assignment is to explore a
hybrid vacuum tube/solid-state current
regulator. For example, you could also
use Fig. 13C with a power triode in place
of M1 (grid to U1-OUT, cathode to
U1-IN, and plate to the input voltage).
The LM317L might be a possible candidate for U1.
I’d be very interested to hear about
your results with these ideas. Write me
at audioXpress via conventional mail, or
contact me via my website, www.waltjung.
org/index, and happy current sourcing
and sinking! aX
Acknowledgments
A number of folks reviewed the
Sources 101 manuscript and made helpful comments toward improvement:
Ken Berg, Erno Borbely, John Curl, Bob
Fitzgerald, Clarke Greene, Chuck Hansen, Bruce Hofer, Mark Kovach, Rick
Miller, and Andy Weekes. My sincere
thanks go out to all of them.
REFERENCES
1. Walt Jung, “Regulators for High Performance Audio, parts 1 and 2,” The Audio
Amateur, issues 1 and 2, 1995.
2. “LM134/234/334 3-Terminal Adjustable Current Sources,” National Semiconductor, March 2005, www.national.com.
3. Arthur D. Evans, Designing With FieldEffect Transistors, McGraw-Hill, ISBN 0-07057449-9, 1981.
4. “The FET Constant-Current Source/
Limiter,” Application Note AN103, Vishay/
Siliconix, March 10, 1997, www.vishay.com.
5. “J500 Series Current Regulator Diodes,”
Vishay/Siliconix, July 2, 2001, www.vishay.com
6. Selected and matched JFETs and
JFET current regulator devices, as well as
other audio components, are available from
Borbely Audio. See www.borbelyaudio.com/
audiophile_components.asp.
7. Bob Dobkin, “3-Terminal Regulator is
Adjustable,” National Semiconductor Application Note 181, October 1975, www.national.
com.
8. Bob Dobkin, “Applications for an
Adjustable IC Power Regulator,” National
Semiconductor Application Note 178, January 1977, www.national.com.
9. “LM337 – 3-Terminal Adjustable Negative Regulator,” National Semiconductor,
www.national.com.
10. “DN2540 N-Channel DepletionMode Vertical DMOS FETs,” Supertex,
www.supertex.com.
11. “IXCP 10M45 Switchable Current
Regulators,” Ixys, www.ixys.com.
12. John Broskie of www.tubecad.com has
written much on current regulators, both
solid-state and tube related. To reveal this
fascinating information, do a Google search
of his site using these terms: “current source
site:www.tubecad.com” (less the quotes).
13. Walt Jung, “Thermal Considerations,” Section 7-5 within Walt Jung, Ed.,
Op Amp Applications, ISBN 0-916550-26-5,
2002, www.analog.com/library/analogDialog/
archives/39-05/op_amp_applications_handbook.
html.
14. “TLV431, TLV431A, TLV431B LowVoltage Adjustable Precision Shunt Regulator,” Texas Instruments, January 2006, www.
ti.com.
15. “LM4041 Precision Micropower
Shunt Voltage Reference,” March 2005, National Semiconductor, www.national.com.
17
3/22/2007 4:17:47 PM
special report
By Jan Didden
We Visit Mundorf
The Mundorf Company is perhaps best known for its high-performance
capacitors and coils for audio applications. The author visited the company
and found that it is active in many more audio fields and is hard at work
on new products for audiophiles.
PHOTO 1: The Mundorf
brothers and the author
(right) discussing crossover issues over coffee and
Kuchen. (All photos by Lou
Jansen).
M
undorf is located in Cologne,
Germany, just an hour’s drive
from my hometown. I was
cordially greeted by Raimund
Mundorf, the founder of the company,
and his brother Norbert ( Photo 1) . Over
a coffee with typical German Kuchen,
we talked about audio and the history
of the company, which started in 1985
when Raimund eventually built a coil
winding machine for the custom filter
coils he handcrafted one at a time. The
machine worked quite well and gradually
a business opportunity presented itself.
The old machine no longer exists but has
been followed by new models producing
new and better products.
COMPANY BELIEFS
The goal of Mundorf is to design “ideal”
components. For example, it is known
that generally capacitors have not only
a capacitance, but also an ohmic series
resistance and a series inductance. Those
parasitic attributes have unwanted effects
on performance and can cause frequency
response deviations and even oscillations
in certain applications. The dielectric
that is used as isolation in capacitors is
also important: Less ideal dielectrics are
known to cause absorption and subsequent release of signal energy that can
distort the sound.
But there is no free lunch, of course.
For example, Mundorf foil caps are often
PHOTO 2:
Large contact
area of the
MCap RXF.
18
audioXpress 5/07
Didden2788-1.indd 18
PHOTO 3: Raimund Mundorf giving away (almost) the secret of ultra-low inductance caps.
shorter, but larger in diameter, than other
caps of equivalent rating. Why? Well,
the usual wide, small diameter caps have
higher inductance because of the longer
pathways of the signal from one connection to the other. On short, large diameter caps, not only is the path shorter, but
also there are more windings that appear
in parallel, further reducing unwanted inductance. The ESR is also lower because
of the wide contact area. You can see this
easily in a type MCap RXF cap (Photo
2).
However, short, large diameter caps
have many more windings, which means
more expensive production. Mundorf believes that the extra cost is worth it for
better performance, and its customers
seem to agree! The latest trick is using
a series connection of two internal caps
with reversed internal current flow. That
means that to get, say 2µF, you need to
build two caps of 4µF in one package,
which is four times the material and effort otherwise required. The result, however, is a cap with almost zero inductance:
the Supreme series, also available in silver/oil and silver/gold. These are exotic,
expensive materials, but they do improve
the characteristics of the components.
Over the years, Mundorf crossover
components have earned their place
among discerning audiophiles and manufacturers alike. For example, if you own
some recent B&W speakers, chances are
that the crossover is by Mundorf.
3/26/2007 9:07:49 AM
COIL CONSTRUCTION
People who know me are aware that I am often critical of
claims that seem to have no basis in physical reality. I was,
therefore, quite skeptical to hear that Mundorf produces coils
with special construction and impregnation treatment to reduce microphonics1. But to my amazement, when we went
over some third-party measurements, this actually became
quite clear! As Fig. 1 shows, some coils happily resonate mechanically in certain frequency bands and the choice of material, construction, and impregnation can help to suppress this
effect.
You can argue that the microphonics as such would not
be audible with the component mounted inside an enclosure.
But remember: that mechanical resonance needs energy, and
that energy must come from somewhere, and it can only come
from the signal passing the components! The specific effect
on the sound depends very much on the particular place in
the circuit, but it is clear that this effect distorts the signal and
thus the sound.
The expertise Mundorf built up with home hi-fi components also allowed it to branch out to car audio. Notable here
are very high value capacitors to hold up the car’s supply for
those watt-hungry systems–––electrolytics that can deliver
peak currents of several hundred amperes. And, oh yes, they
should be mounted on a heatsink. . . and if you need a 300A
13.8V DC supply to demo your car stereo at home, Mundorf
have those, too.
TUBECAPS
One of the disadvantages of electrolytics is that they age and
lose capacitance over their lifetime. The aging is accelerated
with elevated temperatures often encountered in (tube) power
amps. The solution is a foil cap, but traditionally foil caps have
been very, very bulky for the required capacitances and voltage
ratings for tube amps and supplies and consequently are quite
expensive. Using a very thin, rough foil can increase capacitance for a given volume, but the reliability may suffer because
of production defects and foil puncturing.
FIGURE 1: Vibration of coils of various constructions with signal frequency.
19
3/22/2007 4:01:08 PM
With typical creativity Mundorf attacked the problem and solved it: Accept
the fact that after production the caps
may have some shorted spots, but then
use a special process that “burns off ” a
small conductor area around the short to
restore isolation.
This self-healing effect is not new, but
as far as I know has not been used in this
application area. The result is a capacitor
that is roughly equal in volume to an
electrolytic but with no aging, improved
reliability and less series resistance so
it functions much better and reliably in
PHOTO 4: The Mundorf Air Motion
Transformer diaphragm.
20
audioXpress 5/07
(tube) power amps and supplies. It is
non-polar to boot, so is an excellent high
voltage, high capacitance coupling cap.
AMT DRIVERS
In a corner of the lab I spied an unknown
object, which turned out to be a prototype
Air Motion Transformer (AMT) speaker
driver (Photo 4)! Many readers remember
Oskar Heil’s famous Heil speakers of
the 70s with the patented AMT mid/
high driver, which consists of a pleated
diaphragm with a conductor etched on it,
placed in a strong magnetic field. As the
signal current flows across the diaphragm,
alternate sides of the pleats are attracted
to and repelled by each other: the pleats
open and close in a musical rhythm and
move the air outward and inward, generating a sound field.
Over the past three years, Mundorf pushed this principle toward the
next level in terms of rethinking and
researching the designs leading to the
award of new patents. Today the company produces a wide range of varied
AMT drivers for OEMs, but I wouldn’t
be surprised if they also became available
to individual builders. The measurement
graphs I saw in Mundorf ’s own sound
dead room were impressive.
I am not surprised that Mundorf is
so successful in its endeavors. The company’s employees, who obviously enjoy
what they are doing, are tinkerers in
the right sense of the word, looking for
creative solutions to problems. Being a
family business, Mundorf has the flexibility to do what it thinks is worthwhile
without the immediate worry of the
shareholders. Ultimately, that benefits all
of us audiophiles. aX
More info at www.mundorf.com
FOOTNOTE
1. Strictly speaking, the effect described
here is not microphonics. Microphonics means that the component
picks up a signal from vibrations (air
or mechanical); in this case I describe
a component generating vibrations
from a signal. Maybe we should call
this speakerphonics?
3/22/2007 4:01:17 PM
O
GLASS AUDIO
•
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•
GLASS AUDIO
•
GLASS AUDIO
•
GLASS AUDIO
A Versatile Line Amp for Preamp,
Headphone, and Power Use
Here’s a do-it-yourselfer’s dream in tube amp design: four
operating modes, high quality, low cost, and low distortion.
By Joseph Norwood Still
T
his design offers two versions:
a ten-tube $160 line amplifier
and an eleven-tube preamplifier/headphone amplifier/power
amplifier. Either way, it is probably the
most versatile stereo amplifying device
ever offered to the DIY audiophile. It
offers four operating modes with outstanding specifications: line amp, headphone amp, preamp, and power amp.
The 10W, 0.9% distortion single-ended
(SE) amp is the lowest distortion amplifier I have come across. It also has lower
distortion than published tube manual
push-pull amps.
The damping of the 10W amp is incredible: 2.8V across an 8Ω resistor increases to only 3.8V when the 8Ω load
is removed. This is accomplished in a
standard output circuit without the aid of
negative feedback. It is obvious new design ideals for vacuum tubes are still with
us. Behold! The magic of paralleling and
low mu tubes.
Note: If operating in the 10W SE
Stereo mode, you need to add two Hammond output transformers (P-T1640SE,
$93 each) from Antique Electronics to
the cost.
THE CHOICE IS YOURS
The eleven-tube amplifier (Photo 1)
uses very linear low mu, high perveance
12B4s, and a 12AT7 preamp. Five paral-
PHOTO 1:
Finished unit.
leled 12B4s are used in each channel of
the stereo line amp (Fig. 1). In the line
amplifier mode only the 12B4s are used.
In the headphone/power amplifier/preamp mode, the 12AT7 drives the 12B4s.
The line amp and multi-mode amp do
not employ negative feedback. The line
amplifier frequency response is flat from
10Hz to 50kHz, distortion is less than
0.15% at 2V RMS output, and output
impedance is 420Ω. The voltage gain of
the 12AT7 and five paralleled 12B4s is
125.
Note: The gain of a single 12B4 is 2.4
and five paralleled 12B4s have a gain of
3.8 (cathode un-bypassed) and a gain of
5.6 (cathode bypassed). The plate current
requirement of the five 12B4As is 50mA
when using a plate load resistor of 3.1kΩ.
The selection of the low mu 12B4 enables this line amp to rival or surpass the
performance of the most expensive commercial line amps because they use medium mu triodes. For truly superlative sonic
performance a line amp requires a low
mu, high linearity, high perveance tube,
and the 12B4 meets this requirement. I
know of no other tube more suitable for
this application. The amp is designed to
outperform $3k to $10k commercial line
amplifiers.
GENERAL INFORMATION
Great care is required when operating
tubes in a parallel configuration to prevent oscillation or instability. To address
this problem most resistors in the amp
are carbon composition. The 3.1kΩ, 10W
plate load metal-oxide resistors have 2Ω
carbon composition resistors in series
with the plate load resistors. The cathode
resistors are 220Ω, 5W metal-oxide types.
The construction of the amp requires
short leads as used in R-F circuits, and
an aluminum chassis is mandatory to
ensure good grounding. A ground lug
between tubes 1-5 ensures short leads for
the cathode resistors. The “plate stopper”
2Ω resistors are connected to a terminal strip located between tubes 2 and 3.
Single 2Ω resistors are connected to the
plates of tubes 2 and 3, and two seriesconnected 2Ω resistors are connected to
plates of tubes 1 and 5, respectively. This
May 2007
Still-2756-1.indd 21
21
3/22/2007 4:22:41 PM
GLASS AUDIO
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•
GLASS AUDIO
•
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the headphone. I recommend the $20
65Ω headphones from Behringer, model
HPX2000, or the $50 Audio Technology
units, model ATH-M30, from Parts Express (800-338-0531).
Switch Settings:
1. Set S2 to H.P. position.
2. Set S3 to L.A. position.
PREAMP
FIGURE 1: Multi mode amplifier.
ensures a carbon block exists on the long
lead required to connect tubes 1 and 5 to
the terminal strip and the metal-oxide
plate resistors.
Figure 1 shows that much thought
was given to ensure the stability of the
paralleled 12B4s. The pin configuration
of the 12B4 grid with pins 2 and 7 connected to the grid ensured a short path
for the insertion of a “grid stopper” resistor between tube stages. Pins 1 and 9 of
the tube sockets face toward the middle
of the chassis ensuring short runs for
the cathode, plate, and grid resistors. It
is important that you do not substitute
other types of resistors for the carbon
composition resistors. These recommendations are required to ensure stability
when using parallel tube circuits.
HEADPHONE
Application: The headphone amplifier
circuit uses a 12AT7 to obtain sufficient
audio output signal to drive the five paralleled 12B4s line amp. A 0.25V RMS
input signal to the 12AT7 provides an
output of 9V RMS from the 12AT7output circuit to drive the headphone
12B4s amp (maximum output). The output for the headphones is obtained from
the cathodes of the five paralleled 12B4s.
You operate the headphones by simply
plugging the headphone jack into the
chassis mounted ¼˝ or 3.5mm jack. For
best performance, a 55Ω-75Ω impedance
headphone is required.
The amp has a distortion of 0.4% at
1kHz when using a 65Ω Behringer headphone with a measured output signal
of 0.8V RMS. At a normal output level
of 0.6V RMS, distortion is 0.15%. The
0.6V and 0.8V RMS signals provide a
very loud listening level. The frequency response is flat from 20Hz-50kHz.
The distortion and frequency response
measurements are made at the coil of
SPECIFICATIONS OF LINE AMPLIFIER
1. The frequency response is flat from 10Hz to 50kHz.
2. The 100Hz, 1kHz, and 10kHz square waves are perfect.
3. The distortion at 2V RMS (cathode un-bypassed) is 0.15% -20Hz, 0.12% -1kHz, and 0.14% -20kHz.
4. The distortion at 5V RMS (cathode un-bypassed) is 0.25% -20Hz, 0.2% -1kHz, and 0.24% -20kHz.
5. The distortion at 2V RMS (cathode bypassed) is 0.25% -20Hz, 0.15% -1kHz, and 0.36% -20kHz.
6. The distortion at 5V RMS (cathode bypassed) is 0.38% -20Hz, 0.25% -1kHz, and 0.5% -20kHz.
7. Clipping occurs at 30V RMS.
8. Plate impedance is 420Ω.
9. Voltage gain is 3.8 (cathode un-bypassed) and 5.6 (cathode bypassed).
Important: If you require only the line amplifier mode, omit switches (S2/S3), headphone jacks, and the 12AT7.
24
Application (Optional): You may choose
the preamp mode if you prefer to build a
power amp that uses only a single amplifying stage to drive a SE power output
tube or a phase inverter stage to drive
push-pull power output tubes. The preamp mode is the same as the headset
mode except the output signal is obtained from the plates of the 12B4s via
the 0.47µF capacitor. If you require direct
DC coupling of the phase inverter, add a
separate pair of RCA female jacks.
The +115V DC output f rom the
12B4s (3.1kΩ) plate load resistor is compatible with the DC grid voltage requirements of most direct-coupled phase inverters. The voltage gain of the 12AT7
and 12B4s is 125 with cathodes of the
12B4s un-bypassed and 185 with cathodes bypassed. The distortion at 10V
RMS output is 0.4% at 32Hz, 0.34% at
1kHz, and 0.6% at 20kHz with 12B4s
cathodes un-bypassed. The distortion at
30V RMS output is 1.2% at 32Hz, 0.65%
at 1kHz, and 1.5% at 20kHz with cathodes bypassed. The frequency response is
flat from 20Hz-25kHz.
Switch Settings:
1. Set S2 to H.P. position for a voltage
gain of 125 or to P.A. position for a
voltage gain of 185.
2. Set S3 to L.A. position.
POWER AMP
Application (Optional): To operate in
the power amp mode, set the DPDT
switch (S3) from the line amp mode to
the power amp mode and the DPDT
switch (S2) to place the negative end of
the 1000µF capacitor at ground. The S3
switch setting connects the 3.1kΩ resistor from the plate circuit of the 12B4s
to the 1200Ω 25W Hammond output
transformer of the SE-12B4s. The power
output of the SE 12B4s is 10W, and the
distortion at this level is 0.9% with no
May 2007
3/22/2007 4:23:28 PM
GLASS AUDIO
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loop feedback. The frequency response is
flat from 30Hz to 15kHz at 10W output
level. A 0.7V RMS input voltage is required for 10W output from the power
amp.
The plate current of the five 12B4s
in each stereo channel is 148mA. The
damping criterion of the amplifier is excellent. At 10W without feedback, the
distortion at 32Hz is 2.0%, at 50Hz it
is 1.2%, and is 0.9% at 1kHz, 1.6% at
5kHz, and 2.2% at 15kHz, respectively.
At 5W the distortion is 0.8% at 32Hz, at
50Hz it is 0.7%, and 0.6% at 1kHz, 1.5%
at 5kHz, and 1.8% at 15kHz, respectively.
These specifications are outstanding for
a single-ended amp, especially when op-
•
•
GLASS AUDIO
erating with a low plate voltage of 260V
DC.
Switch Settings:
1. Set S2 to P.A. position.
2. Set S3 to P.A. position.
Note: A separate volume control is provided for R and L channels to prevent
crosstalk. Switch S1 (Fig. 2) turns the
heater supply on and off, and switch S2
turns the high voltage supply on and off.
The H.V. switch S2 only operates if the
heater switch is set to the on position.
Of Interest: The right and left channel amps are connected in a monaural
AMPLIFIER PARTS LIST
Qty. Value
Watts
Part No.
Note: Double Parts for Stereo
14
2Ω
½W
30BJ500-2
1
220Ω
5W
286-220
2
330Ω
½W
30BJ500-330
4
680Ω
½W
30BJ500-680
2
1.0kΩ
1W
282-1K
2
6.2kΩ
5W
286-6.2K
6.2 K/5W resistors are paralleled to form 3.1 K/10W resistor.
2
100kΩ
2W
282-100K
1
470kΩ
½W
30BJ500-470K
1
100K, single
½W
31VJ501
1
0.22µF, 400V orange drop
715P22454MD3
1
0.47µF, 400V orange drop
715P47454MD3
1
1.0µF, 250V, metalized
1430-2105
1
220µF, 35V
1
1000µF, 400V
EC-1000
Note: Do not double the parts listed below:
1
6-position, dual deck, 2 pole
275-1386
1
knobs, 4-per pack
275-415
4
5-lug tie point, terminal strips
274-688
10
RCA, female, phono jacks
161-002
1
5-conductor, shielded cable, 10´
P.N. 5C-S22
1
single conductor, shielded cable, 3´
P.N.42-2371
11
T-61/2 tube socket, 9-pins
P-ST9-137R
10
12B4A (NOS)-Order 12
Note: Parts shown below are not required for line amp mode operation.
1
DPDT rocker switch
112-R13-130B
1
DPDT toggle switch
10TF115
1
12AT7 (Note: I recommended you order 3)
1
1,200Ω primary-4,8,16Ω secondary, 25W (P-T1640SE)
Note: Transformer is only required for SE 10W power amp.
Type
Mfg.
Carbon Composition
Metal Oxide
Carbon Composition
Carbon Composition
Metal Oxide
Metal Oxide
M
M
M
M
M
M
Metal Oxide
Carbon Composition
audio taper
polypropylene
polypropylene
polyester
electrolytic
electrolytic
M
M
M
M
M
M
ALEL
ALEL
RS
RS
RS
M
ALEL
RS
AE
AE
micalex
¾˝ hole
miniature
M
M
AE
AE
POWER SUPPLY PARTS LIST
Qty. Value/Designator
1
1
3
2
1
1
1
1
1
1
1
1
1
1
1
1
100kΩ (R-1) 2W
4700µF, 25V (C1)
470µF, 400V (C2-4) 1.4˝ dia. × 1.63˝
SPST switch (S1-2) ¾˝ panel hole, 125V
Bridge rectifier (BR-1) 25A, 200V
Bridge rectifier (BR-2) 4A, 600V
1.0H, 300mA choke (L1)
2.0H, 100mA choke (L2)
115V-12.6V ct, 6A (T1)
115V-230V, 0.860A(T2) 100VA
3AG fuse block
1 pack, 1A, 3AG, 5-fuses
3 connector, AC power cord
Chassis box, aluminum 16˝ × 8˝ × 3˝
1 pack Rubber feet-Self stick, rubber
115V AC Fan, 32 cfm, 80 × 80 × 38mm
26
Part No.
Type
Mfg.
282-100K
metal oxide
electrolytic
electrolytic
Rocker Sw.
1 1/8in2
M
ALEL
ALEL
M
ALEL
ALEL
AE
AE
M
M
ALEL
ALEL
AE
M
RS
M
EC-4740
112-R13-130A
FWB-252
FWB-46
P-T158T
P-T154M
546-166Q12 (Hammond)
553-N77U (Triad)
FHBL-3
FS-1
S-W206
548-1444-28
64-2342
433-3E-115B
GLASS AUDIO
•
GLASS AUDIO
configuration by adding audio interstage
transformer A.E.- P-T124B (10K to 90K
C.T.) to the input of the amp and P-P,
120W, 1900 C.T. (A.E.-P-T1650T) to
the output of the right and left channels. No changes are made to the 11-tube
amp. The power output of the monaural
amp is 24.5W, and the distortion at 1kHz
measures 1.2%. A large percentage of the
distortion is attributed to the input transformer.
The frequency response is relatively flat
from 20Hz to 15kHz. Sensitivity is 0.56V
RMS for 24.5W with 33kΩ resistors connected across the secondary windings of
the interstage transformer. I do not recommend this amp for use as a monaural
amplifier, but I thought the information
here might be interesting. The 35W/60W
amp presented in aX June 2004 is more
cost effective and a more desirable choice.
POWER SUPPLY
The power supply (Fig. 2) contains a
DC heater supply and a high voltage
plate supply. The DC heater supply uses
a 12V AC, 6A transformer connected
to a 25A full-wave bridge rectifier. The
output of the rectifier is connected to a
4700µF capacitor, and the filtered output
voltage from the capacitor is connected
to the heaters of the 12B4s and 12AT7.
The ripple voltage of the supply is 0.9V
RMS.
The high voltage plate supply uses a
100 VA line transformer with the output
secondary winding wired for 230V RMS.
This output voltage connects to a fullwave bridge, and the 260V DC output
from the bridge is connected to a 470µF
capacitor and a 1.0H 300mA choke. The
output of the choke is connected to a
470µF capacitor, and is additionally filtered by a 2H choke and a 470µF capacitor connected to the 12AT7 plate
circuit. The ripple voltage of the 260V
DC supply is 0.3V RMS. The bleeder of
the power supply is 100kΩ, 2W.
CONSTRUCTION
The amplifier I originally built as a line
amp on a 13.5 × 5 × 2˝ chassis is shown
in Photo 1. The chassis was very crowded
(wiring) and required mounting the heater transformer on the rear of the chassis. To avoid the crowding and to mount
the heater transformer on the underside
of the chassis and the two 25W output
May 2007
3/22/2007 4:23:37 PM
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with the power transformer, and position
Headphone
Preamp Line amp 10W Power Amp
the five right-channel
12AT7
12B4
12AT7
12B4
12B4s and five leftEbb
260V
260V
260V
260V
channel 12B4s on eiEb
110V
115V
105V
220V
Ek
1.5V
12V
1.6V
33V
ther side of the 25W
Ib
1.5mA
50mA
1.6mA
148mA
output transformer.
Plate Dissipation (no signal) per 12B4 tube
6.5W
Connect the four
Plate Dissipation (max signal) per 12B4 tube 4.5W
electrolytic capacitors
Efficiency of 10W Power amp
24%
in the power supply to
transformers on the top of the chassis, terminal strips on the underside of the
I recommend that you use a 16 × 8 × 3˝ chassis. Mount the fan facing the power
chassis for this project. Mount the power transformer and notch out the bottom of
transformer at the end-middle of the the chassis to enable the fan to cool the
chassis. Locate the two output transform- electrical components on the underside of
ers in the middle of the chassis in line the chassis.
12AT7& 12B4 OPERATING VOLTAGES
FIGURE 2: The power supply schematic.
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CONCLUSION
I believe this amplifying device is the
most versatile and outstanding in technical specifications for each of its four
operating modes of any device currently
offered to the audiophile. The headphone
and 10W SE amp modes are especially
ideal for apartment dwellers. If you elect
to build this amp, you will enjoy many
hours of listening entertainment. aX
WARNING: Dangerous voltages are
present, exercise extreme caution when
working on the line amp and never leave
the line amp upside down when children
are present.
If you encounter any problems or have
any questions, contact me at 302E. Joppa
Rd., Apt. 1911, Towson, MD 21286.
Distributors:
RS - Radio Shack
ALEL – All Electronics
AE - Antique Electronics
M – Mouser Electronics
May 2007
800-826-5432
480-820-5411
800-346-6873
27
3/26/2007 9:05:52 AM
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Heathkit W-7M Rebuild
Follow this author’s successful method of
rebuilding vintage equipment from Heathkit.
By Bruce W. Brown
PHOTO 1: The finished rebuild.
A
s the first power amplifier that
broke $1 per watt barrier of
listening power, the Heathkit
W-7 was a major breakthrough
for audiophiles. Introduced in 1958, it
carried a list price of $54.95. The W-7
is not as common as the W-5, and when
it comes up at online auctions from time
to time, does not seem to attract the interest of many bidders. Their loss!
FEATURES
FIGURE 1: Heathkit W-7 circuit.
28
Brown-2786-3.indd 28
Each amp used two EL34s as output
tubes, and a single 6AN8 tube as a preamp/phase inverter. The power supply
used four silicon diodes in a voltage doubler. At 8½˝ deep by 6⅛˝ high by 15˝
wide, the unit was quite compact. Early
models featured a satin gold enamel
chassis with black wrinkle cages; later
models had a black chassis and gold
cages. The final models were designated
“AA-91,” and production ended in the
early 60s1.
Specifications were quite impressive.
Frequency response was ±1.0dB from
6Hz to 30kHz at 0.25W, and ±0.5dB
f rom 20Hz to 20kHz, at maximum
rated output of 55W RMS2. Harmonic
distortion was about 0.1% at 1W, rising
to 0.5% at 20W and 1.5% at 55W. (A
2% total harmonic distortion is acceptable in music reproduction.) Hum and
noise was 80dB below 55W. The amp
utilized 25.5dB of negative feedback
and featured 4.8-16Ω outputs as well as
70.7V line output. It also featured a variable damping factor switch on the front
panel, enabling the selection of either
maximum (20) or unity (one)2.
The output transformer in this unit is
huge, much larger than the Mark 3 Dynaco amp that used 6550s, and is probably the reason this amp sounds so clean.
May 2007
3/22/2007 3:39:37 PM
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OPERATION
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The circuit of the W-7 is simple and
straightforward (Fig. 1). The signal is
fed through a gain control and 0.1µF
coupling capacitor to the control grid
of the 6AN8. The amplified signal is
coupled through an RC network to the
grid of the triode section of the 6AN8,
which is used as a split-load phase inverter.
The signal at the cathode of this stage
follows the grid, while the plate voltage
swings in the opposite direction. No
amplification occurs at this stage. The
two out-of-phase signals are coupled
through 0.25µF coupling capacitors to
the control grids of the EL34s. The output stage is operated in Class AB1; the
screen grids of the output tubes are connected to taps on the primary of the
specially designed output transformer.
A unique and elaborate bias supply
and balancing circuit is incorporated in
the W-72. This ensures accurate balance of the output tube’s plate current.
Control R30 adjusts the bias voltage and
is set to yield about –38V on the control
grids of the EL34s, while control R38
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allows you to balance the current between the tubes. This lets you use lessthan-perfectly-matched output tubes.
The high voltage power is supplied
by a voltage doubler from the 180V
AC tap of the power transformer. The
150µF and 200µF electrolytic capacitors are each rated at 300V, and in series
would be equivalent to 75µF at 600V,
and 100µF at 600V. This is a nice safety
margin, because the power supply is de-
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signed to 495V.
As I have mentioned in other articles,
this philosophy was much different than
that of Dynaco. In most Dynaco power
amps, if the voltage used was 500V, then
the capacitors were rated at 525V, and
multicap cans were used to further cut
costs and save chassis space.
Heathkit comments: “The circuitry of
the W-7 was carefully engineered to ensure an unusually high degree of stability
PHOTO 2: The original unit.
May 2007
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3/22/2007 3:39:42 PM
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PHOTO 3: Capacitor replacements.
at both low and high frequencies which
may be verified by observing the 0.25W
frequency response curve—Graph A2.
The smooth rolloffs below 15 cycles and
above 50 kc are indications of the unusually wide stability margins. . . The excellent power supply regulation and low
impedance of the phase-splitter minimize grid current effects as the rated
power is approached and exceeded. This
results in less than 1% IM distortion at
the rated 55W; the overload above 55W
is gradual, and clipping is symmetrical.”2
Another unique feature of this amp is
a sophisticated damping control. Heath’s
explanation of the utility of this control is
interesting: “Damping factor is defined
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as the ratio of load resistance to the internal resistance of the amplifier. . . It has
been found that speaker systems which
have a high degree of acoustical damping
may be over damped when used with an
amplifier having a high damping factor,
with a resulting loss of bass efficiency.
On the other hand, too low a damping
factor will result in boomy or one note
bass which is undesirable”2.
Speaker systems of this era were often
efficient ported systems, or horns, but
newer “acoustical suspension” systems
were starting to appear on the market.
This may be the reason variable damping was provided.
Dynaco equipment provided an economical entry into high fidelity. But
when comparing the Dynaco Mark
IIIs or IVs to the Heathkit W-7s, you
will find that the Heathkit betters the
Dynaco hands down. While the sound
is somewhat similar, the Heathkit has
more bass “kick,” and the treble sounds
cleaner.
Another difference involves the transformers: Dynaco power transformers
run quite warm, and after several hours
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of use are uncomfortable to touch. The
Heathkits run cool, and after several
hours are only warm to the touch. Sideby-side comparison of the Dynaco and
the Heathkit shows where the “big iron”
is; while the power transformers are
similar in size, the output transformer of
the Heathkit is half again the size of the
Dynaco’s. While there is nothing wrong
with the Dynaco transformers (I have
many stock and custom amps that use
them), I would rather get my hands on a
W-7 output than Dynaco!
REPLACEMENTS
The unit as I received it is shown in
Photo 2. As with my other rebuilds, I
usually start with the power supply. In
this unit, all the diodes in the voltage
doubler were fine, but the electrolytic
capacitors were leaky. I was able to find
some 100µF 450V caps to replace the
100µF 300V versions in the first section.
The original caps were about 1½˝ in
diameter and about 4˝ tall. The new ones
are about one-tenth this size (Photo 3). I
removed the originals and mounted a
piece of fiberglass board in the cutout
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30
May 2007
3/22/2007 3:39:47 PM
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and glued the new snap-mount ones to
this board.
I was also fortunate to find four
330µF 350V electrolytic caps to replace
the 200µF 300V originals. These were
short, but sized to fit through the original chassis holes, and I mounted them
using some chassis mounts (Photo 3).
Also shown in this photo is the original
EL34 mount plate, which was extremely
rusty. I wire-brushed and painted it.
Note in Photo 1 the vented cages my
friend Larry built for me to cover the
new power supply capacitors. (Though
these cages were not necessary for operation, he believes they give the amps
a finished look.) I replaced the selenium
diode in the bias supply with a silicon
diode, and also replaced the electrolytic
capacitors in the bias supply, using 100µF
160V units for all (Photo 4). I replaced
the remaining electrolytic caps feeding
the phase inverter and preamp sections.
I then powered up the first amp with
the Variac, and did some voltage measurements. Everything was in spec except voltage to the input grid of one
of the EL34s. This type of problem is
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usually due to a leaky coupling cap, as
was true in this case. I powered down
the amp, and after waiting for the power
supply to discharge, replaced all the coupling caps with new polypropylene. (My
previous articles have covered capacitors
and my thoughts on them, so I will not
repeat here except to say I do not believe
in spending huge amounts of money on
“premium” caps. Use what you like.)
While you are under the chassis, use
your ohmmeter to measure the value of
all the resistor values. You will generally
find several that have drifted off value,
and this will affect the operation of your
PHOTO 4: Diode and electrolytic capacitor
replacements.
•
finished product. The 100K (R15 and
R10) resistors that feed the EL34s should
be very closely matched. Also carefully
check the cathode resistors on the preamp section of the 6AN8 (Photo 5).
These amps use a “surgister” as a surge
suppressor. Located in the corner of the
chassis near the fuse holder, it is about
PHOTO 5: checking the amp's innards
May 2007
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3/22/2007 3:39:56 PM
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1½˝ long and looks like a small tube
with wire wrapped around it. On the
back side of the tube is a bimetallic strip,
which closes the contacts as the wire
heats up, acting essentially as a 100Ω
nichrome resistor that generates heat
and closes the contacts, bypassing itself
(Photo 6). I replaced this with a surge
suppressor (sold for 50 cents by Electronic Goldmine), which works well for
suppressing the turn-on surge of a tube
amp with silicon rectifiers and cold filaments.
PHOTO 6:
Heathkit’s
surgister.
One of the interesting things I found
with my pair of amps was that one was
completely hand-wired as outlined in
the manual copy I had purchased. The
other amp had a factory-wrapped wiring
harness. I have not been able to find additional information on this, but it was a
very professional-looking harness, much
like those seen in vintage military radios.
I contacted the manual copy source and
asked whether they had any information
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on a W-7 manual with a factory-made
wiring harness, but they had not seen
any other W-7 manuals. This may also
be a factory-wired unit.
UP AND RUNNING
The tubes used in this amp are very
common and readily available. NOS
6AN8s are quite reasonable. The choice
of available EL34s provides some outstanding representatives. I often use
Svetlana EL34s, finding them very durable, and producing a nice clean crisp
sound.
One of the nice things about the W7 is that it does not need a matched
set of output tubes, because the balance
adjustment allows you to balance the
current to the output tubes (so you can
even use vintage unmatched tubes). The
manual directs you to rotate the balance
control to mid position and connect a
voltmeter to the meter jacks on the back
of the amp (the bias control should be
fully counterclockwise and no inputs
or speakers should be connected at this
time), then turn on the amp and allow
it to warm up for several minutes. You
can see the two adjustment controls in
Photo 7.
Adjust the balance control to get a
zero reading on the voltmeter, continue
to adjust to the lowest range of your
meter, and readjust until it is zero. Once
this is done, remove one of the leads of
your voltmeter and connect it to the bottom terminal below the 70V output on
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the speaker strip. Adjust the meter for a
reading of 0.36V, measuring across the
6Ω cathode resistor. This corresponds to
60mA current per tube (switch the lead
between the meter jacks and observe the
voltage on each tube).
I dug around in my used tube supply
and found some Mullards, Tung Sols,
and new Sovteks to try, and was fascinated by comparing the sound differences between the new and old tubes. I
liked the Tung Sols the best, followed
by the Svetlana, then the Mullards, and
finally the Sovteks. (Not very scientific,
just personal preference.)
These amps will drive just about any
reasonable speaker system, they run very
cool, and are not the least bit fatiguing.
Very easy listening! aX
[An additional skill I developed in this
rebuild was replication of cosmetics, including logos and cabinet feet. These items are
unique, and replacements are extremely hard
to find. With materials available today, it is
easy and fun to reproduce your own parts,
giving your restoration an authentic appearance (please see my upcoming article, “Casting Replacement Parts”).]
REFERENCES
1. Vacuum Tube Valley, Issue 2, Volume 1,
Fall 1995.
2. Heathkit Model W-7M Assembly Operation Manual, W7FG Vintage Manuals,
www.w7fg.com, 918-333-3754.
3. Schematic.
PARTS LIST FOR W-7 REBUILD, PER UNIT
2 100-200µF 450V electrolytic (C13-14)
2 300-400µF 400V (or better) electrolytic (C17-18)
3 100µF 150V electrolytic (C15, C18, C19)
1 1N4005 silicon diode (replaces selenium)
1 25µF 50V electrolytic (C2)
1 25µF 450V electrolytic (C6)
1 0.1µF 400V coupling cap (C1)
1 0.25-0.27µF 600V electrolytic (C7-8)
1 surge suppressor, 2A minimum
(replaces R27 surgister)
PARTS SOURCES
Electronic Goldmine
www.goldmine-elec.com
Antique Electronics Supply
www.tubesandmore.com
PHOTO 7: Adjustment controls.
32
Mouser Electronics
www.mouser.com
May 2007
3/22/2007 3:40:02 PM
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REVIEW
Tube Imp Mini Tester
By Charles Hansen
PHOTO 1: Tube Imp in its carrying case.
T
he Tube Imp mini tube tester,
designed and built in the UK,
measures the steady-state current,
gain, and transconductance characteristics of any B9A double-triode tube
with the standard ECC83/ECC88 footprint.
Operating at 12V AC from the included external power supply, the Tube
Imp offers matching and parameter testing of a range of commonly used doubletriode tubes to individual tube equipment
enthusiasts as well as retail outlets and
small OEM concerns.
Specifications:
HT voltage maximum: 200V
HT voltage adjustment: 0-200V
HT setting accuracy: Typically better
than ±2%
HT current: 10mA maximum
HT current limit LED: >12mA
HT impedance on gain setting: >1MΩ
Grid voltage range: 0 to -10V
Grid voltage setting accuracy: Typically
better than ±2%
Heater voltage: 6.3/12.6V DC, switchable
Heater current 350mA, 500mA for less
than 5 minutes
Measurement accuracy:
Cathode current: ±2%
Transconductance: ±4%
Gain: (Anode impedance in kilohms)
±5%
Valves that can be tested:
ECC81/12AT7
ECC82/12AU7
ECC83/12AX7
ECC88/6DJ8WA/6922
ECC189/ECC803
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British Audio Products/Moth Group
10 Dane Lane, Wilstead, Bedford,
Bedfordshire, UK MK45 3HT
www.britishaudio.co.uk or
www.tube-imp.co.uk
e-mail: [email protected]
++44(0)123 474 1152
Fax: ++44(0)123 474 2028
Price: £299 UK
Test unit dimensions:
Net weight with carrying case: 4.4 lbs (2kg)
Available online at http://store.securehosting.com/
stores/sh204131/shophome.php?itemprcd=tubeimp
The transformer at the left is a 120/120
to 12/12V AC step-down unit that is
wired in reverse; the external AC adapter
feeds 12V AC to the paralleled transformer secondaries, and this is stepped up to
240V AC via the primaries connected in
series. The AC primary voltage is rectified and used to supply the adjustable HT
(plate) DC voltage. The smaller heatsink
just below the transformer sits on the HT
regulator MOSFET. The adjacent 22µF
450V electrolytic filters the HT DC voltage.
The large heatsink in the middle holds
the filament DC regulator, which is powered directly from the AC adapter. A
switch on the front panel allows you to select either 6.3V or 12.6V DC. The manual
advises starting 12.6V tubes in the 6.3V
position, then switching to 12.6V after a
few seconds to minimize the stress on the
filaments.
The 9-pin gold-plated ceramic tube
socket pins are just to the right of the
transformer. Pin 9 of the tube socket is
not connected. Four small trimpots visible
through holes in the PC boards allow factory adjustment of the Tube Imp.
INSIDE THE TUBE IMP
The Tube Imp comes in a nice carrying case that includes the tester, 12V AC
power adapter, and the manual (Photo
1). I plugged one of my vintage Mullard
ECC83 tubes, err valves, into the Tube Imp
(Photo 2).
The tester is quite rugged, constructed
of red powder-coated heavy gauge steel.
The 2.1mm × 5.5mm AC power jack is located on the rear panel. Photo 3 shows the
Tube Imp with the bottom cover removed.
All chassis components are mounted on
two PC boards with some hard wiring also
involved.
PHOTO 2: Tube Imp testing Mullard ECC83.
PHOTO 3: Tube Imp interior view.
May 2007
Hansen2805-2.indd 33
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TOPOLOGY
The HT supply consists of a source follower MOSFET. The Test Selector allows
you to switch between a voltage source and
a current source in order to provide the
optimum loads for transconductance and
gain testing. The MOSFET is protected
by a 12mA current limit circuit, which also
lights a red LED when the HT supply is in
current limit. A switch on the front panel
allows testing of either the A or B section
of the dual triodes. The Section-1/2 switch
automatically switches the grid and cathode
connections. The tube section not selected
is idled with a -15V grid bias.
When the Test Selector is in the gain (µ)
position, the HT MOSFET is switched
into current source mode with an equivalent impedance of about 2MΩ. An AC signal is fed to the grid, and the anode voltage
is rectified and displayed on the front-panel
digital meter.
When the Test Selector is in the mA/V
(transconductance, or gm) position, the HT
is supplied by the MOSFET as a regulated
voltage. A small AC signal is fed to the
grid, and the resultant AC current is used
to calculate transconductance for the front-
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With 30+ labels (inputs/outputs)
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Hundreds of ways to assemble
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34
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panel digital meter. With the Test Selector
in the mA position, the drop across the low
value cathode resistor is used to measure
the cathode current.
The Tube Imp does not directly measure
plate resistance, but you can easily calculate
it by Rp = µ/gm. By plotting a series of
cathode current versus plate voltage at a
fixed grid voltage, you can produce a plate
curve for your own tube.
MEASUREMENTS
I measured the actual heater, grid, and plate
voltages at the tube socket as selected by the
front-panel control settings. I used a power
resistor decade box set for 18Ω (6.3V) and
36Ω (12.6V) to check the heater voltages
at 350mA. I didn’t use any grid resistor to
measure the grid voltage because my Fluke
DMM (digital multimeter) has a 1MΩ
input impedance. I used a 220k resistor (A
to K) to simulate the plate resistance and
also checked the current limit point using
a 10k 5W 1% load resistor, raising the HT
until the LED was lit.
The filament voltages measured 6.44V
DC and 12.38V DC at 350mA, or within
about 2% of nominal, with pin 4 being
positive. I measured the grid and HT voltages with respect to the scale markings silkscreened on the front panel of the chassis. I
centered each line on the scale at the center
of the notch on the adjustment knob.
The grid voltages were consistently lower
than the scale markings, from -20% on the
low end to -10% at the high end. The grid
voltage at the maximum CW (clockwise)
rotation read -9.2V DC. Perhaps this could
be easily fixed with an internal trimpot adjustment or a tweak of the knob set screw.
The grid voltage of the idled tube section
had -11V DC rather than the specified
-15V DC.
The HT voltage settings were quite accurate, varying only 1.6% above 20V on
the scale. The HT current limit LED just
began to glow at 12.6mA and was fully on
at 13mA.
Finally, with the correction factors in
hand for grid voltage, I checked the gain
and transconductance using the Mullard
ECC83 for the two Class A amplifier conditions specified in the RCA Receiving
Tube Manual RC-30. For the Tube Imp,
I set the specified plate voltage and then
adjusted the grid voltage to produce the
specified cathode current. Note that for
the second Class A amplifier condition I
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FIGURE 1: Audiomatica Sofia plate
curves for Mullard ECC83.
needed to use Va = 200V DC rather than
the 250V DC specified in the RCA manual due to the Va limitation of the Tube
Imp. I also took these same data points
from the plate curve data file I ran on the
Audiomatica Sofia (Fig. 1).
The results of these tests on section 1 of
my Mullard ECC83 are shown in Table 1
and Table 2. The Sofia displays µ, gm, and
Rp directly along with plate current Ia. The
Tube Imp measures gain (µ), mA/V (gm),
and cathode current (Ik), so I calculated Rp
from the formula Rp = µ/gm.
Table 1
Parameter
µ
gm (mA/V)
Rp kΩ
Data Book
100
1.3
80
Sofia
95.2
1.17
81.2
Tube Imp
72.4
1.3
55.7 (calc)
Measurements, Mullard ECC83 section 1
RC-30 Class A: Va = 100V, Vg = -1V Ia = 0.5mA
Note 1: Rp calculated from Rp = µ/gm for Tube Imp
Table 2
Parameter
µ
gm (mA/V)
Rp kΩ
Data Book
100
1.6
62.5
Sofia
98.1
1.64
59.8
Tube Imp
72.1
1.2
58.1 (calc)
Measurements, Mullard ECC83 section 1
RC-30 Class A: Va = 200V, Vg = -2V Ia = 1.2mA
Note 1: Rp calculated from Rp = µ/gm for Tube Imp
Note 2: Va held to 200V due to Va limit of Tube Imp
CONCLUSION
The Tube Imp consistently understated
the gain by about 26% in comparison to
the Sofia. There is some discussion of this
in the Tube Imp manual. The gm was 11%
high in the first test and 27% low in the
second test. Again, there is a discussion
of calculating the theoretical value of true
transconductance in the manual.
I repeated the Tube Imp tests by setting
the plate and grid voltages at the designated
values and accepting whatever cathode current resulted from these settings. The µ and
gm results were essentially the same.
I believe the Tube Imp would be quite
May 2007
3/22/2007 4:14:15 PM
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valuable in matching tubes where the absolute values of µ are less important than the
comparative values.
MANUFACTURER’S RESPONSE:
Thank you for the opportunity to comment
on the in-depth review of the mini TT.
The reviewer gives a fair assessment of the
mini TT and of its target market. There are,
however, a couple of points I need to comment upon.
The grid circuit has a source impedance
of 1MΩ at DC, so reading the voltage at the
grid pin with a DVM (typically of 10MΩ
input impedance) would cause a drop of
around 10% from the actual set value, as
found by the reviewer. The safest place to
measure the grid voltage with a meter of
less than infinite input impedance is at the
wiper of the grid voltage pot, although not
accessible without taking the back off!
The gain readings of the ECC83 measured by the reviewer also need to be addressed. If we understand correctly the
Sophia calculates gain, from measurements of the transconductance and anode
impedances. The mini TT cannot reach
these calculated values for gain. It measures
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gain directly, feeding a signal to the grid
and measuring the amplified signal at the
anode/plate. This requires a high impedance current source for the anode load. The
impedance of the current source will appear
in parallel with the anode impedance, giving
a slightly lower reading.
For the mini TT we originally specified
an IRF730 MOSFET, configured to act as a
voltage or current source. The circuit is quite
simple and relies completely on the MOSFET’s high impedance behavior in current
source mode to work effectively. MOSFETs
are known to work as an almost perfect current source (theoretically), when the gate to
source voltage is held constant. This simple
circuit works well; however, we were disappointed by the gain measurement results
found during the review of the mini TT.
Checking the production unit, we found
that the current source impedance was lower
than we were expecting, at about 200kΩ at
typical ECC83 cathode currents. This in
conjunction with the ≈60kΩ anode impedance of the ECC83 results in the reduced
gain reading seen by the reviewer.
On further investigation it transpires that
not all “IRF730s” are created equal with re-
spect to their current source behavior. After
much measuring of different samples and
scouring datasheets, we found that unfortunately the ST IRF730—which we had
sourced—is probably one of the worst current source MOSFETs there is, although
the specs are all the same! Every other manufacturer’s IRF730 does better!
We have now found a good replacement
with a (dv/di) drain impedance of 240kΩ
at 10mA (rather than 15kΩ of the ST
IRF730), and have modified the mini TT
test rig and test procedure to test for this
parameter.
With the new MOSFET fitted we get
the following results for gain, from a random selection of tubes:
Device
Aged Mullard 12AX7
Sovtek 12AX7WXT
Gold Dragon E83CC
Telefunken E88CC
Mullard 12AT7
Brimar 12BH7
Gain Cathode current
90.5
2.08mA
127.1
1.94mA
94.8
2.88 mA
34.8
5.00 mA
64.4
10.00 mA
19.4
10.00 mA
Anode volts
180V
180V
180V
80V
150V
100V
Best regards,
Hamilton Cleare
Tube IMP aX
May 2007
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An Improved Method of Finding
Vacuum Tube Model Parameters
Designers and others who work with tubes will find this program
handy to determine VT models.
By Bill Elliot
I
first became interested in vacuum tube models when I needed
good triode and pentode models
to use with my SPICE circuit
simulator to simulate an audio power
amplifier I was designing. Searching
the Internet, I discovered the vacuum
tube model work of Norman Koren1.
I then began to produce some model
netlist equations based on Koren’s
phenomenological triode equations
using Texas Instruments’ Derive 6
math software.
At first, I used trial-and-error
methods to find the parameter valTable 1.
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ues until they matched five chosen points on the plate
curves. Despite the tediousness, I managed to produce a
fair number of netlists for the tube types that interested
me.
I have been using Derive since its early DOS days
and have found it to be reliable, easy to use, and, above
all, it does everything I need and more. It is also very
inexpensive—about $115. I know other math programs
are very good, but they are way out of my price range.
I suggest anyone interested in Derive 6 read the article
on the Internet titled “Derive 6: Far too good for just
students.”2
Because of the tedious work and resulting inaccuracies, I decided I needed to find a better way and
wondered whether I could use Derive 6 to calculate the
values of the parameters.
SOFTWARE SOLUTION
Examining Koren’s triode equations, it would seem that
the five unknown parameters can only be found using
trial-and-error methods or some sort of program using
iterative techniques.
Koren’s Triode Equations:
Ip = (E1X/kG1)*(1 + sgn(E1))
where:
E1 = (Ep/kp)*log(1 + exp(kp*(1/u
+ EG/√(kVB + Ep2))))
Upon closer examination of Koren’s equations, I saw
that there are really two separate sets of parameters.
The first set of two, kG1 and X, are independent of the
second set and are quite easily found using the zero bias
line of the triode plate curves. The second set of three,
kp, u, and kVB, are more difficult to find, but can be
directly calculated using algebraic and numerical programming techniques.
After numerous attempts, I wrote three very simple
Derive 6 programs that directly calculate triode, pentode, and diode model equation parameters based on
Koren’s triode equations.
The triode and pentode programs run in 15–60 seconds on my computer, depending on tube type and data
point selection. Some tube types and data point selections may run on and on without giving a result. You
may even get negative or complex numbers for results.
In these cases, check your data entry numbers first. If
the numbers are OK, then try moving data points B1,
B2, and C to higher current levels. I cannot promise
that these programs will work on any tube type, but I
think they will work on most.
May 2007
37
3/22/2007 4:03:01 PM
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The Pentode Program is shown in
Table 1 . All the programs plus the
step-by-step equation development are
available on my URL3 in Rich Text
Format (RTF) and Derive 6 format
(DFW ). You can download and read
RTF files with Windows Wordpad. If
you have Derive 6, you can download
the Derive files and run them on your
computer.
The pentode equations for a pentode netlist are shown in Table 2. Normally, you would use only equations 3
and 4 in the netlist because a triode
can be obtained by connecting a plate
and screen together. The pentode-triode connection equations are used to
check the triode curves using the Derive 6 curve plotter.
aX
1/3 page ad for Glass Audio Insert
All fonts are Helvetica
REFERENCES
Contact: Phil Marchand
Phone (585) 423 0462
1. www.normankoren.com/Audio/
Tubemodspice_article.html.
FAX
(585) 423 9375
2. www.scientific-computing.com/
scwmarapr04derive6.html.
[email protected]
3. www.knology.net/~billelliot.
s
r
e
s
o
r
v
c
t
o
e
s
l
r
o
n
i
C
Tube Electronic Crossovers . Solid State Electronic Crossovers.
E
c
Marchand Electronics Inc.
PO Box 18099, Rochester, NY 14618
(585) 423 0462
[email protected] www.marchandelec.com
Phono Preamp. Moving Coil Transformers. SP/DIF Phono Options. Custom Solutions. We
Passive line Level
crossovers. Tube Electronic Crossovers. MOSFET SE Amp. Tube Phono Preamp.Solid State
can add notch filters, baffle step, etc..... Tube Electronic Crossoverrs.Kits and Assembled
O
May 2007
39
3/22/2007 4:03:10 PM
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AUDIO AID
Greek Gifts:
Amplifiers from Athens
A
lex Arion writes that his only
regret is discovering audioXpress in “. . . the autumn of
[his] life.” We’re sorry too,
since his odyssey in audio construction appears to be exceptionally interesting. He began life, including
his audio activities and his engineering education, in Bucharest, Romania. Today, Alex lives and works in
Athens, Greece, where he plies a
busy trade in exotic amplifiers, both
tubed and solid-state under the firm
name of Arco Sound. He makes ex-
PHOTO 1: Single-ended, stereo 3.5W/channel using 2A3s
for output, driven by 6SL7s.
PHOTO 2: Stereo 2 × 40W power amp using 211 outputs driven by 6L6 in triode mode, plus 6SN7 inputs and an interstage
transformer.
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May 2007
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tensive use of wood and wood veneers
in his productions. We include a few
samples.---ETD
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You may reach Mr. Arion at Arco
Sound, 13 Koritsas Str. 14561 Kifissia,
Athens, Greece. 001 310 210 8083008.
PHOTO 3: Two 180W/
channel stereo power
amps using six EL34s,
driven by 6SN7s and
ECC81s and classically
supported, of course.
PHOTO 4: A hybrid
two-channel, 240W
amplifier with an
integral tube preamp.
May 2007
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3/22/2007 3:49:14 PM
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Cakepan Chassis
Have your cake and chassis, too!
By Ken Bird
B
uilding tube amplifiers has always required a good sturdy
chassis to handle the heav y
transformers and necessar y
drilling and hole punching for tube
sockets and other components. The solution was always a Bud or Hammond
chassis from the local parts jobber or
mail-order house. Electronic parts jobbers have gone the way of the steam
locomotive, while mail order is more
expensive.
I found an ideal source of aluminum
chassis at www.wilton.com, which is a supplier of baking accessories for cake baking professionals and home based cake
decorators. Their line of square baking pans, which are perfect for even the
largest power amplifier chassis, is sold
through local dealers found in most cit-
PHOTO 1: Pans ready to be put to a good use.
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ies, and the variety of sizes will meet your
most demanding chassis needs. These
units are made of heavy ¹⁄₁₆ aluminum
construction but punch and drill easily,
and have flanges for attaching bottom
covers or feet if desired. After cleaning
and applying a coat of auto primer, you
can paint them or polish them to their
natural finish. Photo 1 shows two different sizes of the square pans in their “raw”
format. Photo 2 shows the amp chassis.
The black chassis (left) is an 8˝ × 8˝ × 2˝
pan with a dual channel SE amp using a
6L6 in each output stage, and the “Red
Hot Amp” (right) is built on a 12˝ × 12˝
× 3˝ and is a dual channel PP amp using
a dual triode 6BX7 in the output stage.
The pans offer a lot of real estate for the
builder, and soon they will be baking up
some good sound in my workshop. aX
PHOTO 2: Amp chassis.
May 2007
3/22/2007 3:19:57 PM
sound solutions
By Ed Simon
Speech Intelligibility – Part 3
The author concludes his series with a look at problems—and solutions—
associated with reverb, resonances, and reflections.
PHOTO 3: Treatment too thin.
A
classic method of measuring reverb is to clap two boards together and use a stopwatch to
time how long the sound takes
to decay. The better method of the day
was to sound an organ pipe and time
its decay. More sophisticated measures,
of course, evolved as technologies advanced. Much of the early work was at
500 or 512Hz, which is still the design
center frequency for music; for speech
2000Hz is preferable. The modern version of this is a small sound system,
such as the “Sound Strobe” (aX 3/06).
Now you can just twiddle a knob to get
the impulse you need for listening tests.
REVERB PROBLEMS
Most of the computer-based audio measurement systems will also give a reverb
time number or graph. They typically
derive the reverb time from other kinds
of test signals. When you use computerbased measurement systems to do this,
you encounter that important warning:
“To err is human, to really foul things up
requires a computer!” (Anon.)
The reverberant field decay is usually
presented as a small matrix or graphs
of the actual frequency and the time it
takes to decay by 60dB. A typical reverb
time meter such as the Goldline GL60
model uses octave bands from 125Hz4000Hz. It is different in that it measures only the first 20dB of decay, then
presents the number as though it were
60dB of decay.
That is because when I designed it, I
was aiming for 20dB of signal-to-noise
ratio to ensure good speech intelligibility. I knew then that it was the first
drop in the sound that affected this. The
60dB number is still the classic reference, so that was my way of presenting
useful data to the less experienced user.
The meter also has the “bad” habit of
not settling on a single reverb number
if the field does not have a uniform rate
of decay.
The model of speech I use is a burst
of sound followed by silence for twice as
long. I had a recording of Andrew Carnegie speaking at the turn of the century
in a large reverberant room. He spoke
slowly by today’s standards—about three
syllables per second. A fast talker today
will speak six. For my model I use a rectangular gate that is open five times per
second. Thus the on time is ¹⁄₁₅ of a second and the off time is ²⁄₁₅ of a second.
Others who have studied actual speech
use different time intervals. As wrong as
they may be, my numbers work for me!
You can use a meter such as the Goldline to determine not only a matrix of
reverb numbers, but also how reverb
changes in a given room. The manual
(not mine) suggests you use a pink noise
source with the meter. The sound directly from the source drops off as you
move farther away. The reverberant field
in a room will continue to build until the
absorption and leakage out of the room
equal the rate at which energy is added.
This is the steady-state condition.
Turning off the noise while you are
far away from the sound source will trigger the meter and give you a classic reverb time number. Just remember, this
is for only the first 20dB of the curve.
There are some cases in which this does
not happen. The meter bounces all over
and does not settle on a single number,
which is a strong indication that the
room is not linear and there may be specific echoes that interfere with speech
intelligibility. The room is most likely
not very music friendly either.
If you produce a sound impulse right
next to the meter, the number it computes will be much lower than if you
repeat the experiment with the meter
across the room. If you use a Sound
Strobe or other impulse generator
SimonPt3-2761-2.indd 45
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3/22/2007 4:20:54 PM
through an existing or portable sound
system and move away with the reverb
time meter, at some point the meter or
your calibrated ears will indicate a reverb
time of 1.5 seconds at 2000Hz. Note
this distance, which is most likely the
same distance you would get from the
speech articulation tests for an acceptable loss distance.
Assuming that you speak at five syllables per second and you require about
5dB of decay before the next syllable to
get acceptable understanding, then you
would like to see a decay rate of 25dB in
two-thirds of a second. The other third
of a second is your sounding time. This
gives you a classic good speech reverb
time of 1.44 seconds, basically what the
meter just showed you.
CRITICAL DISTANCE
As you move away from the constant
sound source, the level directly from the
source decreases. The level of the rever-
FIGURE 2: Post absorption reverb.
FIGURE 1: Pre-absorption reverb.
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berant field for a steady source will reach
a constant level throughout a room. The
distance at which the direct field equals
the reverberant field is called the critical
distance, which you can measure with a
constant noise source and a sound level
meter.
Start at the farthest point and walk
toward the sound source. You are at the
critical distance when the level rises
by 3dB. With some practice you can
get close with just your ears and not-
�
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by John Linsley Hood
NEW REPRINT OF THE 2ND EDITION. Starting with lucid descriptions of the parts involved
(resistors, capacitors, inductors, tubes, and transistors), the author then begins joining them together
to form circuits offering the reader an accessible approach to linear electronics design. Focusing mainly
on audio amplifiers, the book features two of Linsley Hood’s classic designs which have become popular
with the DIY crowd. Includes excellent coverage of amplifiers,
oscillators, power supplies, filters, and feedback using clear examples
and easily solved equations. Reprinted 2006, 1998, 348pp., 7 3/8” × 9
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“For many years ‘do-it-yourself’ audio enthusiasts have asked me,
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audioXpress 5/07
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3/22/2007 4:21:02 PM
ing where the volume begins to increase. The rise
means the sound level has
doubled. One-half is the
direct field, the other half
is the reverberant field.
In a really bad room this
distance should be about
the same as your speech
articulation loss distance.
That is why when
you move away from the
pulsed noise the reverb
time meter shows agreement with the articulation
tests. This is the maximum distance for unaided acceptable speech in
the room. This will be at
least the critical distance.
At the critical distance
the signal-to-noise ratio
is by definition 0dB for
continuous sounds.
Because we speak in
bursts, the reverberant
field will not build to the
same level as for continuous sound. With the model I use onethird on, two-thirds off—the ratio at the
critical distance is at least 4.8dB signal
to 0dB noise. This would be the worst
case with an almost infinite reverb time.
If you know the reverb time is less
than infinite, you can calculate how
much decay will occur between syllables
and add that to the signal-to-noise ratio.
You also know that the talkers’ speech
drops off with distance squared. You can
now calculate where the signal-to-noise
level drops to 5dB in your room. You
may wish to raise this goal depending on
the expected use of the room.
The critical distance can change if you
approach from different directions when
you measure the critical distance with
a pulsed noise source. It is possible that
this is due to room geometry or construction producing a nonuniform reverberant field. The most common cause
is that the sound source has directional
characteristics.
You can use this ability of loudspeakers to be more directional than a plain
human talking to your advantage. The
idea is to put sound on the audience only
where it will be absorbed. That way you
Altitude 3500
Integrated Valve Amplifier
PHOTO 4:
Round trap
in use.
do not feed energy into the reverberant
field. The directional loudspeakers will
increase the effective speech intelligibility distance. It is possible to deliberately
design very directional loudspeakers.
One theory states that you can make
a sound system overcome bad acoustics. As some ancients believed, the gods
made us vain so they can enjoy the fall!
You can make some improvements—
horrible to bad or bad to okay, for example. But horrible to good will just not
happen with loudspeakers alone. You
must improve the room.
One of humans’ annoying traits is our
preference for simple answers. Is the
room good or bad? Exactly what does it
take to be good? In this case, you could
hire one of my acoustician friends, pay
them lots of money, follow their advice,
pay attention to the details, and you will
have a good room. But for those of you
who want to know more. . .
A CHURCH STORY
The Altitude 3500 is the
culmination of many years of
development and refinement by
Fountek Electronics.
Component
quality and circuit design have been
carefully chosen to deliver the highest
quality of musical reproduction, with
accurate and emotional delivery of the
sound stage.
The Altitude 3500 Integrated valve
amplifier features only the best
available components with the shortest
and cleanest signal paths possible with
direct coupling used on the input stages
to improve signal transit response.
Specifications:
Channels
Inputs
Output Power
Output Class
Output
T.H.D.
S/N Ration
Valves
2-channel
CD/Tuner/Aux1/Aux2
32WPC@1KHz /8ohm
AB1 Push Pull
4 or 8 ohm speaker load
less than 1%
89dB/A
2 x 12AT7 Twin triode
2 x 12BH7 Twin triode
4 x EL34B Pentode
Chassis
heavy gauge brushed
aluminum on all sides
Dimensions
350mm*190mm*320mm
Net Weight
18Kgs (39.7lbs)
Conformity
CE Rating
$1150.00
Introductory Price
A local large church had a problem with
speech not being well understood after a
recent building renovation. The church,
as far as I could tell, was originally built
47
3/26/2007 9:06:18 AM
with an interior finish of plaster and
stone. When this proved to be far too
reverberant, the interior plaster ceilings
were sprayed with a sound-absorbing
plaster and the main ceiling covered with
absorbing panels. This left the room adequate to good for speech but very unpleasant for music.
When 30 years had passed and no
one left could remember the original
problems, the room was renovated to
improve the musicality of the room. No
acoustician was consulted. The designer
plastered over the absorbent tile ceiling and painted the absorbent plaster.
Now the room had lots of reverb and a
few folks incorrectly thought it was improved, but it was now good for neither
speech nor music. In addition, the room
was much noisier.
The noise problem involved an organ
blower chamber isolated from the remodeled room by a masonry wall except
where it had been cut to accept a heat-
ing radiator. Only a piece of sheet metal
blocked the visible opening. Closing the
opening with a double layer of drywall
and sealing reduced the noise.
Many folks proposed installing large
and very expensive loudspeakers to project speech the length of the room. Three
different systems were tried and none
proved adequate. I had a few ideas regarding the problem, so I contacted Sam
Berkow (Sia Acoustics-Jazz at Lincoln
Center, Hollywood Bowl, and Smaart
Software) to visit the church. He took
measurements using Smaart software
and determined the major problem was
that all the sound energy was not decaying uniformly. In fact, the energy at
the 250Hz octave band was decaying so
slowly that it was masking speech in the
room. In addition, he thought that the
curved roofs on the transepts (side seating areas) were the primary cause.
He suggested adding enough absorption to drop the reverb time of the
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48
audioXpress 5/07
T
he decibel is one of those wonderful bits of jargon that we use
to confuse the uninitiated. Please do
not read further unless you already
know the secret sign and password.
The base unit is the bel, which is the
logarithm base 10 of the ratio of the
sound being measured to the threshold
of hearing. Thus, if the sound energy is
ten times louder than our threshold of
hearing it is log(10/1)=1. If the sound
is 100 times louder, it is 2 bels, 1000
times louder is 3 bels, and so on. Just
count the number of zeroes after the 1.
You can hear over a range of greater
than 12 bels (120 decibels), which,
when written out, is 1,000,000,000,000
to 1. Writing 120dB is much easier.
The decibel becomes confusing
when you have, for instance, only twice
the energy. The math works out to 10
(to convert from bels to decibels) ×
logarithm (2/1). Use a calculator to
do the logarithm, which works out to
a two-fold increase, or 3.010299957. .
. decibels, which you can round off to
3. With ten times as much energy (1
bel), you get 10 decibels. If you double
the sounds’ energy again, it cannot be
20 decibels because that is 100 times
as loud.
You now know the secret of logarithms: add the 3 from the doubling
to the 10 from the ten-fold increase to
determine that the energy increases by
13dB. Thus, 40 times as much energy
would be 3 + 3 + 10, or 16dB. Eighty
times, of course, is 3 + 3 + 3 + 10, or
19dB.
If you know that 20dB is 100 times
and 19dB is 80 times, then what is
1dB? Well, 100 divided by 80 is 1.25.
The log of 1.25 is .0969100. Ten
times that is close enough to 1 that
you should see how this works. The
precise answer is 10 to the exponent
(1dB/10 decibels to bel), which equals
1.258925412. . ., or 1.
Another consideration is the accuracy to which you can measure. It is
fine measuring a voltage on a digital
meter that is accurate to four digits.
A change in barometric pressure of 1˝
will change some sound levels by about
.2dB. Wind and air movement also affects the readings. That is why sound
pressure levels are rarely used with a
resolution of greater than 1.0dB.—ES
3/22/2007 4:21:09 PM
room to 1.5 seconds with a reasonable
crowd; in particular, putting absorbers
in the transepts. As soon as we added
two absorbers to a transept, it became
immediately clear that he had identified the source of the resonance. Treating both transepts and adding material
along the side walkway ceilings provided
the acoustic treatment needed to change
the room from very bad to good. In addition, because of the location of the
absorbers, the aesthetics of the church
remained unchanged. The moral of the
story—find out exactly what the problem is before trying to fix it.
RESONANCES
Any two parallel walls will bounce sound
back and forth. Most rooms have at least
three pairs of such surfaces. If the walls
are good reflectors, this will show up at
the frequencies which have a half wavelength that is a multiple of the spacing
distance. There are other kinds of resonators besides parallel surfaces.
It is easy to understand that a wind
instrument such as a flute uses the air
motion inside it to produce a tone.
Sound travels at a speed of about 1132´
per second. If you have the air bouncing
between two acoustic nonlinearities, it
will form a standing wave. Even random
air motions will excite the wave and add
energy to it as it bounces back and forth.
The other common form of resonator
is like a whistle in which the air rolls
around in a circle and reinforces the resulting sound wave.
All rooms have many resonances. So
the concern involves resonances that
occur at particular frequencies to which
we are especially sensitive or are very
strong.
Finding out you have a problem resonance is simple. It will show up as a
frequency or band of frequencies that do
not decay as rapidly as the surrounding
bands of frequencies. You can measure
this with test equipment or use a musical
instrument such as a piano or organ. You
just listen for notes that hang around too
long. This is sometimes called “color.”
Figuring out which part of the room
causes the problem resonance is a bit
harder. Once you are able to produce
or measure the resonance on demand,
you can move a large absorber around
until you find out where it has the most
49
3/22/2007 4:21:12 PM
effect. Sometimes you will need to use
many absorbers and cover most of a wall
to produce a noticeable effect. The difficult times occur when you have a bell
tower that may be causing problems and
you want to try material on the ceiling
120´ above you!
This may sound difficult, and it is.
Finding out the cause of a problem is
more art than science. Although there
are some instruments that help, they are
not common or readily available. The
good news is that from my experience,
resonances that cause a real problem
with speech intelligibility are present
in about 10% of the rooms I examine.
Many small resonances at the right frequencies can enhance music.
The common mistake in trying to fix
a resonance is to place foam on the walls
until the problem goes away. If the foam
is not thick enough to absorb the frequencies of interest, it is not doing what
you want and is just killing the mids and
highs. This also colors the sound. If you
want to use movable panels, be sure they
are really big and thick.
Once you have identified the area that
causes the resonance, you can add just
enough absorption to damp it. This is
different than trying to add absorption
to get a desired reverb time. Because of
the nature of resonance, the absorption
has twice as much or more effect than it
would if it were just reverb.
A second approach would be to
change the surface. Building out, adding
angled surfaces, or features of interest
such as reliefs, or even using shelving to
break up the characteristics that allow a
single frequency to build are often used.
If you can’t absorb it, bounce it around
enough so that it spreads out and dissipates.
REFLECTIONS
Of course, by now you are thinking, this
is nuts. “How can this be right if you
don’t need to use a computer or any ludicrously expensive test gear?”
That’s right! With simple measurements and calculations, you can achieve
good reverb times in a room with nonuniform absorption and nonlinear reverb
time curve. The only equipment you
need is a pair of ears, paper, and a pencil.
A calculator helps.
Of course, sometimes this is not
50
audioXpress 5/07
enough. You have until now been looking at statistical averages of the sound
field. Another problem area to address
involves echoes or sharp distinct reflections that are not muddled in the statistical average.
If you add bookcases to the side and
front walls, a small room would sound
much better even with the same reverb
time. The decay curve would now be
much smoother due to the diffusion you
have added.
Helmut Haas (1958) determined that
a single echo can reinforce speech intelligibility or destroy it. He concluded that
if the repetition of a sound comes within
30ms, you perceive it as part of the original sound, perhaps adding a bit of spaciousness or volume to the original. The
second sound must be 10dB louder than
the first to be perceived at all.
When the second sound comes after
this period, it disturbs the speech. The
effect is greater for faster speech. Some
forms of late energy can even destroy
speech intelligibility. Increasing the volume of the speech does not change this
effect because the echo also increases.
The direction of the echo does not have
a great effect on sounds that originate
from the front.
As a demonstration, I ask two singers
to sing a capella the same piece while
walking away from each other. When
they get to around 30–35´ apart, they
find they cannot sing together. There is
a slight variance due to the tempo of the
piece.
If you have ever tried Sound Strobe
on a loudspeaker that has a path length
difference at the crossover frequency,
under some conditions you can hear a
distinct double click even though the
path length difference may be less than a
foot. Haas used speech for his model; it
does not encompass dissimilar sounds or
frequency effects. It does work great for
speech reinforcement design if you do
not act silly trying to stretch the results
of his work.
RAY TRACING
When you try to predict what the echo
pattern of the sound will be like, you use
the ray tracing model of sound dispersion. Simply put, you draw arrows pointing out from the sound source. When
the rays strike a wall, ceiling, or other
surface, you draw them bouncing onward as though they hit a mirror. Continue this path until the rays (arrows)
reach where the listener(s) should be.
You can then assume that the length
of the ray is directly proportional to the
time it takes the sound impulse to reach
the listener. It is useful to plot the first
direct ray, the second rays that bounce
once off a wall or ceiling, and the third
rays that bounce off two surfaces. This
is usually enough to get an idea of what
the ray response will be like. You can do
more if you like, but the task becomes
almost impossible for five to seven
bounces.
You can also scale the amplitude of
the sound rays by multiplying the inverse square law value by the absorption coefficient of the surface materials.
Sometimes the reflecting surface is small
enough or angled so that it reflects only
part of the ray and spreads out part of it.
The wavelength or frequency of interest
determines what size of non-uniformity
becomes effective.
Sometimes the room is too small to
use the ray simplification at the lower
frequencies. I stop using the ray model
if the room is too small to allow seven
wavelengths of the frequency of interest.
This allows you to produce a graph of
amplitude versus time.
If you want to cheat a bit, you can also
plot the reverb time curve on the same
scale. Because the reverb time is usually
based on steady-state energy excitation
and the ray tracing assumes a unit impulse, you may wish to move the reverb
time curve down a bit on the X axis. If
you plot enough ray bounces, you will
find it merges into a giant jumble that
about matches the classic reverb time
and that is where you can merge the
curves. Computer programs are useful
for this, but you can still do this by hand!
All the unit spikes up to 30ms are
combined as though they are one great
sound. Some folks combine the rest
of the mess for a second value. When
the first value is 5–10dB greater than
the second, speech intelligibility will
be good. The exact value is influenced
by your exact methodology, the noise
level, a few other semi-mythical issues,
and the intended audience.
When the combination curve shows a
nice drop of about 5–7dB for the initial
3/22/2007 4:21:12 PM
break and a smooth linear tail without
any major jaggies, you probably have a
good room for music. When the reflections coming in past 30ms are stronger
than desired, you can add absorption to
the reflecting surfaces. One mistake is to
treat one sidewall and not the other. The
reflections off the sidewalls contribute to
the spaciousness of a room. If sidewall
treatment is needed, use it sparingly and
split it up onto both walls and stagger it.
Sometimes this will decrease the overall
reverb time too much.
The trick, then, is to use a diffusing
surface—anything from columns along
a wall, moldings, or even strips of absorbent material. This will spread out the
reflection, decreasing its level and adding
to the reverberant tail.
Sometimes you wish to reinforce the
early energy of the first 30ms, which
tends to improve the intimacy of a
space. This is where a low ceiling angle
to maximize projection to the audience
works wonders. I once was involved in a
project with acoustician David Klepper,
who designed a glass reflector system
that was on a remote control. At the first
use Mimi Lerner was the soloist.
I did not like the sound and asked her
what she thought. She was diplomatic
Mouser
but agreed to sing a very brief bit (conserving her voice for the second performance). I lowered the reflectors by about
3´ and heard an amazing difference as
the path length locked into the magic
30´. The second performance was much
better!
In the 1940s studies of movie theaters
surprisingly concluded that some of the
“good” examples had worse reproduction
system frequency response than some
of the “bad” examples. An excellent example that the nature of the sound reflections, reverb, and noise can be more
important than just “fidelity.”
One question that comes up from
time to time is, “If we know so much
these days, how come all the old concert
halls are good halls and many newer
halls are not as good?” The answer is
simple: When you build a bad concert
hall you either fix it or tear it down
and build another one! I hope you have
picked up enough useful information
to improve the acoustic spaces you care
about and will not need to resort to the
wrecking ball! aX
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51
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TONEREX, Nichicon Fine Gold Muse,
KZ, Jensen Four-pole capacitors Stepped
attenuators, Teflon cables, Connectors
Custom designs for OEM customers
www.borbelyaudio.com
Selected BORBELY AUDIO kits in Japan:
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transformers made by Trafomatic.
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Lots of audio information on our website.
Please visit WHAT’S NEW on our homepage,
www.tdl-tech.com (505-382-3173)
stereo gear old/new/quality, speakers,
amps, preamps, turntables, reel to reels,
vacuum tubes, tube testers, guitars, parts,
etc. 850-314-0321
Email: [email protected]
What’s New
on the aX website?
Web-exclusive content:
Word Test files from Ed Simon’s “Speech
Intelligibility, Part 1” (aX 3/07).
“Modifying Mighty Mouse,” By Bob McIntyre
(aX 3/07).
Jesse W. Knight’s review of Sony CD/DVD
Player Model DVP-NS55P (aX 1/07).
Articles from past issues:
“A Phase Meter Calibrator,” By Charles
Hansen (aX 11/06).
Dan Ferguson’s review of Installer car stereo
installation software (aX 12/06).
“Low-Level Analog Switching,” By Dennis
Hoffman (aX 1/07).
“Grounding and System Interfacing,” By
Gary Galo (aX 1/07).
Dennis Colin’s review of The Art of Linear
Electronics (aX 1/07).
“How Lound Is Real?” By Larry Klein (aX
2/07).
Yard
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Luther Wright
16901 Parksdale Road
Petersburg, VA 23805
“Transferring LPs to DVDs in High
Resolution,” By Victor Staggs (aX 2/07).
“The Rocky Mountain Audio Fest,” By Bob
Cordell (aX 2/07).
Dennis Colin’s review of High Performance
Loudspeakers Sixth Edition [plus table of
contents] (aX 2/07).
“A Prototyping System for Passive
Crossovers,” By Ramkumar Ramaswamy (aX
3/07).
David Milford’s review of PNF Audio Cables
(aX 3/07).
“A Low-Noise Measurement Preamp,” By
Dennis Colin (aX 4/07).
“Yard Sale” is published in each issue of
aX. For guidelines on how subscribers
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audioXpress 5/07
Classified-Ad Index507.indd 52
For information on these and others, visit
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check out the links to our other magazines,
Voice Coil and Multi Media Manufacturer!
3/22/2007 3:58:33 PM
review
By Tom Perazella
Revue Du Son
Test CD Number 17
the 17th in a series that started in October
1993. The music is highlighted by female
and male vocals (soloists and groups), old
woodwind instruments, strings, piano,
orchestra, organ, bells, large drums, and
gongs. Tracks 16, 21, and 22 are not musical, but include applause, street sounds,
and the killer helicopter.
Track 1: The Magic of
Kasarova—“Deh! Tu, Bell’anima”
This track opens with very smooth woodwinds and strings. The vocalist is mezzosoprano Vesselina Kasarova, whose dynamic range is very large. Your speakers
must be able to handle very sotto-voce
inflections all the way to huge peaks.
My Tenma sound level meter measured from 61-to-88dBsplA. If you have
any midrange or tweeter peaks or distortions, this recording will sound very
harsh. On a good system it will sound
hugely powerful and engrossing.
Track 2: Kari Bremnes: Svarta Bjorn—
“Sangen om fyret ved Tornehamn”
Close-miking can make recordings sound
in-your-face, especially with mikes that
have a presence peak. However, done
properly it can impart a sense of immediacy and detail. This track walks the fine
line in between.
The voice is very clean and detailed
without being overly bright. It is a great
test piece because there are some highlevel low-frequency drums in the background that center around 40Hz and hit
peaks of 102dBsplC, while the average is
78dBsplA. Anyone for an IM distortion
test? A small two-way speaker will be
hard-pressed to handle the high low-frequency levels while simultaneously keeping the voices clean.
T
est CDs fill a basic need to evaluate system and room performance. They come in various
f lavors—some with lots of test
signals, others with mostly music. Each
has its place.
A CD with properly designed and recorded/produced signals can save a lot
of money otherwise spent on specialized
test equipment. However, at the end of
the day it’s all about music, so picking
recordings that can reveal your system’s
performance in various categories is invaluable.
There is no shortage of musical test
CDs. I have quite a few and use them
extensively. Some are commercially available, others were custom made.
A test CD should have a wide variety
of music to test frequency response, transient response, dynamic range, freedom
from distortion, and spatial characteristics. By the way, it should be listenable. If
the music sounds like test tones, I’d rather
use test tones.
Test CD No. 17 was produced by the
French magazine Revue Du Son. Usually I
first read the track descriptions to get an
idea of what to expect. In this case, I not
only had liner notes, but also an article in
the January/February 2005 Revue Du Son
that gave more complete descriptions.
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audioXpress 5/07
Perazella2803.indd 54
PHOTO 1: Revue du Son.
Great! Well, maybe not. Everything is
in French, which I can’t read. I’ve done
blind equipment tests, but this is my first
blind CD test. I would like to have provided insight from the magazine descriptions, but not being able to read them
certainly minimized any listener bias.
My response to a quick run-through
was quite positive. The CD has 22 tracks
and runs a total of 71:26. Deciphering
what I could from the notes written by
Jean Hiraga, the disc was mastered at Le
Studio Acoustique de Passavant and is
Track 3: Kari Bremnes:
Svarta Bjorn—“Byssan lull”
Strong, but not very deep-bass percussion opens this track. Peaks of around
95dBsplC centered around 50Hz provide
a backdrop for more smooth vocals. The
vocals are clear and separate from the
background, resulting in a very intimate
feeling. There are also some delicately
struck bells that float above the vocals.
At the end of the cut, Kari’s voice
evolves to what sounds like quiet synthesized surf. This is a good test of separation of a clean vocal from a strong per-
3/22/2007 4:19:24 PM
cussive background.
Track 4: Trésors Moyen-Age: Musique
sacrée et Profane ensemble Sequentia—
“Bonne amourette me tient gai”
This is a short vocal track that was recorded in a hall with good acoustics. A
solo voice provides the chance to hear
its definition along with hall reflections.
Other voices join, adding to the complexity and testing the ability of your system
to separate the voices and their harmonics not only from each other, but from
the hall reverberance. Each voice should
sound rich and spacious, yet distinct.
sacrée et Profane ensemble Peceval—
“Aisso es viadera”
Ancient instruments add distinctive flavors to this track. The opening woodwinds have a strong reedy sound. A female vocal enters and is accompanied by
background percussion. The separation of
the voice from the percussion is excellent.
sacrée et Profane ensemble Peceval—
“Ja nulls om pres”
The opening strings are quite distinct
with well-defined plucking and delicate
harmonics. A male voice with realistic
tonal balance and clear vibrato separates cleanly from the accompaniment.
Sibilants are well controlled. A flute enters solo and is later added to the mix.
During the vocals it should be audible,
though in the background.
Track 7: J. Brahms: Sonate Nr 1 op 1
pour piano, M.J. Jude—“Scherzo”
This is an OK piano recording that shows
good spatial and tonal balance. Harmonics are well presented. However, I have
heard recordings with a much greater
sense of dynamics. It’s a good recording,
but would not be my primary reference.
Track 8: Stefano Bollani:
Smat-smat—“La Vita Intensa”
This is a good track to show how demanding a piano can be to reproduce. It
is quite dynamic with excellent hammer
sounds and natural resonances from the
lower strings. Being able to separate the
chords’ harmonics while sounding tonally
balanced is a definite challenge. All this
happens in an atmosphere of transients
and rhythms that can easily lead to a
sense of musical confusion if your system
cannot resolve all that is going on. It may
not be everyone’s choice in music, but it
is a great test piece.
Track 9: Chopin: Ballades et Scherzos,
A. Rubinstein—“Scherzo nr 2 op 31,
Siminear”
This piano piece is all about dynamics,
transients, high- and low-level detail, and
room acoustics. There is a good sense
of the instrument while still having hall
ambience that is not overly reverberant.
Attacks are good. Low-level detail is excellent.
Track 10: Orchestre de Contrebasses:
CH. Genet—“Bass, Bass, Bass, Bass,
Bass & Bass”
As you might gather from the title, this is
all about bass. And bass there is, plucked,
bowed, and struck—string bass heaven!
The detail is extraordinary. Harmonics
extend to all frequency ranges. This is
clean bass with enough higher frequencies to add spice. It is a great test of the
ability to separate transients from the
background bass lines.
I ran it through my Behringer
DEQ2496 RTA and found some of the
bass line centered around 28Hz. There is
quite a bit of energy all across the audio
band with an interesting change from
bass to midrange at one point. At a little
over three minutes into the track several
voices appear that move around the stage.
It’s an entertaining change of pace. The
piece ends with a sharp impact.
Track 11: Vivaldi: Concerto nr 2, La
Stravaganza, C. Todorovski—“Largo”
As with a piano, it is often very difficult
to make an organ recording that really
sounds right. This one does justice to the
instrument—a complex, powerful, subtle
combination of tones that separates the
organ from all other instruments. Balance is very good with a sense of position
that is neither too close nor too far. The
hall sounds believable.
Upper registers are clean with good
separation of upper-midrange and treble,
especially in the presence of bass. The
notes seem to float above the ambience.
This is not a fireworks piece, but a believable simulation of being with the instrument.
Track 12: Beethoven: Concerto pour
violon, Heifetz/Munch—“Larghetto”
Strings, beautiful strings. How they can
make your spirit soar. Though if reproduced badly, they can send you running
for the door.
Here they are done well. The strings
are very smooth without sounding soggy.
Ambience adds to the realism. The violin is clean and well defined. Massed
strings sound like massed strings, not
some fur-ball of strident sound. Individual harmonics are easily heard. The
violin’s intonation, bow sounds, and vibrato are excellent and well separated
from background instruments. Plucked
strings are believable.
There is a lot going on in this recording. It is an excellent test of definition in
which the result is a composite rich in
texture rather than a lot of impressive,
individually dramatic sounds. Detail is
important to producing a realistic image
even if not accompanied by striking dynamics. This is an excellent piece to see
how well your system performs. It’s either right or the magic will be gone.
Quad speaker owners will love this track.
Track 13: Rachmaninoff: Concertos
Nr 3 pour piano, Horowitz/Ormandy
—“Allegro ma non tanto”
I’m torn over this track. As I review my
notes, I see the phrase “can’t put my finger on the problem.” There is something
about this recording that I don’t like.
The sound of the solo piano is good
and clean, except for weakness in the
lower register. However, when listening over speakers or headphones, there
seems to be some glare when it is going
full tilt with other instruments. Dynamic
range also does not seem to be as wide
as other piano recordings. This is the
weakest track as a reference recording. If
not in such strong company as the other
tracks, it might be considered very good,
but it pales in comparison.
Track 14: CD Test Nr 7: “Presentation
des Grandes Orgues de Saint Eustache
à Paris par Jean Guillou”
This is another great organ recording,
not only because of the instrument, but
also because of the hall and the ability
to hear a male voice so clearly and with
great intonation. Ambience is excellent.
The different voices of the organ are
55
3/22/2007 4:19:26 PM
explained with short excerpts following
each spoken section.
There is first-rate definition in a highly
reverberant space, displaying good highfrequency clarity and tremolo. The low
pedals provide a nice rumble. The article
mentions a stop capable of producing a
16Hz fundamental, but if it is there, it is
at a low level.
Never fear, this track is not about earth
shaking bass, but rather about definition
and ambience. The house shaker comes
in track 22. Enjoy this one for its realism.
Track 15: CD Test Nr 10: “clochettes”
The bells are an extreme test of the highfrequency system performance. It is ripe
with harmonics that can easily smear together into high-frequency oatmeal. The
strikes produce strong transients and the
decay sounds natural with gobs of harmonics and beat frequencies. Good luck
if you have tweeter or crossover problems.
Track 16: CD Test Nr 10:
“applaudissements”
The first of the non-musical tracks, this
one features almost three minutes of applause in a large hall, with an ability to
identify individuals yelling at different
distances from the mikes. It is a very
complex sound that will lose its realism
if the playback system is lacking in any
critical aspect.
At a little over a minute into the track
there is so much going on that the sound
almost produces pink noise. However, through it all, the handclaps have
a sharp attack, the voices are real and
distinct, the hall size palpable, and the
enthusiasm of the audience apparent.
This should not sound overly sharp or
muted. Pay special attention to how the
voices separate from the applause.
Track 17: Percussions XX—CD Test
Nr 14: “Appendice alla perfezione”
Delicate bells introduce this track. While
the level is low, the attack and decay
should be quite clear. The intensity increases and the sound, especially the attacks, should become louder without
becoming harsh.
With the harder strikes, note the increase in the ratio of the initial impact to
the decay. There is extended harmonic
content. You should also hear movement
across the stage. As the track ends, there
56
audioXpress 5/0
5/077
are complex sounds at high volume. Regardless of the volume, they should always sound like bells.
Track 18: Percussions XX—CD Test
Nr 14: “Towards”
Drums, cymbals, gongs! A friend once
said I would stop while walking down
the street to listen to a garbage can fall
down a flight of stairs. I guess that is because highly dynamic, wide-bandwidth
sounds can tell a lot about what is right
and wrong with your system.
This track is a great broadband transient test. I played it back at average levels around 100dBspl and peaks as high
as 111dBspl, where everything must be
right. Your speakers need a lot of linear
displacement at all frequencies, you need
enough clean power to avoid clipping
the peaks, and a good radiation pattern
will help it sound realistic. Use caution
when playing this track because it can
damage wimpy speakers, and your hearing if you play it for extended periods.
Track 19: CD Test Nr 10: “Gongs”
This is the gong show. They can really
sound nasty if you have problems anywhere in the frequency band. The transients are brutal. There is energy over
much of the audio spectrum. Harmonics
are abundant.
Check out the levels of the strikes at
23 and 32 seconds. Also check out the
shimmering effects later in the track. If
your system has good low-level resolution, you will hear voices close to the end.
“Grosse caisse”
Forget the dog, beware of the drum! This
track is brutal. There is a huge drumstrike impact. At my listening position,
I recorded a peak level of 115dBsplC. If
your speakers survive, listen for the detail
in the decay before the next hit.
“Bruits de rue et de moto”
This is one of only a few recordings I
have of an outdoor venue that sounds
real. It was made on a street and the
sounds are amazingly real in terms of the
image width and depth. The superimposition of the voices with the footsteps
sounds uncannily real.
At around 23 seconds there is some
kind of mechanical sound I cannot identify that is clearly separate from the rest
of the foot and voice traffic. The placement of different voices is excellent. At
around 1:06 a motorcycle starts up in
the background. It comes into the foreground moving to the left and across to
the right, with low and high frequencies
as the rider blips the throttle.
As with the earlier applause track, the
sound of footsteps is the delineating factor in the realism. The heel attacks are
crisp without being overly sharp, and you
can hear the slap of the soles as they hit
pavement. The natural voices float above
the foot sounds providing the frosting on
the cake. This is a very good test track.
“hélicoptère, decollage”
OK, you have been warned about other
tracks, but if you have not paid attention
before, do so now. This track is brutal. If
you value your speakers, please use restraint when setting the volume. I suggest
extra caution if you have a small two-way
speaker because it just won’t cut it. In
fact, it will be a total letdown; you will
probably hear nothing partway through
the cut except gobs of distortion and the
sound of your speakers self-destructing.
This track sneaks up on you because
it starts out pretty normal. You hear the
sounds of a turbine-powered helicopter preparing to lift off. The compressor
whine is realistic, plus the hiss characteristic of gobs of air being drawn into a jet
engine. The hiss becomes dominated by
the sound of the rotors.
If you cannot reproduce frequencies
below 20Hz, you will miss the action.
There are very high levels at low frequencies. To determine just how low, I ran the
signal from my preamp to my Tektronix
oscilloscope and recorded the waveform.
Figure 1 shows a large low-frequency
component around 19Hz.
This is serious low-frequency energy.
Even with my two very long-excursion
15˝ woofers, I started to get ugly noises
at 107dBspl. Death to small speakers!
I needed to reduce the peak levels to
around 105dBspl to stop the really serious uglies.
After running the tests, I consulted
the calibration chart for my sound level
meter and saw that the response at 20Hz
was 5dB down. Therefore, the sustained
3/26/2007 9:03:54 AM
level was really 110dBspl. The sub I had
built into my former house would have
been a better match for this track. It had
eight 12˝ drivers (see “True Bass,” 5/96
Speaker Builder). That is the kind of volume displacement necessary to reproduce
serious bass at high levels.
At around 2:55 there are some high
frequencies that are either the rotor tips
exceeding Mach 1 or the digital recording instruments exceeding 0dBFS. In any
case, this track is an interesting high-frequencies test if you are familiar with compressor whine. It is certainly a source of
very high levels of low frequencies from
the rotors. Small woofers need not apply.
SUMMARY
The real question is did it pass as a useful
musical test disk? The answer is definitely yes. There are enough different musical and non-musical but natural sounds
to provide a wide range of stress tests for
any system. More important, there are
many pieces that have sufficient content
quality to test not only whether it sounds
real, but real enough to elicit emotional
responses. Highly recommended! aX
FIGURE 1:
A large low-frequency
component around
19Hz.
Appendix
Home System:
CD player
Preamp
EQ
Crossover
Amps—high
Amps—mid
Amps—low
Speakers—mid/high
Speakers—mid bass
Speakers—sub bass
Portable System:
CD player
Headphone Amp
Headphones
Test Equipment:
Sound level meter
RTA
Oscilloscope
Sony 707ESD
Custom-built, based on AD797 op amps and
BUF03 output buffers
Behringer DEQ2496
Behringer DCX2496 (Linkwitz-Riley 48dB/
octave slopes at 71 and 303Hz)
One two-channel Crown Macro Reference amp
Three—AudioSource Amp3
Two—KG-5230 “plate” amp
Two—each consists of a Bohlender-Graebener
RD75 in a custom baffle. See “On Angel’s
Wings” in the January 2001 audioXpress.
Two—each consists of six Peerless 831727 10˝
woofers in a custom baffle. See “A Dipole
Midbass” in the June 2004 audioXpress.
Two—each consists of a Dayton 15˝ DVC
woofer in a 5ft3 sealed box
Panasonic SL-CT520
Headroom Total Airhead amp
Etymotic Research ER4S
Tenma 72-860, calibrated against an
ACO 7012 microphone
Behringer DEQ2496
Tektronix TDS210
57
3/22/2007 4:19:33 PM
XPRESSMail
LESS WAR, MORE PEACE
(OR VALVET AMPLIFIER)
Due to a slight mis-communication, I
didn’t see the final proofs of my article
“Amplifier War and Peace” about the
Valvet™ hybrid amplifier before printing (March ’07, p. 14). Thus, potential
builders should note that C7 is shown
in Fig. 1 as 47µF, but should be 0.47µF
(plastic film).
Also, if built as shown, there is a nasty
(but not fatal) “turnon thump” through
the speakers. This is due to the HT
turn-on upsetting the MOSFET biasing momentarily. The simple fix is to
add another step to the power switch, so
heaters, then HT, then MOSFETs are
progressively turned on.
Simon Brown
[email protected]
OSCILLATOR PARTS
Reading Dennis Colin’s “A Wide-Range
Audio Sweep Oscillator” (aX 2/07, p.
26), I found that he is, as usual, making
it difficult to obtain parts! Harris Semiconductor no longer exists as a separate
entity; they were eaten by Intersil, long
ago.
Allied Electronics used to carry this
part (under Harris), but no longer carry
Intersil. A google search turns up loads
of suppliers—in Hong Kong, and other
exotic places—bad news if you live here.
I did find one stocking distributor in the
US: www.1sourcecomponents.com. I have
requested a quote for small quantities.
We’ll see what happens.
John Nickerson
[email protected]
Dennis Colin responds:
My sincerest apology; I hadn’t known that
the CA3280AE wasn’t available. Digi-Key
has some Intersil parts, but not this one.
What a shame; this part is extremely versatile. I have six Harris units and one original
RCA; I’d better preserve them in a time
capsule!
Please let aX know if your search is successful. If not, and there’s no substitute,
I could try to design a discrete transistor
equivalent, or change the oscillating circuit
58
audioXpress 5/07
xpressmail-1.indd 58
to use analog multipliers. That could reduce
the frequency range because of voltage
offsets.
Thanks for your letter; in future projects
I’ll make sure all parts are available.
DVD/SACD PLAYERS
I decided to document my recent experiences with SACD/DVD player modifications. I first pursued this back in 2003
when I procured a Philips SACD1000
unit. I then had a lot of trouble obtaining technical information (tech service
manual) until I wrote your magazine,
detailing my experiences.
With help from your readers, I eventually did obtain a CD-ROM containing the information I wanted. However,
that’s “ancient history” right now—I was
diverted from that path by involvement
in other projects and put it aside. My
experience now is with a pair of cheap
players—Sony DVP-NS500V and the
oppo-digital DV-970 universal disc
player.
Back in 2003, I had also (easily and
quickly thanks to Sony’s professional
services) procured the tech manual for
the Sony unit. At that time, I did an
ultra simple (and simple-minded) revision, installing “pull-down” resistors
(4.32k, RN55D) at the outputs of the
main stereo output op amp and also bypassing the output coupling caps (47µF/
16V) with small polypropylene caps (the
well-known 50V Panasonic units available from Digi-Key). The audio output
channels in the Sony unit are all handled
by the cheap and ubiquitous 4558 dual
op amp.
The results were modest but apparent, and listening quality did improve
a bit. I put that aside for a while until I
remembered how useful the AD827 had
been in my previous projects. This fastsettling video op amp works very well as
a substitute for other unity-gain stable
devices in common audio circuits—it
has always outperformed the garden variety chips I have encountered in other
pieces of consumer gear (line-level applications).
Its one drawback is that it is not available in an 8-pin surface-mount package.
Luckily a friend came to the rescue and
mentioned the AD828, which is stable
down to a gain of 2 (no problem in these
players I am referring to) and is otherwise reasonably similar in performance
to the AD827. It is also cheaper and is
available in an 8-pin SOIC package
(surface-mount).
The temptation was too great. I tried
it out in the Sony DVP-S500V (IC202
position) and the results were what I’d
been dreaming about for SACD playback (and the 24/96 Classic Records
DAD discs—which are DVD-V in twochannel stereo).
All of the problems with the SACD
and the DAD were eliminated, including
the new “Living Stereo” hybrid SACD
series, the older Sony Classical SACDonly discs dating back to 1999/2000 and
later, and the original circa-1998 Classic
Records DAD discs. By comparison,
the playback results are thrilling. My
resources are very limited, so I don’t have
access to more than about 20 discs with
which to demonstrate these results; others will need to go further afield than I
can right now.
How do these hi-res discs compare to
CD? (My current CD reference is any
transport used with the heavily modified
Philips DAC960. . . work done on it in
1993 and in 1998—all documented in
the essential POOGE book by Jung and
Galo.) They are decisively superior—in
every way. And this is with an el-cheapo
(but now modified with the AD828)
SONY DVP-NS500V DVD/SACD/
CD player. It’s possibly the most dramatic transformation I’ve experienced in
several years.
My suspicions that SACD and DVDV (at 24/96) formats were capable of
outstanding results are now confirmed
(and almost nobody I know is aware of
this!). I am surprised and pleased that
my suspicions were correct. However,
beware of producers who issued SACD
discs made from CD masters ( John
Atkinson—and others?—have exposed
these charlatans in the past. . . and I can
tell you now, the differences are amazingly obvious).
I also performed this “trick” on my
3/22/2007 3:50:07 PM
oppo-digital DV-970 player (the main
stereo outputs only—and yes, it also uses
the 4558 chip too—perhaps everything
does). . . and with some trepidation, because I have no documentation for it.
Luckily a scope and the old CBS CD-1
test disc come in handy. The sonic improvements are similar, and are also obvious when playing-back a movie DVD
(PCM - stereo option). This is a good
all-round improvement, with good results for every application.
R. K. LeBeck. Jr.
Kirkland, Wash.
MATT HAMILTON—IN
MEMORIAM
In January 1994, I placed an ad in Speaker Builder and Audio Amateur. It brought
Matt Hamilton and me together. I wrote
about my relationship with Matt as my
mentor in the May 2004 issue of audioXpress (“Letters,” p. 68).
It is with great sadness and a profound
sense of personal loss that I report Matt’s
passing on October 29, 2006, in Bradenton, Fla., after a prolonged illness.
William Matthew Hamilton was born
on May 21, 1925, in Audra, W.Va. While
growing up, he and a friend built radios,
and never lost his love for electronics.
He graduated from Buckhannon Upshur
High School, W. Va., in 1943; attended
Fenn College in Cleveland, Ohio, and
married Wanita Waugh in 1947. He
worked as a mechanic on both automobiles and buses until 1955.
Matt then worked for the Thew
Shovel Company in Lorain, Ohio, in
the engineering department until 1959,
when he started working for the Burroughs Corporation. During the period
he worked for Burroughs, he assisted in
selling and installing many large computer systems in Ohio, Michigan, Wisconsin, Pennsylvania, California, and
also in Amsterdam, The Netherlands.
He returned to the US in 1971, retired
from Burroughs in 1986, and moved
from Pennsylvania to Bradenton, Fla.,
in 1988.
Sometime in the 1990s, he started
giving computers to academically motivated, needy children. He accepted computers and associated equipment from
individuals and companies, and made
sure the computers were in good work-
ing condition and configured them so
the children could use them with ease.
He gave away dozens and dozens of
entire computer systems, complete with
monitors, mouse devices, printers, and
speakers, to children of all backgrounds.
I recall one very poor Mexican family
who had previously been living in a van.
They’d finally gotten a home to live in,
and the little girl who was the recipient
of Matt’s generosity of heart was recommended as an outstanding student to
Matt to be a computer recipient by her
school teacher. Another fortunate beneficiary of Matt’s kindness was a little
Mexican-American girl whose mother
my wife had seen through four pregnancies (my wife is an Advanced Registered
Nurse Practitioner—Certified NurseMidwife), and she has mentored the
entire family. Matt was happy to give her
a computer system, too. He never asked
all the dozens and dozens of children
and their families for anything more
than thanks.
His wife, Wanita, says of Matthew:
“He was very capable—no job was impossible for him.” Wanita figures prom-
inently in this story because she was
always very supportive of his involvement in his many and varied projects.
She supplied the grille material for his
D’Appolito speakers and actually helped
him stretch the material over and around
the speakers. She also provided the foregoing biographical information on Matt.
I miss my friend more than words can
say. He will be forever in my mind and
in my heart.
Angel Luis Rivera
[email protected]
CD/VINYL TESTING
After scanning the Feb. '07 issue of audioXpress, especially the CD/vinyl system in Indonesia (p. 31), it occurs to me
that an overlooked issue in the entire
controversy may be related to whatever
occurred in mastering.
I roundly criticize the current practice
of severe peak limiting on many CD
releases in a misguided attempt to make
something with a brickwall peak ceiling
play louder. I thought this practice came
about with the adoption of CDs as our
primary music transport media.
59
3/22/2007 3:50:13 PM
In the process of transferring some
vintage vinyl to CD, I was surprised
when I looked at the audio envelope of
the transferred disc on my audio workstation and found it looked like toothpaste, as most contemporary CD releases
look.
My conjecture, at this point, is that
at least part of the differences heard in
CD/vinyl comparisons are differences
in mastering practice rather than differences between the two methods of
delivery. It is folly to assume that the
master tapes are transferred verbatim
to either medium, because they aren’t.
The record companies have finally recognized the value of their back catalog
and are bringing more and more of their
old recordings back into circulation. But
when they do this, the recordings are
inevitably subjected to “mastering,” and
in the process they may be caressed or
abused.
I plan to explore this further by comparing the long-term envelopes of the
recordings between vinyl and CD releases of the same material. I know that I
have an original Capitol Records release
of Sgt. Peppers, a British pressing of the
same, and the CD release of the same. It
will be interesting to compare the three.
Rick Chinn
Sammamish, Wash.
SAFETY FIRST
I would like to comment about certain
technical viewpoints expressed in the
article “Grounding and System Interfacing” ( Jan. ’07, p. 26).
While I certainly agree with most of
Mr. Galo’s statements and those he attributes to Mr. Whitlock, I must take
exception to one major issue: Attributions to Mr. Whitlock that “. . . earth
grounds are for lightning protection,”
and that “. . . . earth ground plays no role
in protecting people from electrocution.”
These seem to be contradictory statements. If earth grounds are for lightning
protection, then they certainly play a role
in protecting people from electrocution
(from lightning and from a little-known
effect I describe later).
I can only assume that Mr. Whitlock
was only considering his example shown
in Fig. 1 on page 26, in which defective
equipment develops an internal short
60
audioXpress 5/07
and, if the equipment’s chassis were to
be not grounded, the chassis would be
“hot” and thus dangerous. While lightning strikes occur very rarely in most of
the country, Mr. Whitlock’s figure shows
the components of a more menacing,
24-hours-a-day threat to equipment and
humans. This threat is the utility distribution system itself; and this is where
the building ground, in my opinion, does
its “thing.”
The utility transformer does not have
perfect isolation, as depicted in the schematic, between its primary (typically
4kV, 12kV, or more volts) and the building’s load (typically 120, 208, 240, or
480V). This lack of isolation “in the real
world” produces leakage currents that
are primarily due to capacitance effects
(proximity of windings and wires) and
resistance (dirty insulators, poor insulation, and so on). While the leakage
current may be relatively small as far as
building loads are concerned, it can easily provide more than enough amperage
to electrocute someone. This current is
always there to varying degrees, depending upon transformer and wiring condition and age. Admittedly, the ground rod
(or any other grounding system) is not a
perfect connection to “ground” (it has
a low, but not zero resistance), and this
is one of the reasons that it is possible
to measure voltage differences between
grounds.
Stated another way, if the ground rod
(or ground system) were not present,
people would be electrocuted every day
in their homes, office buildings, or factories just by touching perfectly functional
equipment that has no internal faults.
Another point I’d like to make is that
utility companies ground their power
transformer secondaries in addition to
the customers’ grounds. This is due partly to the fact that utility transformers
rarely feed just one customer. If they
had multiple customers, they would have
multiple grounds anyway (and could not
depend upon one customer to provide
an adequate ground for all of the other
customers). Additionally, there are times
when they might need to disconnect the
lines to customer(s), so they certainly
need to be concerned about the safety of
their own personnel and wiring.
Imagine an ungrounded transformer
with a 12kV primary and its secondary
3/22/2007 3:50:14 PM
is not grounded. The utility’s wiring and
the customer’s wiring are rated at 600V.
Without a ground, the wiring is subject
to voltages (thought to be zero) but that
could be many kilovolts, maybe 12kV.
Over a prolonged time, even new wiring
would fail, and—more important than
this—personnel could be subjected to
the same voltage (very, very dangerous).
Ironically, new wiring would lead to a
more dangerous condition, because more
leakage current would be available.
With 50 years as an electrician, I
am personally aware of several dangerous—and lethal—instances of this type
of problem. In one case, a utility worker
was killed near a pumping plant I was
maintaining when he was working inside
a relatively small ground-mounted transformer feeding several houses nearby.
He was working on the secondary when
he received a “high-voltage” burn even
though the 12kV was not accessible.
Another instance occurred when an
electrician working at a pumping plant
received a call that the pumps were not
working in the facility and decided to
measure the voltage-to-ground of a
480V service to the pump building. This
was a three-phase system with a 12kV
primary and 480V secondary (he was
not aware at the time that the secondary
was ungrounded). Normally, the voltageto-ground is 277V on each phase; but
this is only true in a grounded system.
The utility source was in trouble (apparently one or two of the three transformers' primary fuses had blown due to insulation breakdown in the old secondary
wiring as described previously).
Even though the blown fuses would
not permit any pumps to run, that one
good fuse was allowing leakage current
to flow from the high to the low voltage connections. When the electrician
touched the leads of his voltmeter, which
was set to the 750V scale, the meter exploded in his hands and he was slightly
burned and very surprised and frightened. If he had touched any electrical
connections in the building with any
part of his body—thinking that power
was off—he could have been killed by a
current that might not have been strong
enough to even light a light bulb.
As far as the article’s discussion of
good and bad ways to try to eliminate
noise in audio gear, I couldn’t agree
more. Three-prong to two-prong adapters (used as “ground lifters”) is a misunderstood, and thus, dangerous item to
deploy by persons testing or trying to
eliminate ground loops. They are used
because they are convenient; and alternatives may not be known or available at
the time. Perhaps their discontinued use
for these purposes, thanks to Mr. Galo’s
article with information from Mr. Whitlock, may prevent someone’s tragedy.
Gene Davis
[email protected]
CAR AUDIO
I thank you for publishing audioXpress,
without which a large part of my life
would be missing. I do, however, have a
few complaints.
Please, more modern topics and less of
the “good old tube amps.” I understand
why you run so many tube articles, and
empathize. But, may I suggest you take
even a brief look at the automotive amp
industry (repair, maintenance, building)?
61
3/22/2007 3:50:19 PM
I am an automotive installer, but prefer
the permanence of “home” equipment/
installs. I’ve seen Class BD, Class D,
Class T, and so on each hit big in the
automotive field (and last), years before
home stereo even considered them. Also,
nearly all of the innovations for speakers,
especially subwoofers, have come from
auto needs/desires.
So please, do music a favor by getting
more young people involved in how all
this stuff works. (I’m 47 and have saved/
repaired massive quantities of amps,
EQs, subwoofers, and so on through
simple repairs and total rebuilds.) My
daily driven “grocery getter” van has
done 160dB at the dash.
Topics of most interest, I think, might
be:
1. The amp power converter/supplies.
2. Amp topologies and special needs in
autos.
3. Matching subs with amps. (None of
our customers seem to understand
that you don’t need 3000W RMS
and subs to match, when 300W with
efficient drivers will rattle the mirror
off the windshield.)
I’m not sure whether you’re aware—or
care—but it is interesting that in competition concrete filled cars are nearing
190dB. Personally, I prefer sound quality.
Again, thank you for what you do.
Glenn Ray Dõrzök
Dimock, SD
NIGHT SCHOOL
I love your magazine. I also think there
are a few subscribers like myself who
love the hi-fi hobby, love to solder and
rebuild equipment, but may have needed
extra work in areas of mathematics. In
that respect many of your articles go
over my head. But one did not.
The February issue featured an article
by Jan Didden about a visit with some
Indonesian audiophiles. It was a really
great article! Keep those special reports
coming, they make your magazine more
adaptable to all audio nuts. In turn, I will
go back to night school to brush up on
decimals, fractions, and graphs. Thanks
for a great mag.
Mark Korda
[email protected]
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audioXpress 5/07
3/22/2007 3:50:22 PM

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