Wing TV-WP4-D5 Compa..

Transcription

Project Report
Wing TV
Services to Wireless, Integrated, Nomadic, GPRSUMTS & TV handheld terminals
D5 – Wing-TV Companies lab tests
Editor: Olivier Rousset, TeamCast
Abstract
This document contains the contributions writing by several Wing TV partners. These
contributions resulted from individual test campaign realized by each partner. Seven
contributions are included in this document:
•
RAI laboratory test: Evaluation of DVB-H receiver’s performances with hierarchical
modulation
•
Nokia laboratory test: Complete results from the first common laboratory session (Turin)
in order to estimate the relation between the different DVB-H modes
•
Ericsson laboratory test: Evaluation of an algorithm for ICI cancellation in an FPGA, and
measure the performance
•
TeamCast laboratory test: Assessments the performances of a transmission using DVBH with a bandwidth shrunk to 1.75 MHz
•
DiBcom laboratory test: Test and validate the DVB-H receiver in pure SFN environment.
•
Mier Comunicaciones laboratory test: Evaluation of on-channel repeaters in DVB-H
transmission environment.
•
Sidsa laboratory test: Design of an automatic test bench in order to estimate the relation
between the different DVB-H modes.
Project
Wing TV
Confidential
May 2006
Participants in project Wing TV are:
•
Åbo Akademi University Turku (AAU)
•
Antenna Hungaria
•
Dibcom
•
DIGITA OY
•
Elektrobit Ltd.
•
Ericsson AB
•
Fundació Privada Universitat I Tecnologia (FUNITEC) - Universitat Ramon Llull
•
Mier Comunicaciones S.A.
•
Nokia Corporation
•
Nozema Services
•
Philips Electronics Nederland B.V. Research
•
RAI – CRIT (Centro Ricerche e Innovazione Tecnologica)
•
Retevisión (abertis telecom group)
•
Rohde&Schwarz, Broadcasting Division
•
SIDSA
•
Tampere University of Technology (TUT)
•
TeamCast
•
Technical University Braunschweig, Institut für Nachrichtentechnik
•
Telefónica I+D (TID)
•
Thales Broadcast & Multimedia
•
T-Systems International GmbH Media&Broadcast
•
University of Turku (UTU)
Wing TV - Services to Wireless, Integrated, Nomadic, GPRS-UMTS & TV handheld terminals
Hierarchcal Modulation Issues
Editor: Name, company
Project coordinator: Jesús Fernández, Retevision (abertis telecom group)
CELTIC published project result
•
2005 CELTIC participants in project Wing TV
Disclaimer
This document contains material, which is copyright of certain CELTIC PARTICIPANTS and may
not be reproduced or copied without permission. The information contained in this document is the
proprietary confidential information of certain CELTIC PARTICIPANTS and may not be disclosed
except in accordance with the regulations agreed in the Project Consortium Agreement (PCA).
The commercial use of any information in this document may require a licence from the proprietor
of that information.
Neither the PARTICIPANTS nor the CELTIC Initiative warrant that the information contained in this
document is capable of use, or that use of the information is free from risk, and accept no liability
for loss or damage suffered by any person using the information.
CELTIC Wing TV project report
Preface
(to be provided by the project coordinator)
 2006 CELTIC participants in project Wing TV
page 3 (3)
page 4 (4)
Executive Summary
This document contains the contributions writing by several Wing TV partners. These contributions
resulted from individual test campaign realized by each partner. The same measurement
methodology has been used in all laboratories tests (common and individual) based on the
document “Validation task force report” and Wing TV D8 report. Thank to the same methodologies,
all results should be compared and analysed.
The RAI, NOKIA and Sidsa tests are the continuity of the common laboratories tests in order to
complete and to confirm the DVB-H receiver performances (C/N and Frequency Doppler) with the
TU6 profile in the different DVB-H modes. An analysis has been realized to determine the influence
of each DVB-H modulation parameters and, then, to help the user to choose the best DVB-H mode
according to this application.
The Ericsson tests permit to evaluate a new ICI canceller algorithm at low complexity and achieve
good performance by using a window function and cancellation of ICI on one sub-carrier from two
adjacent sub-carriers. The functionality of the ICI cancelling algorithm has been proved using both
frequency offset and a time-variant channel causing Doppler spread, with good results, very close
to the theoretical improvement.
The TeamCast tests are a first approach of the bandwidth shrinkage effect in a context no DVB-H
standardized but compatible of the T-DMB (or DAB) standard. The performances are consistent
between 1.75MHz and 7MHz: Maximum Frequency Doppler four times lower, receiver
consumption four times less important. The remaining work consists to evaluate DAB modulation
with the same methodology.
The DiBcom tests have consisted to validate their DVB-H receiver in environment combining SFN
and mobility. This receiver is not sensitive to the 2 paths channel if the delay between these 2
paths is less than the guard interval. There are not notion of pre-echo or post-echo in these results
(the curve a symmetrical around 0). Even with a delay higher than the guard interval the
demodulation is possible.
The Mier tests have permitted to evaluate the performance of DVB-T gapfillers with standard echo
cancellers under different simulated conditions for DVB-H networks (this is closer to urban areas),
showing limitations due the multiple objects and variant conditions. Advanced echo cancellers
should be considered to manage these harder conditions.
page 5 (5)
List of Authors
Andrea Bertella, Davide Milanesio, Bruno Sacco, Mirto Tabone (Rai-CRIT)
Pekka Talmola (Nokia)
Leif Wilhelmsson (Ericsson)
Vincent Recrosio (DiBcom)
Olivier Rousset, Thibault Bouttevin, Gérard Faria (Teamcast)
Raimon Casals, Domenec Iborra, Xavier Fustagueras, Eduard Gil (Mier Comunicaciones)
Juan M. Roldán (Sidsa)
page 6 (6)
Table of Contents
Preface................................................................................................................................................ 3
Executive Summary............................................................................................................................ 4
List of Authors ..................................................................................................................................... 5
Table of Contents ............................................................................................................................... 6
Abbreviations ...................................................................................................................................... 9
Definitions ......................................................................................................................................... 11
1 Introduction................................................................................................................................ 12
2 Wing TV laboratory trials at Rai................................................................................................. 13
2.1
Measurements on hierarchical modulation ........................................................................ 13
2.1.1
Introduction ................................................................................................................. 13
2.1.2
Preliminary laboratory tests on DVB-T receivers........................................................ 13
2.1.3
DVB-H laboratory tests: performance evaluation ....................................................... 15
3 NOKIA laboratory tests.............................................................................................................. 17
3.1
Purpose of the test ............................................................................................................. 17
3.2
Tests done.......................................................................................................................... 17
3.3
Test bed ............................................................................................................................. 17
3.4
Results ............................................................................................................................... 17
3.5
Analysis .............................................................................................................................. 18
4 Ericsson laboratory tests ........................................................................................................... 22
4.1
Implementation................................................................................................................... 22
4.1.1
System Design ............................................................................................................ 22
4.1.2
Design Methodology ................................................................................................... 22
4.1.3
Overview ..................................................................................................................... 23
4.1.3.1
Window Unit......................................................................................................... 24
4.1.3.2
Channel Estimator ............................................................................................... 24
4.1.3.3
ICI Canceller ........................................................................................................ 24
4.2
Detailed Description ........................................................................................................... 24
4.2.1.1
Window Unit......................................................................................................... 24
4.2.1.2
Channel Estimator ............................................................................................... 26
4.2.1.3
ICI-Canceller ........................................................................................................ 28
4.3
Measurements.................................................................................................................... 29
4.3.1
Laboratory Setup ........................................................................................................ 29
4.3.2
Measured Results ....................................................................................................... 30
4.3.2.1
Frequency Offset ................................................................................................. 30
4.3.2.2
Doppler Spread.................................................................................................... 33
4.4
Conclusions and Improvements......................................................................................... 34
4.4.1
Conclusions................................................................................................................. 34
4.4.2
Improvements ............................................................................................................. 35
4.4.2.1
Window Unit......................................................................................................... 36
4.4.2.2
Channel Estimation.............................................................................................. 36
4.4.2.3
ICI Unit ................................................................................................................. 36
5 TeamCast lab tests.................................................................................................................... 37
5.1
Purpose of the test ............................................................................................................. 37
5.2
Test bench presentation..................................................................................................... 37
5.2.1
Synoptic ...................................................................................................................... 37
5.2.2
Method ........................................................................................................................ 38
5.3
Results ............................................................................................................................... 39
5.3.1
Mode 2K...................................................................................................................... 39
5.3.2
Mode 4K...................................................................................................................... 40
5.3.3
Mode 8K...................................................................................................................... 41
5.4
Analysis of the DVB-H in 1.75MHz bandwidth................................................................... 41
5.4.1
Comparison with the 7MHz bandwidth ....................................................................... 41
5.4.2
Comparison with DAB standard.................................................................................. 42
6 DiBcom lab tests........................................................................................................................ 44
6.1
Pure SFN : 2 paths channel ............................................................................................... 44
6.1.1
General ....................................................................................................................... 44
6.1.2
Paths channel without added noise ............................................................................ 44
6.1.3
Paths channel with added noise ................................................................................. 45
6.2
DiBcom results ................................................................................................................... 45
6.2.1
8K FFT mode .............................................................................................................. 45
page 7 (7)
6.2.2
2K FFT mode.............................................................................................................. 47
6.2.3
Comparison between Guard Interval.......................................................................... 50
6.2.4
Code rate influence for SFN mode............................................................................. 52
6.3
Conclusion ......................................................................................................................... 52
7 Mier Comunicaciones laboratory tests...................................................................................... 55
7.1
WingTV laboratory trails at RAI ......................................................................................... 55
7.1.1
On-channel repeaters test at RAI............................................................................... 55
7.1.2
Results........................................................................................................................ 57
7.2
Design of a RF multipath generator................................................................................... 57
7.3
DVB-H On-channel repeaters with echo cancellers laboratory tests ................................ 59
7.3.1
Profile 1....................................................................................................................... 60
7.3.2
Profile 2....................................................................................................................... 60
7.3.3
Profile 3....................................................................................................................... 61
7.4
Conclusions ....................................................................................................................... 61
8 SIDSA SuperLode automatic test tool ...................................................................................... 63
8.1
Introduction ........................................................................................................................ 63
8.2
Setup.................................................................................................................................. 63
8.3
Static Channel Tests.......................................................................................................... 64
8.4
Dynamic Channel Tests..................................................................................................... 65
8.5
Output Report .................................................................................................................... 65
8.6
WingTV measure results: .................................................................................................. 66
9 Conclusions............................................................................................................................... 68
References ....................................................................................................................................... 69
page 8 (8)
Abbreviations
ACI
Adjacent Channel Interference
AWGN
Additive white Gaussian noise
BB
Base Band
BER
Bit Error Rate
C/N
Carrier to Noise ratio
CCI
Co-Channel Interference
DUT
Device Under Test
DVB
Digital Video Broadcasting
DVB-H
Digital Video Broadcasting - Handhelds
DVB-MUX
Digital Video Broadcasting - MUltipleX
DVB-SI
Digital Video Broadcasting - Service Information
DVB-T
Digital Video Broadcasting – Terrestrial
EIT
Event Information Table
END
Equivalent Noise Degradation
ESR
Erroneous Second Ratio
ETSI
European Telecommunications Standards Institute
FEC
Forward Error Correction
FER
Frame Error Rate
FFT
Fast Fourrer Transform
FPGA
Field Programmable Gate Array
GI
Guard Interval
HP
High Priority bit stream
ICI
Inter-Carrier Interference
IF
Intermediate Frequency
INT
IP/MAC Notification Table
IP
Internet Protocol
IPE
IP Encapsulator
IT
Interoperability Tests
LP
Low Priority bit stream
MBRAI
Mobile and Portable DVB-T/H Radio Access Interface Specifications
MCP
Multimedia Car Platform
MER
Modulation Error Ratio
MFER
Multi Protocol Encapsulation frame Frame Error Rate
MFN
Multi Frequency Network
MIP
Mega-frame Initialization Packet
MPE
Multi-Protocol Encapsulation
MPE-FEC
MPE Forward Error Correction
MPEG
Motion Picture Expert Group
MUX
MUltipleX
NIT
Network Information Table
OFDM
Orthogonal Frequency Division Multiplex
PAT
Program Association Table
PER
Packet Error Rate
PID
Packet IDentifier
PL
Physical Layer
PMT
Program Map Table
PSI
Program Specific Information
QAM
Quadrature Amplitude Modulation
QEF
Quasi Error Free
QoR
Quality of Restitution
QPSK
Quaternary Phase Shift Keying
RF
Radio Frequency
SDT
Service Description Table
SFN
Single Frequency Networks
SI
Service Information
SI/PSI
Service Information / Program Signaling Information
TPS
Transmission Parameter Signalling
TS
Transport Stream
UHF
Ultra-High Frequency (300 MHz to 3 000 MHz)
VHF
Very High Frequency (30 MHz to 300 MHz)
page 9 (9)
page 10 (10)
Definitions
FER/MFER
The TU6 propagation channel lead to a QEF criterion instability. Because of this instability, it is very
difficult to get a reliable measurement with the BER before Reed-Solomon decoder. It is preferable
to use the FER (Frame Error Rate) for DVB-T system and the MFER (MPE Frame Error Rate) for
DVB-H system. This criterion detects errors after complete demodulation and decoding, i.e. after
Viterbi and Reed-Solomon decoding for DVB-T and Viterbi/Reed-Solomon/MPE-FEC for DVB-H.
This detection is realised by monitoring the error flag in the header of each demodulated TS
packet. A threshold of FER/MFER = 5%, i.e. 5% of erroneous frames on the total received frames
is considered to be the limit of an acceptable quality of image. This threshold is called FER5 for
DVB-T and MFER5 for DVB-H.
Doppler frequency shift
The Doppler frequency shift is the offset between the emitted frequency and the received
frequency when the receiver is in motion compared to the emitter. This frequency shift depends of
the relative speed between emitter and receiver.
1
page 11 (11)
Introduction
In WING-TV project, WP4 is in charge to coordinate all the laboratory tests produced during the
project, including common laboratory test sessions and individual tests campaigns.
The purpose of this document is to compile the work and results of laboratory tests from individual
companies during the Wing-TV project
These individual laboratory tests campaigns have assessed the following points:
RAI laboratory test:
Evaluation of DVB-H receiver’s performances with hierarchical modulation.
Nokia laboratory test:
Complete results from the first common laboratory session (Turin) in order to estimate the
relation between the different DVB-H modes.
Ericsson laboratory test:
Evaluation of an algorithm for ICI cancellation in an FPGA, and measure the performance.
TeamCast laboratory test:
Assessments the performances of a transmission using DVB-H with a bandwidth shrunk to
1.75 MHz.
DiBcom laboratory test:
Test and validate the DVB-H receiver in pure SFN environment.
Mier Comunicaciones laboratory test:
Evaluation of on-channel repeaters in DVB-H transmission environment.
Sidsa laboratory test:
Design of an automatic test bench in order to estimate the relation between the different
DVB-H modes.
page 12 (12)
2
Wing TV laboratory trials at Rai
2.1
Measurements on hierarchical modulation
2.1.1
Introduction
Hierarchical modulation allows for the transmission of two streams, having different bit-rates and
performance, in the same RF channel.
The sum of the bit-rates of the two streams is equal to the bit-rate of a non-hierarchical stream
using the same modulation (even if the net data rate is slightly lower, due to the double MPEG-2
TS overhead).
As regards performance, the better protected HP stream has about the same noise sensitivity as a
standard QPSK stream (an α factor of 2 can be chosen to improve the noise sensitivity of
the HP stream), with an impairment of 1-2 dB due to the “noise-like” presence of the LP stream;
the LP stream has the same noise sensitivity as the overall scheme in case of α=1, and slightly
impaired in case of higher values of α.
DVB-H networks will be designed for different reception scenarios (i.e. portable, indoor, mobile,
etc.) with respect to fixed DVB-T reception with roof antenna. Therefore, thanks to hierarchical
modulation, it is possible to compensate the differences in the coverage areas of the two streams,
i.e. [1]:
•
DVB-H on HP (i.e. for indoor coverage) and DVB-T on LP (i.e. for fixed reception),
•
DVB-H on both HP and LP, with different robustness and coverage areas.
2.1.2
Preliminary laboratory tests on DVB-T receivers
Investigations made on some commercial DVB-T Set-Top-Boxes showed that, unfortunately, many
of the consumer equipments currently available of the market are not able to correctly decode
hierarchical modulation streams: generally, only the HP stream is decoded, and not the LP stream.
This in principle could be a problem in case of introduction of hierarchical modulation for DVB-H
services sharing the same RF channel of DVB-T services. However, according to the
manufacturers, a software upgrade of the Set-Top-Boxes should be sufficient.
Among ten DVB-T receivers tested at Rai-CRIT, only two of them were able to receive both LP and
HP streams.
The following Table 1 summarizes the results for these receivers for hierarchical and non
hierarchical modes.
Table 1 Laboratory measurements on commercial DVB-T receivers
with hieararchical modulations
page 13 (13)
Mode
Constellation
QPSK in 64QAM
QPSK in 64QAM
QPSK in 64QAM
QPSK in 64QAM
QPSK in 64QAM
QPSK in 64QAM
QPSK in 64QAM
QPSK in 64QAM
QPSK in 64QAM
QPSK
64QAM
64QAM
64QAM
Useful bit
Reference (2)
rate
C/N
LP
HP
C/N LP C/N HP C/N LP C/N HP C/N LP HP
16.3
8.8
14.8
7.9
14.6
8.9
17.8
7.1
16.5
6.3
16.3
6.5 12.06 6.03
20.8
5.7
19.5
5.5
NA
NA
18.7
8.8
17.3
7.9
16.9
8.9
20.4
7.1
19.0
6.3
18.9
6.5 16.09 6.03
23.7
5.7
22.7
5.5
NA
NA
19.9
8.8
18.7
7.9
18.6
8.9
22.0
7.1
20.6
6.3
21.0
6.5 18.10 6.03
25.5
5.7
24.8
5,5
NA
NA
5.7
4.6
3.1
6.03
14.6
12.7
14.4
18.10
16.0
16.5
24.13
17.5
19.0
17.7
18.0
27.14
STB 1 (1)
α
1
2
4
1
2
4
1
2
4
x
x
x
x
Rate Rate
LP
HP
1/2
1/2
1/2
1/2
1/2
1/2
2/3
1/2
2/3
1/2
2/3
1/2
3/4
1/2
3/4
1/2
3/4
1/2
1/2
1/2
2/3
3/4
STB 2 (1)
(1) = Threshold Of Visibility.
(2) = Values referred to QEF threshold for Gaussian channel and do not consider the implementation margin [2].
In HP (Figure 1), the loss due to hierarchical modulation varies from -3.3 dB (with α=1) to –0.9 dB
(with α =4)
In LP (Figure 2), considering the 64-QAM 2/3 as reference, the loss due to hierarchical modulation
is (negative means loss, positive means gain):
•
64-QAM ½ hierarchical: from +1.2 dB (with α=1) to -3.5 dB (with α=4)
•
64-QAM 2/3 hierarchical: from -1.3 dB (with α=1) to -6.7 dB (with α=4)
•
64-QAM ¾ hierarchical: from -2.7 dB (with α=1) to -8.8 dB (with α=4)
QPSK 1/2 vs Hierarchical modes
NML (Hierarchical-Non
hierarchical)
0,0
-0,5
-1,0
-1,5
-2,0
-2,5
-3,0
-3,5
QPSK in 64QAM, 1/2, a=1
Figure 1 Loss in the HP stream with respect to non-hierarchical modulation (DVB-T signal)
page 14 (14)
hierarchical)
64QAM 2/3 vs Hierarchical modes
2.0
1.0
0.0
-1.0
-2.0
-3.0
-4.0
-5.0
-6.0
-7.0
-8.0
-9.0
QP SK in QP SK in QP SK in QP SK in QP SK in QP SK in QP SK in QP SK in QP SK in
64QA M 64QA M 64QA M 64QA M 64QA M 64QA M 64QA M 64QA M 64QA M
1/2, a=1 1/2, a=2 1/2, a=4 2/3, a=1 2/3, a=2 2/3, a=4 3/4, a=1 3/4, a=2 3/4, a=4
Figure 2 Loss in the LP stream with respect to non-hierarchical modulation (DVB-T signal)
2.1.3
DVB-H laboratory tests: performance evaluation
DVB-H performance with respect to non-hierarchical modulation has been evaluated on Gaussian
channel by means of laboratory measurements, using the Teamcast/DiBcom Showcast receiver.
The DVB-H stream was included in the HP stream (QPSK ½, MPE-FEC ¾).
The C/N values for MFER = 5% are reported in Figure 3. The losses due to hierarchical modulation
are reported in Table 2 (compared with simulation results coming from DVB-H VTF) and Figure 4.
Non Hierarchical vs Hierarchical modes
C/N@MFER<5%
7
6
C/N (dB)
5
4
3
2
1
0
QPSK 1/2
QPSK 1/2 in
QPSK 1/2 in
QPSK 1/2 in
64QAM 1/2, a=1 64QAM 1/2, a=2 64QAM 1/2, a=4
Figure 3 C/N for DVB-H signal in the HP stream
Table 2 Loss in the LP stream with respect to non-hierarchical modulation
(DVB-H signal, QPSK ½ in 64-QAM)
Laboratory measurements
DVB-H Simulation Task Force
α=1
-4.1 dB
-5 dB
α=2
-2.6 dB
-3 dB
α=4
-1.6 dB
page 15 (15)
Non Hierarchical vs Hierarchical modes
Hierarchical)
0
-0,5
-1
-1,5
-2
-2,5
-3
-3,5
-4
-4,5
QPSK 1/2 in 64QAM 1/2, QPSK 1/2 in 64QAM 1/2, QPSK 1/2 in 64QAM 1/2,
a=1
a=2
a=4
Figure 4 Loss in the HP stream with respect to non-hierarchical modulation (DVB-H signal)
Such values can be useful for planning purposes, evaluating the reduction of the coverage area
with respect to non-hierarchical modulation.
page 16 (16)
3
NOKIA laboratory tests
3.1
Purpose of the test
The laboratory measurements performed by Nokia in the Nokia laboratory were direct continuation
of the work done in the common laboratory session in Turin. As time did not allow all the agreed
measurements to be done in the Turin laboratory it was agreed that the measurements will be
completed at home.
3.2
Tests done
The performed tests were according the agreed Turin test plan:
1. Performance with top 22 modes in AWGN, Rayleigh and TU-6 channel ( Fdmax, C/N at
Fdmax/2, Fd at 3dB, C/N at 10 Hz and 2 Hz)
2. Effect of the FFT-Size with 2k and 8k
3. Effect of the guard interval
4. Effect of the Burst Length
3.3
Test bed
The test bed was a similar R&S SFQ, which was used in Turin. The device was first calibrated, so
that similar C/N values were obtained as in the Turin measurements with the same receiver.
The tested receiver was the Nokia receiver with MPE-FEC simulation. MFER 5% error criteria was
used throughout the testing.
3.4
Results
The results for the Top 22 Modes are shown in the following table.
Table 3 Top modes results
"Top modes"
8K
1/4 QPSK
8K
1/4 QPSK
8K
1/4 QPSK
8K
1/4 QPSK
8K
1/4 QPSK
8K
1/4 QPSK
8K
1/4 QPSK
8K
1/4 QPSK
8K
1/4 QPSK
8K
1/4 16QAM
8K
1/4 16QAM
8K
1/4 16QAM
8K
1/4 16QAM
8K
1/4 16QAM
8K
1/4 16QAM
8K
1/4 16QAM
8K
1/4 16QAM
8K
1/4 64QAM
8K
1/4 64QAM
8K
1/4 64QAM
8K
1/4 64QAM
8K
1/4 64QAM
1/2
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
1/2
1/2
2/3
2/3
2/3
1/2
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
5/6
7/8
2/3
3/4
5/6
512
512
512
512
512
512
512
512
512
512
512
512
512
512
512
512
512
512
512
512
512
512
250
250
250
250
250
250
250
250
250
250
250
250
250
250
250
250
250
250
250
250
250
250
AWGN Rayleigh Fd Max Fd Max / 23 dB
4.1 dB
6.8 dB 105 Hz
9.0 dB 100 Hz
4.3 dB
7.2 dB 105 Hz
9.5 dB
95 Hz
4.3 dB
7.1 dB 100 Hz 10.0 dB
95 Hz
4.4 dB
7.4 dB 100 Hz 11.0 dB
90 Hz
4.4 dB
7.5 dB
95 Hz 11.0 dB
90 Hz
6.0 dB
9.9 dB
95 Hz 12.0 dB
90 Hz
6.0 dB 10.0 dB
95 Hz 12.5 dB
90 Hz
6.0 dB 10.0 dB
90 Hz 13.5 dB
85 Hz
6.0 dB 10.1 dB
90 Hz 14.5 dB
85 Hz
9.4 dB 12.0 dB
90 Hz 14.5 dB
85 Hz
9.5 dB 12.0 dB
90 Hz 15.0 dB
85 Hz
9.5 dB 12.1 dB
90 Hz 16.0 dB
80 Hz
9.6 dB 12.2 dB
85 Hz 16.5 dB
80 Hz
11.7 dB 15.6 dB
85 Hz 18.0 dB
80 Hz
11.8 dB 15.7 dB
80 Hz 18.5 dB
75 Hz
11.9 dB 15.7 dB
80 Hz 19.5 dB
70 Hz
11.9 dB 15.9 dB
75 Hz 20.5 dB
70 Hz
14.0 dB 17.4 dB
70 Hz 21.5 dB
65 Hz
14.1 dB 17.4 dB
70 Hz 22.5 dB
60 Hz
17.5 dB 22.2 dB
60 Hz 25.0 dB
50 Hz
17.6 dB 22.2 dB
50 Hz 25.5 dB
45 Hz
17.7 dB 22.3 dB
45 Hz 27.0 dB
40 Hz
10 Hz
9.5 dB
9.5 dB
10.5 dB
11.0 dB
11.0 dB
12.0 dB
13.0 dB
13.5 dB
14.5 dB
15.5 dB
15.5 dB
16.5 dB
17.0 dB
18.5 dB
19.0 dB
19.5 dB
20.5 dB
21.5 dB
22.5 dB
25.0 dB
25.5 dB
27.0 dB
2 Hz
10.0 dB
10.5 dB
11.0 dB
12.0 dB
12.0 dB
13.5 dB
13.5 dB
15.0 dB
15.0 dB
16.5 dB
16.5 dB
17.5 dB
18.0 dB
19.5 dB
20.0 dB
20.5 dB
21.0 dB
22.0 dB
22.5 dB
25.5 dB
26.0 dB
27.5 dB
Results of the FFT Size test are shown in the table below.
page 17 (17)
Table 4 FFT-size results
Effect of FFT-size
2K
1/4 QPSK
2K
1/4 16QAM
2K
1/4 16QAM
2K
1/4 64QAM
4K
1/4 QPSK
4K
1/4 16QAM
4K
1/4 16QAM
4K
1/4 64QAM
8K
1/4 QPSK
8K
1/4 16QAM
8K
1/4 16QAM
8K
1/4 64QAM
1/2
1/2
2/3
2/3
1/2
1/2
2/3
2/3
1/2
1/2
2/3
2/3
3/4
3/4
3/4
2/3
3/4
3/4
3/4
2/3
3/4
3/4
3/4
2/3
512
512
512
512
512
512
512
512
512
512
512
512
250
250
250
250
250
250
250
250
250
250
250
250
4.3 dB
7.2 dB 410 Hz 10.5 dB 380 Hz
9.4 dB 12.0 dB 360 Hz 15.5 dB 340 Hz
11.8 dB 15.5 dB 330 Hz 18.5 dB 290 Hz
17.5 dB 21.7 dB 215 Hz 24.5 dB 180 Hz
10 Hz
10.5 dB
15.5 dB
19.0 dB
24.5 dB
2 Hz
11.5 dB
16.5 dB
20.0 dB
25.0 dB
4.3 dB
9.5 dB
11.8 dB
17.5 dB
95 Hz
85 Hz
75 Hz
50 Hz
10.5 dB
15.5 dB
19.0 dB
25.0 dB
11.0 dB
16.5 dB
20.0 dB
25.5 dB
4.3 dB
7.2 dB 410 Hz 10.5 dB 380 Hz
4.2 dB
7.2 dB 450 Hz 10.5 dB 420 Hz
3.9 dB
6.8 dB 470 Hz 10.0 dB 450 Hz
3.6 dB
6.5 dB 500 Hz 10.0 dB 470 Hz
11.8 dB 15.7 dB
80 Hz 18.5 dB
75 Hz
11.6 dB 15.5 dB
90 Hz 18.5 dB
80 Hz
11.2 dB 14.9 dB
95 Hz 18.5 dB
85 Hz
11.0 dB 14.7 dB 100 Hz 18.0 dB
90 Hz
17.6 dB 21.4 dB
55 Hz 24.5 dB
45 Hz
17.5 dB 21.4 dB
60 Hz 24.0 dB
50 Hz
17.0 dB 20.7 dB
65 Hz 23.5 dB
50 Hz
10 Hz
10.5 dB
10.5 dB
10.0 dB
10.0 dB
19.0 dB
19.0 dB
18.5 dB
18.0 dB
24.5 dB
24.0 dB
23.5 dB
2 Hz
11.5 dB
11.5 dB
11.0 dB
11.0 dB
20.0 dB
20.0 dB
19.5 dB
19.0 dB
25.5 dB
25.5 dB
25.0 dB
7.1 dB
12.0 dB
15.7 dB
22.2 dB
100 Hz
90 Hz
80 Hz
60 Hz
10.0 dB
15.0 dB
18.5 dB
25.0 dB
Results of the Guard Interval test are shown in the table below.
Table 5 Guard interval results
Effect of Guard Interval
2K
1/4 QPSK 1/2
2K
1/8 QPSK 1/2
2K 1/16 QPSK 1/2
2K 1/32 QPSK 1/2
8K
1/4 16QAM 2/3
8K
1/8 16QAM 2/3
8K 1/16 16QAM 2/3
8K 1/32 16QAM 2/3
8K
1/4 64QAM 2/3
8K
1/8 64QAM 2/3
8K 1/16 64QAM 2/3
8K 1/32 64QAM 2/3
3/4
3/4
3/4
3/4
3/4
3/4
3/4
3/4
2/3
2/3
2/3
2/3
512
512
512
512
512
512
512
512
512
512
512
512
250
250
250
250
250
250
250
250
250
250
250
250
Results of the Burst Length test are shown in the table below.
Table 6 Burst length results
Effect of burst size and length
8K
1/4 16QAM 1/2
8K
1/4 16QAM 1/2
8K
1/4 16QAM 1/2
8K
1/4 16QAM 1/2
8K
1/4 16QAM 1/2
8K
1/4 16QAM 1/2
3.5
3/4
3/4
3/4
3/4
3/4
3/4
Rows Length PID
AWGN Rayleigh Fd Max Fd Max / 2 3 dB
20 Hz
1024
500
1003 9.7 dB 12.4 dB 85 Hz
16.0 dB 75 Hz 16.0 dB
512
250
1002 9.7 dB 12.5 dB 85 Hz
16.2 dB 75 Hz 16.2 dB
256
125
1004 9.7 dB 12.4 dB 85 Hz
16.2 dB 75 Hz 16.3 dB
1024
250
1005 9.7 dB 12.4 dB 85 Hz
16.2 dB 75 Hz 16.2 dB
512
125
1006 9.7 dB 12.5 dB 83 Hz
16.7 dB 75 Hz 16.9 dB
256
62.5 1007 9.6 dB 12.4 dB 83 Hz
16.6 dB 75 Hz 16.9 dB
15 Hz
16.0 dB
16.2 dB
17.2 dB
16.2 dB
16.8 dB
16.9 dB
10 Hz
16.2 dB
16.5 dB
17.1 dB
16.4 dB
17.0 dB
16.9 dB
7 Hz
16.5 dB
17.0 dB
17.4 dB
17.2 dB
17.2 dB
5 Hz
17.0 dB
17.0 dB
17.3 dB
17.2 dB
17.2 dB
17.1 dB
2 Hz
17.2 dB
17.2 dB
17.2 dB
17.2 dB
17.3 dB
1 Hz
17.2 dB
17.2 dB
17.3 dB
17.2 dB
17.3 dB
16.9 dB
Analysis
The results in the top 22 modes have mainly been used to find out how the MPE-FEC code rate
affects the C/N performance in the TU-6 channel. The purpose of this was to be able to extend the
“Typical” reference receiver presented in the DVB-H Implementation Guidelines. Currently this
includes only MPE-FEC code rate ¾. To see the effect of the code rate the following deltas were
calculated from the results.
page 18 (18)
Table 7 MPE-FEC code rate influence
Mode
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
Delta
Fd Max / 2 3 dB
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
1/4
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
16QAM
16QAM
16QAM
16QAM
16QAM
16QAM
16QAM
16QAM
1/2
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
1/2
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
10Hz
2Hz
-1
-0.5
0
1
1
5
0
0
-5
-5
-1
-1
0
0.5
0.5
-1
-0.5
0
1
1
-0.5
0
1
2
0
0
-5
-5
-1
0
0.5
1.5
0
0
1.5
1.5
-0.5
0
1
1.5
0
0
-5
-5
0
0
1
1.5
0
0
1
1.5
-0.5
0
1
2
5
0
-5
-5
-0.5
0
0.5
1.5
-0.5
0
0.5
1
Based on this table the following extended reference receiver was created.
Table 8 Reference receiver
"Typical" Reference Receiver
Guard interval = 1/4
Modulation Code rate
Bitrate
[Mbit/s]
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
16-QAM
16-QAM
16-QAM
16-QAM
16-QAM
16-QAM
16-QAM
16-QAM
64-QAM
64-QAM
64-QAM
64-QAM
64-QAM
1/2
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
1/2
1/2
2/3
2/3
2/3
4.98
4.98
4.98
4.98
4.98
6.64
6.64
6.64
6.64
9.95
9.95
9.95
9.95
13.27
13.27
13.27
13.27
14.93
14.93
19.91
19.91
19.91
2k
MPE FEC
CR
1/2
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
2/3
3/4
5/6
7/8
5/6
7/8
2/3
3/4
5/6
Bitrate
[Mbit/s]
2.49
3.32
3.74
4.15
4.36
4.43
4.98
5.53
5.81
6.63
7.46
8.29
8.71
8.85
9.95
11.06
11.61
12.44
13.06
13.27
14.93
16.59
Speed at Fd3dB [km/h]
4k
8k
C/Nmin
Fd3dB
474
698
C/Nmin
Fd3dB
474
698
C/Nmin
Fd3dB
474
698
[dB]
8.5
9
9.5
10
10.5
12
12.5
13.5
14.5
15
15.5
16.5
17.5
18
18.5
19.5
20.5
21.5
22.5
25
25.5
27
[Hz]
400
380
380
360
360
360
360
340
340
340
340
320
320
320
320
300
280
260
240
200
180
160
MHz
911
866
866
820
820
820
820
775
775
775
775
729
729
729
729
684
638
592
547
456
410
365
MHz
619
588
588
557
557
557
557
526
526
526
526
495
495
495
495
464
433
402
371
309
279
248
[dB]
8.5
9
9.5
10
10.5
12
12.5
13.5
14.5
15
15.5
16.5
17.5
18
18.5
19.5
20.5
21.5
22.5
25
25.5
27
[Hz]
200
190
190
180
180
180
180
170
170
170
170
160
160
160
160
150
140
130
120
100
90
80
MHz
456
433
433
410
410
410
410
387
387
387
387
365
365
365
365
342
319
296
273
228
205
182
MHz
309
294
294
279
279
279
279
263
263
263
263
248
248
248
248
232
217
201
186
155
139
124
[dB]
8.5
9
9.5
10
10.5
12
12.5
13.5
14.5
15
15.5
16.5
17.5
18
18.5
19.5
20.5
21.5
22.5
25
25.5
27
[Hz]
100
95
95
90
90
90
90
85
85
85
85
80
80
80
80
75
70
65
60
50
45
40
MHz
228
216
216
205
205
205
205
194
194
194
194
182
182
182
182
171
159
148
137
114
103
91
MHz
155
147
147
139
139
139
139
132
132
132
132
124
124
124
124
116
108
101
93
77
70
62
Next the measurements with different Guard Intervals were analyzed to understand the effect of
the Guard Interval and to be able to apply this in a standard format to the proposed extended
reference receiver. The relative Doppler performance of the receiver with different modulations is
shown in the following picture. As can be seen the drops with increased GI in a rather regular
manner. Thus it was possible to propose a standard behavior for the GI versus Doppler. This is
shown in the following table.
page 19 (19)
GI-effect in Doppler
1.0
1.00
0.95
0.9
Fd/Fd(1/32)
0.90
0.8
0.80
QPSK
16QAM
64QAM
0.70
0.60
1/32
1/16
1/8
1/4
Figure 5: Guard interval effect in Doppler
Table 9 Guard interval effect
GI
1/4
1/8
1/16
1/32
Fd/Fd(1/32) Fd/Fd(1/4)
0.80
0.90
0.95
1.00
1.000
1.125
1.188
1.250
When this is applied to the GI=1/4 extended reference receiver it is possible to get a full set of
performance figures for all Guard Intervals.
From the FTT-size measurements it can be seen that the expected theoretical effect of the FFTSize is verified, thus it is possible to have the 2/2/2 relation in the reference receiver tables.
The burst length measurements were mainly done to study the effect of C/N increase in the low
Doppler frequencies in the TU-6 channels. There the theory is saying that the increase will happen
at different Doppler frequencies depending on the burst length, which roughly corresponding the
interleaving depth.
The results of this test are shown in the picture below. This zoomed picture shows the bend of the
curves at different frequencies as expected. The overall increase in the C/N is rather small, in the
order of 1 dB.
page 20 (20)
Burst Size and Length
20.0 dB
19.0 dB
18.0 dB
17.0 dB
1024-500
512-250
1024-250
256-62.5
C/N [dB]
16.0 dB
15.0 dB
256-125
512-125
14.0 dB
13.0 dB
12.0 dB
11.0 dB
10.0 dB
0.0
5.0
10.0
15.0
20.0
25.0
30.0
35.0
40.0
45.0
50.0
Fd [Hz]
Figure 6: Burst size influence
4
page 21 (21)
Ericsson laboratory tests
The report of the Ericsson laboratory tests is extracted from a Master Thesis report. The scope of
the Master Thesis was to implement an algorithm for ICI cancellation in an FPGA, and measure the
performance. Special attention was paid to the possibility to use a (non-rectangular) window
function in case the delay spread of the channel was (considerably) smaller than the GI, to further
enhance the performance.
Concerning the theory, reference is therefore made to a paper that has been accepted for
presentation at VTC 2006 Fall [4]
4.1
Implementation
In this chapter the implementations of the three different units needed for effective cancellation, in
the DVB-H prototype, are described. Starting with a short introduction of the basics of design
strategy and hardware design methodology. This is followed by an overview of the design and a
more detailed description is then presented along with simulation results explaining the choices
made during implementation.
4.1.1
System Design
When designing a chip, design strategy is an important issue, especially when the system has
many design constraints. The major design constraints are functionality, speed, area, power
consumption, and cost. The goal of the designer is to find an optimal point in the design space, i.e.,
sufficient throughput at minimum power consumption and area requirement while keeping the
functionality intact. However, optimizing the design takes a great deal of time and work effort and
thus cost starts to increase rapidly. Therefore a designer, to keep the costs at reasonable level,
often settles for a sufficiently good point in the design space which meets the requirements.
In this thesis work, the most important criterion is to prove functionality for the ICI reducing
algorithm. The power aspect of the design is more or less disregarded. There are consequently no
power saving measures taken, such as turning off inactive parts or running a high speed design at
low voltage and speed. Area and speed, on the other hand, could not be disregarded completely.
First, the design had to fit the FPGA without causing routing or speed problems. Second, the
design also had to meet the speed requirements of the receiver and be able to finalize all
calculations for one OFDM symbol before the next arrives. The cost being time as the design had
to be verified during the thesis work.
4.1.2
Design Methodology
In Figure 7, a simplified design flow is depicted, where the major design steps are shown. First, a
floating point model is derived according to specifications. When functionality has been verified, a
fixed point model is implemented. These models are implemented in a suitable programming
language, e.g. Matlab for floating point and C/C++ for fixed point model. With the fixed point model,
it is possible to determine word lengths needed to meet design constraints. From these models test
vectors are extracted, which are to be used in later steps for verification.
During the second step, the system’s design criteria have to be taken into consideration as the
architecture will depend on these. For example, if there is an area constraint, a time-multiplexed
architecture would be preferred whereas, if high throughput is desired, a parallel architecture would
be beneficial. The architecture is described in a hardware description language, such as VHDL or
Verilog. After the design is implemented, its functionality is verified with the previously generated
test vectors.
page 22 (22)
Figure 7: Schematic illustration of the design flow
Next, the design is synthesized into a netlist of components. These components are, in most
cases, taken from a standard library containing arithmetic, logic, and registers. However, in some
cases, the need to implement custom components can arise when the design constraints are hard
to meet. The netlist is then downloaded to an FPGA and the functionality can be verified.
Every step in the design flow must often be iterated several times before a fully functional design,
that reaches all requirements, has been derived.
For this design there was no fixed point model derived, as the cost probably would be too high, i.e.,
too time consuming. The floating point model implemented in Matlab was used for verification.
4.1.3
Overview
Figure 8 is an overview of the top level. Data flows in one direction and is processed by one
algorithm after the other. Output samples from different blocks are accompanied by a valid signal
and are in most cases assumed to be correctly received by the following block. However, there are
exceptions from this, for example, memories which use an acknowledge signal when receiving a
sample. The different blocks have been designed with this in mind.
As the design should be incorporated with an existing DVB-H receiver implementation, wrappers
were developed for the different blocks containing interfaces to the surrounding blocks. This
allowed fairly independent development of the different parts.
The transmission parameters most likely to be used for DVB-H in Sweden are 8K transmission
mode and 16-QAM. Further, the carrier frequency will be located in the TV frequency band (470862 MHz) and probably use an 8 MHz channel. Only the 8K mode is implemented in the prototype
ICI canceller. The major reason for this is that only this mode is implemented in the DVB-H
prototype. Also, only a CP of ¼ is considered.
page 23 (23)
Figure 8: Schematic overview of the design. Grey blocks were implemented during the
Thesis work
4.1.3.1
Window Unit
To reduce the ICI level, the distortion free part of the CP is used for windowing, as described in [4].
The unit applies one of four different window functions available to the incoming OFDM symbol.
The complex baseband samples from the CP are stored in a memory and are then weighted
together with the corresponding part at the end of the symbol. The result is then passed on to the
FFT. It is possible to apply one of four different stepped window functions with equal step lengths
and heights and a maximum window length corresponding to 75 % of the longest CP.
4.1.3.2
Channel Estimator
Doppler shift severely degrades the performance of a channel estimator operating in time direction.
For a system using 8k FFT and CP = ¼, the Nyquist frequency is at 111.6 Hz.
As the existing channel estimator in the DVB-H prototype receiver implemented time then
frequency interpolation, a new channel estimator was developed which only operates in frequency
direction to be able to handle higher Doppler frequencies. This, on the other hand, results in a
receiver with worse performance regarding channels with large delay spreads.
Seven different cubic (third order filter with four filter taps) interpolation filters are implemented. Six
are Wiener filters and one is a Lagrange filter. The results from incoming OFDM symbols
processed by the FFT are stored in a memory which the channel estimator reads pilots from. With
these pilots, interpolation is performed and the resulting estimates are stored in another memory
from where the ICI canceller then fetches the channel estimates.
4.1.3.3
ICI Canceller
The unit that requires most computational resources is the ICI-canceller, as divisions have to be
performed to retrieve the sent data symbols needed in the cancellation. The current implementation
tries to cancel the ICI introduced to a sub-carrier from the two neighbouring sub-carriers. The
design also incorporates the channel equalizer. The unit is implemented using a pipelined structure
to achieve high throughput, and this also results in fairly simple control logic.
4.2
Detailed Description
The following sections give a detailed description of the different implemented blocks. There is also
simulation results presented along with discussion of choices made during implementation. All
simulations are made using a cubic Lagrange interpolation filter in the frequency direction only and
windows of 1563 samples length, unless otherwise stated. The simulations are made using 8K
mode with a CP = ¼, 16-QAM, and a convolutional code rate of 1/2. The byte error rates are
calculated after the Reed-Solomon decoder.
4.2.1.1
Window Unit
As the window is located prior to the FFT, special care must be taken when performing calculations
regarding operations such as rounding. If there is a bias in the rounding this will be accumulated as
a DC offset and might cause overflow and error propagation to other frequencies. Due to this
page 24 (24)
effect, all calculations performed in the window unit is rounded to nearest even, i.e., perform a
rounding that is symmetric around zero.
As described in [4],, the task of windowing is to combine weighted samples from the CP with the
corresponding weighted samples at the end of the OFDM symbol.
Figure 9: Schematic layout view of the window unit, where the RTNE denotes round to
nearest even
As seen in Figure 9, the window unit consists of three major parts, a memory, a calculation unit,
and control logic. The length of the window can be set from N = ¾ CP down to N = 0. The first
quarter of the CP will never be used due to distortion from the channel impulse response. If the
channel impulse response is estimated to be longer than a quarter of the CP, the window length
can be shortened as described above. The estimation of the impulse response length is outside the
scope of this thesis work.
The window unit actually implements all three transmission modes, although only the 8K option is
used. The only difference between a window in 2K and 4K is the length. In 2K the CP is one
quarter of the length of the corresponding CP in 8K, and half the length in 4K. Therefore, the
window length and step size only has to be divided by two (4K) or four (2K) and this is easily
performed by a right bit shift.
From system simulations of the complete receiver, see Figure 10 and Figure 11, it looks like the
impact of the number of steps is not that large. It rather seems that more steps in the window
worsen the performance of the receiver. Despite of this, windows with 1-4 steps with equidistant
step sizes and heights are implemented.
C/N = 25dB
Brickwall
1 step
2 steps
3 steps
511 steps
Triangle
−1
Byte error rate
10
−2
10
160
170
180
190
200
210
220
230
240
Doppler (Hz)
Figure 10: Simulation results for different window functions when no ICI cancellation is
performed
page 25 (25)
The window unit starts to fill the memory with the incoming complex data samples where the real
part is stored in the high bits and the imaginary in the low bits. It continues to do so until it receives
a sync signal, i.e., the first sample in the OFDM symbol r(0) arrives.
In 8K mode using a CP of ¼ there are 10240 samples arriving during the OFDM symbol duration
1.12 ms. As the receiver runs on a 36~MHz clock, this results in approximately one sample every
fourth clock cycle. For this reason, the window calculations are performed in a time multiplexed
fashion. First sample $r(-n)$ is retrieved from the memory and the real part of $r(-n)$ and $r(N-n)$
is then multiplied with the corresponding weight, $w(-n)$ and $w(N-n)$. The results are added
together, rounded to the nearest even, and stored in a register. The same operation is then
performed on the imaginary part and the resulting windowed sample, $r_{w}(N-n)$, is sent to the
FFT. The weights of the different available window functions are located in a small LUTaccessible
by the controller.
C/N = 25dB
1 step
2 steps
3 steps
511 steps
Triangle
−1
Byte error rate
10
−2
10
210
220
230
240
250
260
270
280
Doppler (Hz)
Figure 11: Simulation results for different window functions when interference from one
sub-carrier on each side is cancelled
4.2.1.2
Channel Estimator
During channel estimation the interpolated values in the central interval of the interpolation
polynomial, i.e., 0 < t <= 1, are the interesting ones. As the interpolation is performed in a discrete
(digital) environment t = k/M, where M is the number of interpolation points in the interval, thus k
takes the integer values from 1 to M. Knowing this, the interpolation can be viewed as a discrete
filter, and this task can be solved by a FIR filter. Therefore, the channel estimation can be
performed by a simple structure, see Figure 12, using a FIFO, a FIR filter, a LUT with filter
coefficients, and control logic. Further, it can be seen that the unit is located between two
memories, one containing the received samples, R, and the second one where the estimated
channel responses for different frequencies, H, are stored. In these memories, the samples are
stored in the same fashion as in the window unit with the real part in high bits and imaginary in low.
Figure 12: Schematic layout view of the channel estimator unit
page 26 (26)
The DVB-H standard provides pilot signaling. Of the two different types of pilots the most common
is the scattered pilots, which are evenly distributed in the time frequency grid. The scattered pilots
are separated by 12 steps in the frequency direction and four in the time direction in a staggered
fashion. This dense and even distribution makes the scattered pilots suitable for channel
estimation. As explained before, to benefit from ICI cancellation a channel estimator that operates
in frequency direction first is needed. Therefore, no virtual pilots will be calculated in between the
actual ones, but instead the channel estimation is performed by directly interpolating in the
frequency direction.
Interpolation FIR filter
The implemented cubic FIR interpolation filter is depicted Figure 13. The four filter taps use
complex multipliers and adders as all calculations are complex valued. Prior to the FIR filter a FIFO
of length four is located. These registers hold the four scattered pilots, P, used during the filtering
process.
Figure 13: Schematic view of the cubic interpolation FIR filter
The interpolation coefficients are stored in a LUT and this makes the structure versatile as it is
possible to realize different filters by changing the coefficients. In the current implementation, the
LUT contains filter coefficients for linear filters of lengths M=3, 6, 9, and 12, a cubic Lagrange filter
of length M=12, and six cubic Wiener filters of length M=12.
To calculate H one should use ¾ * P as the pilots are sent with boosted power level. This is,
however, not done as it only results in a scale factor and this is instead handled later in the receiver
chain.
Memories
Both memories R and H, see Figure 12, are addressed with row and column, where a column is of
the same length as the information part of an OFDM symbol. For 8K transmission mode there are
6817 sub-carriers, in one OFDM symbol. Instead of using two-port memories for simultaneous read
and write, one-port memories with a two-port wrapper are used. The wrapper translates the row
and column addresses for the memory and handles a simultaneous read and write request
according to a priority rule, where one port has higher priority than the other which has to wait. The
reason for using one-port memories with a wrapper instead of two-port memories is that a two-port
memory is considerably larger and consumes more power than a single port memory.
As there are three units which need access to the R memory, the channel estimator and the ICI
canceller share one port and the FFT uses the other one. This works fine as ICI cancellation is not
performed until channel estimation is finalized.
In the current implementation, the H memory holds the estimates from three OFDM symbols as this
is the number of symbols needed to perform the estimation of the channel derivative, H’.
Controller
The task of the control logic is to retrieve the pilot values from the memory containing the received
OFDM symbols, feed the FIR filter with the appropriate coefficients and write the result to the
memory containing the channel estimates, H. These tasks are performed by two controllers; one
that fetches the pilots from the R memory and another that handles the coefficients for the FIR filter
and writes the result to the H memory.
The first controller addresses a unit in the receiver to determine if a sample in the symbol is a pilot
or not. This unit contains counters for the two different pilot types which are decremented for every
frequency that is stepped through. When a counter reaches zero, the unit flags the event, and thus
telling when a certain pilot is reached. For the scattered pilots the counter is set to either 3, 6, 9, or
12 at reset and every time it becomes zero it resets to 12. With this information the controller can
retrieve the pilots and switch the sign if necessary and store it in the FIFO. As the FFT has the
page 27 (27)
highest priority to the memory the channel estimator controller has to stall until the memory is
available, when a collision occurs. Now the controller signals which filter to use, starts the second
controller and continues to step through the frequencies to retrieve the next pilot. At the beginning
of every OFDM symbol the FIFO is empty and the first pilot is only switched in and sent to the H
memory. When the second pilot arrives linear filtering is started, the length of the filter, either 3, 6,
9, or 12, depends on where the first scattered pilot is located in the symbol. For the third pilot either
a linear filter of length 12 or a cubic filter of length 12 is used. The same applies for the last three
pilots in the OFDM symbol, for all pilots in between a cubic filter will be used. Which cubic filter to
use is determined by the user, i.e., external control signals. The second controller fetches the
interpolation coefficients, q, from the LUT according to the filter used, and writes the result to the H
memory.
4.2.1.3
ICI-Canceller
The ICI cancellation unit and the equalizer are closely interlinked, see Figure 14 and Figure 15, and
are implemented as, more or less, one unit.
Figure 14: Schematic overview of ICI cancellation unit and channel estimator
From system simulations, it is seen that the number of neighbouring frequencies canceled has a
low impact on the performance. Therefore, the major benefit from ICI canceling is achieved by
removing the interference from the closest sub-carriers. For this reason the ICI cancellation unit
only tries to remove the interference from the closest sub-carrier on each side of the active carrier.
Figure 15: Detailed schematic view of ICI cancellation unit
The pipelined structure of the unit can be seen in Figure 15, where the samples are shifted into
registers and the result then propagates through the design. FIFOs of different lengths are used to
ensure that data and control signals arrive at the correct time. This structure makes it easy to
extend the implementation to cancel the ICI from more sub-carriers.
Controller
The ICI cancellation unit controller has a very basic control sequence as the design is pipelined.
Most of the work consists of handling memory locations for all the different samples needed in the
ICI calculation. When a calculation is started the data and control signals ripple through the design
and FIFOs of different length are used to ensure correct timing. When the unit receives a go signal
page 28 (28)
from the channel estimator the controller starts to retrieve channel estimates and received data
samples.
The unit has to stall if the FFT writes to the R memory as the FFT has the highest priority to this
memory and the sequence therefore takes four or five clock cycles to complete. If the current subcarrier contains a continual or scattered pilot, instead of estimating the sent data symbol, the actual
known value is switched in to the corresponding FIFO.
The controller also retrieves the modified folding factor coefficients, to be used in the calculations
according to the window function used. These modified coefficients are stored in a LUT and cover
the same window types and lengths as the window unit.
Derivative estimation
To estimate the derivative, or rather the change of the channel during the OFDM symbol, it is
assumed that the channel has changed in a linear fashion from the previous symbol to the next.
The channel estimates for the previous symbol and the next one is therefore used to estimate the
derivative, see [4] for a more detailed description.
ICI calculation
The ICI is calculated using the (possibly) estimated signal sent on a sub-carrier, the estimated
channel change, and a weighting function, which depends on what window shape is used. See
[4]for a more detailed description.
4.3
Measurements
This chapter begins with a short description of the laboratory equipment used for measurements in
followed by a section where the actual results from the measurements are presented. Two different
measurements were performed, one with a pure frequency offset applied to the signal and one
using a time-varying channel.
4.3.1
Laboratory Setup
The digital DVB-H baseband prototype decoder runs on an Altera Stratix FPGA, the laboratory
setup is depicted in Figure 16. In addition, a second board containing an Ethernet circuit is
connected to the main FPGA board. This allows for Ethernet signaling at 100 Mbps with a PC.
To generate the radio frequency signal needed for measurements, an I/Q modulation generator,
AMIQ-B3 model 04, together with a vector signal generator, SMIQ03, are used. Both instruments
are from Rhode & Schwartz. The tuner board consists of a Philips TU1216 tuner module and a
discrete analog to digital converter. From the PC, via a RS232 serial interface, it is possible to set
control bits in the receiver and ICI canceling units, e.g. which interpolation filter to use, the number
of steps in the window and also the length of the window.
Figure 16: Schematic overview of the laboratory setup used for measurements
Reference information used for measurements is generated in Matlab. This reference sequence is
then transferred to the AMIQ, which plays the sequence repeatedly. The SMIQ converts the
baseband signal to radio frequency, performs channel modulation and adds noise. The radio
page 29 (29)
frequency signal is then translated to intermediate frequency by the tuner board, which is fed to the
DVB-H receiver. The output from the receiver is sent to the PC over the Ethernet interface and is
compared to the reference sequence.
The functionality of the receiver has also been verified by real-time reception of a live DVB-H
broadcast transmitted by Sony Ericsson. This was done using the radio part of a DVB-T set-topbox from Nokia.
4.3.2
Measured Results
Two different measurements have been performed. The first one where a frequency offset was
introduced to the signal and a second one with a time-varying channel causing Doppler spread.
The measurements are performed using 8K mode with a CP of ¼, 16-QAM, and a convolutional
code rate of ½. The byte error rates are calculated after the Reed-Solomon decoder.
4.3.2.1
Frequency Offset
The frequency offset is imposed by multiplying the time domain signal by an exponential. A
frequency offset can be viewed as a special case of a time-varying channel, where the variation
consists of a phase change. The theoretical gain than can be obtained by using a one-step window
with and without ICI cancellation is shown in Table 10. Excess window length does in this table
refer to how much of the GI that is used for the windowing.
Table 10: Ideal improvement by using a one-step window with different lengths with and
without ICI cancellation
No ICI canceling
Canc. one tap on each side
Excess window length: 0
0.00 dB
4.07 dB
Excess window length: N/32
0.41 dB
5.13 dB
0.84 dB
6.21 dB
1.73 dB
8.16 dB
Excess window length: 3N/16
2.65 dB
9.12 dB
The corresponding measured gains obtained in case of a frequency offset are shown in Table 11.
Comparing with the ideal gains presented in Table 10, it can be seen that the gain obtained by
employing a window function agrees very well with the theory, whereas the gain obtained by
canceling is considerably smaller.
As an example, using a 1024 samples long one-step window, this would ideally lead to an
improvement of 1.73~dB in 8K mode. As the ICI is proportional to the frequency offset squared, an
increase of the frequency offset by about 22% would be possible to achieve by using such a
window.
Table 11: Measured improvement in dB by using a one-step window with different lengths
with and without ICI cancellation when a frequency offset is applied
No ICI canceling
0.00 dB
1.28 dB
0.82 dB
2.19 dB
1.50 dB
2.77 dB
Excess window length: 3N/16
2.16 dB
3.20 dB
In Figure 17 through Figure 21, the measured performance in compared for different choices of
window function with and without ICI cancellation. Since the MPE-FEC is not implemented, the
performance is determined as the byte error rate at the output of the RS decoder. The line at 6.1%
byte error rate at the output of the RS decoder, corresponds to MPE-FEC error rate of 5% if the
rate of the MPE-FEC is ¾ and it is assumed that the errors are independent. Of course, one should
page 30 (30)
not use the figures to compare with other implementations where the MPE-FEC is implemented,
but merely to estimate how much larger frequency error can be handled.
C/N = ∞ dB
0
10
Byte error rate
Brickwall
Brickwall, ICI cancel
1 step, 1024 samples long
1 step, 1024 samples, ICI cancel
−1
10
−2
10
180
200
220
240
260
280
Frequency error (Hz)
300
320
340
360
Figure 17: Measured result when a frequency offset is applied to the signal using brickwall
and one-step window with and without ICI cancellation
C/N = ∞ dB
0
10
Byte error rate
Brickwall
−1
10
−2
10
180
200
220
240
260
280
300
320
340
360
Figure 18: Measured result when a frequency offset is applied to the signal using different
lengths in a one-step window and no ICI cancellation is performed
page 31 (31)
C/N = ∞ dB
0
10
Byte error rate
Brickwall
−1
10
−2
10
220
240
260
280
300
320
340
360
lengths in a one-step window and the ICI from the two closest neighbouring sub-carriers are
cancelled
C/N = ∞ dB
0
10
Byte error rate
Brickwall
1 step
2 steps
3 steps
4 steps
−1
10
−2
10
180
200
220
240
260
280
300
320
340
360
number of steps in a 1536 samples long window function and no ICI cancellation is
performed
C/N = ∞ dB
0
10
Brickwall
1 step
2 step
3 step
4 step
−1
Byte error rate
10
−2
10
−3
10
220
240
260
280
300
320
340
360
page 32 (32)
number of steps in a 1536 samples long window function and the ICI from the two closest
neighbouring subcarriers is cancelled
4.3.2.2
Doppler Spread
The theoretical improvements (in dB) by using different lengths of a one-step window with or
without ICI cancellation are the same for time-varying channels as for frequency offset. The
corresponding measured results are given in Table 12.
Table 12: Measured improvement in dB by using a one-step window with different lengths
with and without ICI cancellation in a time-variant channel causing Doppler spread
No ICI canceling
0.00 dB
1.22 dB
0.46 dB
1.67 dB
1.77 dB
3.12 dB
Some measured results are given in Figure 22 through Figure 25 for a TU6 channel.
−2
Byte error rate = 6.1 ⋅ 10
28
Brickwall
27
26
25
24
Required C/N (dB)
23
22
21
20
19
18
17
16
15
14
0
20
40
60
80
100
120
Doppler (Hz)
140
160
180
200
Figure 22: Measured result when a Doppler spread is applied to the signal using different
lengths in a one-step window and no ICI cancellation is performed
−2
28
Brickwall
27
26
25
24
Required C/N (dB)
23
22
21
20
19
18
17
16
15
14
0
50
100
150
200
250
Doppler (Hz)
page 33 (33)
lengths in a one-step window and the ICI from the two closest neighbouring subcarriers is
cancelled
−2
28
Brickwall
1 step
2 steps
3 steps
4 steps
27
26
25
24
Required C/N (dB)
23
22
21
20
19
18
17
16
15
14
0
20
40
60
80
100
120
Doppler (Hz)
140
160
180
200
number of steps in a 1024 samples long window function and no ICI cancellation is
performed
Byte error rate = 6.1 ⋅ 10−2
28
Brickwall
1 step
2 steps
3 steps
4 steps
27
26
25
24
Required C/N (dB)
23
22
21
20
19
18
17
16
15
14
0
50
100
150
200
250
Doppler (Hz)
number of steps in a 1024 samples long window function and the ICI from the two closest
neighbouring subcarriers is cancelled
4.4
Conclusions and Improvements
First, in this section, conclusions are drawn and problems encountered during the thesis work are
listed. Then, suggestions for improving the design regarding complexity, functionality, power
consumption, and speed are given.
4.4.1
Conclusions
During this thesis, most problems that have arisen are related to timing of either data and/or control
signals. It gets especially problematic when there are a lot of data being shuffled between
memories, as in this design, which requires many control signals. Finding what causes these
problems can in many cases be very time consuming and a major part of the design time is spent
on implementing and testing controllers.
page 34 (34)
Even tough wrappers were used and the interfaces to the surrounding blocks in the DVB-H
receiver were, more or less, ready the incorporation of the units encountered more problems and
consumed more time than expected. One reason for this was that control signals regarding pilot
positions were for a channel estimator working in time direction first and was slightly askew for the
new estimator working in frequency direction only.
As this thesis has shown, it is possible to implement an ICI canceling algorithm at low complexity
and achieve good performance by using a window function and cancellation of ICI on one subcarrier from two adjacent sub-carriers. The complexity can be reduced further, by implementing
some of the suggestions in the next section. The functionality of the ICI canceling algorithm has
been proved using both frequency offset and a time-variant channel causing Doppler spread, with
good results.
It is seen that the best result is achieved by using a one-step window function and measurements
show that the performance is very close to the theoretical improvement. This gives that an
improvement up to 2.65dB, i.e., 35% higher Doppler frequency, is possible to reach using only a
one-step window function of 1536 samples length. This is a very satisfying result as the simple unit
works close to its maximum performance.
According to theory, ICI cancellation should give a large improvement of 4.07dB when the ICI from
two adjacent sub-carriers are cancelled. Unfortunately, the measured performance of the
implementation is far lower, about 1.2dB, this is, however, still an improvement of 15% and higher
than the expected outcome from simulations. The reduced performance is probably due to that the
sent symbol and channel change are estimated. Thus, to improve the performance of the ICI
cancellation, the quality of these estimated values has to be improved, and as both values depend
on the quality of the channel estimates, the channel estimation has to be improved. Further, it
seems that the performance of the ICI cancellation is increased when used together with a window.
This as the channel estimates become better when there is less ICI remaining. This increase in
performance is, however, only seen when using a channel causing Doppler spread and not when a
frequency offset is applied, where the performance is actually decreased when the longest window
is used.
Using both a long window and ICI cancellation an improvement of about 55 % is expected and
Doppler frequencies well above 200Hz can be handled without a too high required C/N, which
correspond to a receiver speed of at least 250km/h at a carrier frequency of 862
4.4.2
Improvements
In a mobile terminal, where a DVB-H receiver is likely to be found, power consumption is one of the
most important design criteria. In most cases, lower power consumption comes at the expense of
larger chip area. As this implementation more or less disregarded the power aspect, there are quite
a lot to improve.
Since this prototype was implemented to prove functionality, no word-length optimization was
performed. Therefore the area and the power consumption of the design could be reduced by
reducing the word-lengths. The channel estimate uses 16 bits and after the channel equalization a
resolution of six bits is believed to be sufficient, so a word-length of 8-10 bits will probably be
enough. The large word-length is to ensure that noise is caused by the transmission and not
truncation errors. All word-lengths can easily be changed by a constant in a configuration file.
All complex multiplications
BD)+j(AD+BC).
are
implemented
straightforward
as
E=(A+jB)
(C+jD)=(AC-
This multiplication could be optimized by using a distributed arithmetic approach and give a
significant improvement in both speed and power consumption.
As discussed, a real implementation should cover all three transmission modes. The parts that
have to be extended are the channel estimator and the ICI cancellation unit. This is on the other
hand easily obtained, with some additional logic in the controllers, as the only difference between
2K, 4K, and 8K is the length of the OFDM symbol. Pilots have the same location in all three modes.
Therefore, adding logic that handles different symbol lengths is the only thing necessary to make
the ICI canceling compliant with all three transmission modes. In the ICI unit the folding factors are
scaled. This scale factor depends on the different modes and CP used. To solve this, the derivative
estimate can be scaled according to the mode and CP used with a multiplication or have different
sets of coefficients.
4.4.2.1
page 35 (35)
Window Unit
As seen, the best performance is reached using a one-step window function and that the
performance is close to the theoretical limit. By only implementing a one-step window the window
unit could be simplified, this as no step sizes has to be calculated and the weight would be ½, i.e.,
a right bit shift. It will also reduce the size of the LUT containing coefficients in the ICI canceller.
In the current implementation, the length of the window is set by external control signals. To make
this unit independent the length of the channel impulse response needs to be estimated.
If the window is placed to early, the output from the FFT will be phase rotated and this will affect
the channel estimate and thereby the ICI cancellation. The window position optimization is, in the
current implementation, done manually, i.e., there is a need to implement adaptive control for the
window position.
4.4.2.2
Channel Estimation
As seen, the improvement from using ICI cancellation is far from the theoretical. One reason for
this is that the sent values, S, has to be estimated and to improve the estimates performance of the
channel estimator has to be improved. This can possibly be achieved by longer interpolation filters
and implementing this can be done by adding more filter taps to the interpolation filter and at the
same time increase the length of the FIFO holding pilots. Additional control logic is also necessary
to decrease the order of the interpolation filter at the edges of OFDM symbols.
There is also a possibility to apply a filter in the time direction to increase the tolerance toward
delay spread.
Another way of improving the tolerance towards delay spread is to implement pre-rotation. Delay
spread will have the same effect on the signal as a too early placed FFT window, i.e., a rotation of
the different sub-carriers dependent on their position. In pre-rotation the output from the FFT is
rotated backwards according to the estimated rotation caused by delay spread and/or incorrect
window placement. The rotation can be estimated by summation of phase differences between
pilots from which the rotation can be calculated.
4.4.2.3
ICI Unit
As seen in Figure 14, and earlier mentioned, the design uses two equalizing units. Remembering
the control sequence used, by the ICI controller it is noted that the first equalizer only starts a
calculation every fourth (or fifth) clock cycle. It would therefore be possible to use only one
equalizer to perform both divisions. This improvement would decrease the area of the ICI
cancellation unit significantly, approximately 25% as the two equalizers make up for about 65% of
the total area in the ICI canceller.
The pipelined structure of the ICI canceller makes it possible to start a new calculation every clock
cycle. This makes the H memory a bottle neck as there are three different values that need to be
fetched, and this takes three clock cycles. One possible way to speed up this is to have three small
memories containing one symbol each instead of one large holding all. Thus, enabling access to all
three samples in one clock cycle and a new calculation could be started every cycle, unless the R
memory is accessed by the FFT and one has to stall until the memory becomes available. This
would speed up the calculations considerably and the unit would be inactive longer, thus saving
power. Combining this with the improvement proposed above, one could start a new ICI calculation
and channel equalization every other clock cycle.
page 36 (36)
5
TeamCast lab tests
5.1
Purpose of the test
Teamcast with the DiBcom collaboration would like to experiment in its labs in Rennes, the
performances of a transmission using DVB-H with a bandwidth shrunk to 1.75 MHz instead of 8
MHz. The purpose of this internal test is to evaluate the DVB-H in comparison to DAB (T-DMB) in
the same configuration.
DAB transmission uses a 1.536 MHz signal bandwidth broadcast in a 1.712 MHz channel. To
adapt the DVB-H signal in such channel, while optimising channel usage and broadcast bitrate, the
“7 MHz bandwidth” DVB-H mode has been selected as a starting point. The 7 MHz system clock (8
MHz) has been divided by four (2 MHz) in order to produce a DVB-H signal having a 1.66 MHz
bandwidth, fitting in the 1.712 MHz DAB channel.
Accordingly, the effective DVB-H bandwidth is 1.66MHz instead of the DAB 1.536MHz.
The performed tests have tried to evaluate the impact of the bandwidth shrinking in 3 domains:
1. Impact on the maximum Doppler,
2. Impact on the C/N,
3. Impact on the receiver consumption,
5.2
Test bench presentation
The automatic test bench is very close to the test bench built in Rennes session. It mainly consists
of an IP Encapsulator, a DVB-T/DVB-H modulator, a channel simulator and a receiver under test.
All equipments are controlled by a computer running a special software.
Teamcast and DiBcom have modified respectively the modulator and the receiver to accept the
new bandwidths.
5.2.1
Synoptic
The test bench is composed of five main elements:
•
A Sidsa DVB-H IP encapsulator. It performs two tasks. It plays a video file to generate
an IP stream with VLC software. Then this IP stream is converted to a MPEG-TS stream
adding MPE-FEC table and using time-slicing. MPE-FEC code rate, table rows and time
slicing parameters are fully tunable. Output of this IP encapsulator consists of an ASI
link. This equipment is not controlled by the test bench software. Manual operations are
required to set IP encapsulator specific parameters such as time slicing or MPE-FEC
code rate.
•
A Teamcast modulator. It is fully tunable for DVB-T and DVB-H modulation. It allows
setting bandwidth, FFT size, native or indepth interleaver, guard interval, code rate or
constellation. Input consists of an ASI link provided by the IP encapsulator. Outputs
consist of two analog signal in band de base, I and Q. The modulator has been modified
to support 1.75MHz, 3.5MHz, 5MHz, 6MHz, 7MHz and 8MHz channel bandwidth.
•
A Rohde&Schwartz SFU channel simulator. It includes a fading simulator, a noise
generator and a Doppler frequency shift generator. Frequency, bandwidth, output power
and C/N are finely tunable. For tests, standard channel profiles can be applied: AWGN,
Rice, Rayleigh or TU6 for example. Other custom profiles can be fully defined. For
mobile tests, a Doppler frequency shift can be added to output signal. This equipment
outputs a ready to use signal at 666 MHz at an output power about -40 dBm.
•
A DiBcom receiver under test. It is able to handle either DVB-T or DVB-H signal.
Monitoring parameters are obtained USB interfaces. Parameters include FER (Frame
Error Rate) and MFER (MPE Frame Error Rate). The receiver has been modified to
accept different channel bandwidth (1.75 to 8MHz).
•
page 37 (37)
A computer runs a software written with LabView allows setting modulation parameters,
propagation channel characteristics and getting monitoring information. This software
implements the measurements method described bellow.
DVBDVB-H
IP encapsulator
(SIDSA)
TS
DVBDVB-H / DVBDVB-T
Modulator with BW shrunk
(TeamCast)
TeamCast)
I/Q
Channel Simulator
(Rohde & Schwartz)
RF (666MHz)
WING
TV
WP4
TEST
BED
Control
Software
(Thales
&
TeamCast)
TeamCast)
DVBDVB-H Receiver
Custimized with BW shrunk
(DiBcom)
DiBcom)
Figure 26: Test bench synoptic
5.2.2
Method
The measurement method is illustrated by the fig 3. Four points have to be determined to build the
characteristic curve. The automatic test bench processes as following:
1. The C/N is set at a high value (40dB). The Doppler frequency is adjusted to reach the
FER5/MFER5 criterion. This allows determining the first point.
2. The Doppler frequency get in (1) is divided by two and fixed, and then the C/N is adjusted
in order to reach the FER5/MFER5 criterion.
3. The C/N level get in (2) is increased by 3dB and the Doppler frequency is adjusted until
reaching FER5/MFER5 criterion.
4. Finally, the last point is get by setting frequency Doppler at few hertz and the C/N level is
adjusted to reach the FER5/MFER5 criterion.
page 38 (38)
The same process is run twice, once to get the DVB-T curve with FER5 criterion, the second time
to get the DVB-H curve with MFER5 criterion.
DVB-T / DVB-H
Mobile Performance
(C/N versus Doppler)
DVB-T @ FER5
DVB-H @ MFER5
40
1
30
2
4
20
3
10
0
1
10
TU6
100
1000
Figure 27: Mobile measurement result (sample)
Test process is quite long: the FER5/MFER5 criterion requires waiting at least 40 frames to be
measured. Getting a curve takes roughly 1 hour either manually or with an automatic test bench.
Precision on these measures is 0.5 dB for C/N and 5Hz for Doppler frequency shift.
All tests have been performed at 666MHz in single antenna reception (no diversity) with a constant
input power of -40dBm.
5.3
Results
For all measurements, the IP encapsulator is configured with rows=512 and FEC=3/4.
5.3.1
Mode 2K
Table 13: Numerical values for BW=1.75MHz and FFT=2K in DVB-T (FER criteria)
FER
FFT
Guard
InDepth
Constellation
Code Rate
Doppler 5
Doppler 6
Doppler 7
Doppler 8
C/N 5
C/N 6
C/N 7
C/N 8
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
1/32
1/4
1/4
1/16
1/32
1/4
1/4
1/16
1/32
1/4
1/4
1/16
1/32
1/4
1/4
1/16
1/4
1/4
1/16
1/32
1/4
1/4
1/16
indepth
indepth
native
indepth
indepth
indepth
native
indepth
indepth
indepth
native
indepth
indepth
indepth
native
indepth
indepth
native
indepth
indepth
indepth
native
indepth
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
3/4
3/4
3/4
3/4
1/2
1/2
1/2
1/2
2/3
2/3
2/3
3/4
3/4
3/4
3/4
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
67,5
55,0
57,5
70,0
67,5
60,0
50,0
60,0
60,0
50,0
37,5
60,0
60,0
55,0
50,0
65,0
47,5
42,5
57,5
47,5
42,5
27,5
50,0
130,0
105,0
105,0
130,0
125,0
100,0
85,0
115,0
115,0
95,0
70,0
115,0
115,0
100,0
90,0
120,0
90,0
80,0
110,0
90,0
80,0
45,0
90,0
135,0
110,0
115,0
140,0
135,0
120,0
100,0
120,0
120,0
100,0
75,0
120,0
120,0
110,0
100,0
130,0
95,0
85,0
115,0
95,0
85,0
55,0
100,0
10,5
11,0
12,5
11,0
13,0
13,5
16,0
14,0
16,5
17,5
18,5
15,0
16,0
15,0
17,5
16,0
19,5
21,0
20,0
23,0
23,0
22,0
22,0
11,0
11,5
13,0
11,5
14,0
14,0
17,0
14,5
18,0
18,0
21,0
15,5
16,5
16,0
18,0
16,5
20,0
23,0
21,0
23,5
23,5
24,0
22,5
14,0
14,5
16,0
14,5
17,0
17,0
20,0
17,5
21,0
21,0
24,0
18,5
19,5
19,0
21,0
19,5
23,0
26,0
24,0
26,5
26,5
27,0
25,5
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
page 39 (39)
Table 14: Numerical values for BW=1.75MHz and FFT=2K in DVB-H (MFER criteria)
FFT
Guard
InDepth
Constellation
Code Rate
Doppler 1
Doppler 2
Doppler 3
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
2K
1/32
1/4
1/4
1/16
1/32
1/4
1/4
1/16
1/32
1/4
1/4
1/16
1/32
1/4
1/4
1/16
1/4
1/4
1/16
1/32
1/4
1/4
1/16
indepth
indepth
native
indepth
indepth
indepth
native
indepth
indepth
indepth
native
indepth
indepth
indepth
native
indepth
indepth
native
indepth
indepth
indepth
native
indepth
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
3/4
3/4
3/4
3/4
1/2
1/2
1/2
1/2
2/3
2/3
2/3
3/4
3/4
3/4
3/4
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
10,0
72,5
60,0
60,0
75,0
70,0
62,5
60,0
72,5
72,5
55,0
60,0
70,0
70,0
60,0
60,0
70,0
50,0
60,0
60,0
65,0
60,0
55,0
62,5
140,0
110,0
110,0
140,0
135,0
105,0
100,0
135,0
115,0
100,0
80,0
130,0
115,0
110,0
100,0
115,0
95,0
110,0
115,0
110,0
70,0
90,0
100,0
5.3.2
MFER
Doppler 4
C/N 1
145,0
120,0
120,0
150,0
140,0
125,0
120,0
145,0
145,0
110,0
120,0
140,0
140,0
120,0
120,0
140,0
100,0
120,0
120,0
130,0
120,0
110,0
125,0
8,5
5,5
9,5
8,0
12,0
11,0
13,0
13,5
13,5
13,5
14,5
14,5
11,0
12,0
11,0
12,5
15,5
17,0
15,5
18,0
18,5
18,5
18,5
C/N 2
C/N 3
C/N 4
7,5
9,0
9,5
9,5
11,0
10,5
14,0
12,5
11,0
14,0
16,0
12,5
12,0
13,5
15,0
13,5
19,0
19,0
16,5
19,0
17,5
20,0
17,5
10,5
12,0
12,5
12,5
14,0
13,5
17,0
15,5
14,0
17,0
19,0
15,5
15,0
16,5
18,0
16,5
22,0
22,0
19,5
22,0
20,5
23,0
20,5
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
Mode 4K
FER
FFT
Guard
InDepth
Constellation
Code Rate
Doppler 5
Doppler 6
Doppler 7
Doppler 8
C/N 5
C/N 6
C/N 7
C/N 8
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
3/4
3/4
3/4
3/4
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
3/4
3/4
3/4
3/4
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
25,0
22,5
27,5
27,5
30,0
22,5
25,0
27,5
27,5
22,5
22,5
27,5
30,0
22,5
25,0
25,0
27,5
22,5
25,0
27,5
22,5
17,5
20,0
20,0
45,0
40,0
50,0
50,0
55,0
40,0
45,0
50,0
50,0
40,0
40,0
50,0
55,0
40,0
45,0
45,0
50,0
40,0
45,0
45,0
40,0
30,0
30,0
30,0
50,0
45,0
55,0
55,0
60,0
45,0
50,0
55,0
55,0
45,0
45,0
55,0
60,0
45,0
50,0
50,0
55,0
45,0
50,0
55,0
45,0
35,0
40,0
40,0
12,0
9,0
11,0
10,5
14,0
14,0
13,5
13,0
16,5
13,5
7,5
16,0
16,5
16,0
16,5
15,0
20,0
19,0
19,5
19,5
19,0
22,0
22,0
24,5
13,0
10,0
11,5
11,0
16,0
14,5
16,0
14,5
17,0
16,0
19,5
16,5
17,5
18,5
17,5
17,5
21,5
19,5
21,0
20,5
26,0
24,0
24,0
25,0
16,0
13,0
14,5
14,0
19,0
17,5
19,0
17,5
20,0
19,0
22,5
19,5
20,5
21,5
20,5
20,5
24,5
22,5
24,0
23,5
29,0
27,0
27,0
28,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
page 40 (40)
FFT
Guard
InDepth
Constellation
Code Rate
Doppler 1
Doppler 2
Doppler 3
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
4K
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/16
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
indepth
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
3/4
3/4
3/4
3/4
1/2
1/2
1/2
1/2
2/3
2/3
2/3
2/3
3/4
3/4
3/4
3/4
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
4,0
37,5
27,5
32,5
30,0
35,0
27,5
27,5
30,0
32,5
27,5
32,5
32,5
35,0
27,5
27,5
30,0
32,5
27,5
30,0
32,5
35,0
27,5
30,0
30,0
65,0
45,0
55,0
55,0
65,0
50,0
50,0
55,0
60,0
50,0
60,0
60,0
60,0
50,0
50,0
40,0
55,0
45,0
50,0
45,0
40,0
35,0
50,0
40,0
5.3.3
MFER
Doppler 4
C/N 1
75,0
55,0
65,0
60,0
70,0
55,0
55,0
60,0
65,0
55,0
65,0
65,0
70,0
55,0
55,0
60,0
65,0
55,0
60,0
65,0
70,0
55,0
60,0
60,0
9,0
6,0
9,5
9,5
11,0
10,0
12,5
12,0
14,5
7,0
6,5
14,0
14,5
14,0
11,0
13,5
12,0
14,5
12,0
14,5
13,0
18,5
14,5
16,5
C/N 2
C/N 3
C/N 4
9,0
9,0
9,5
7,5
13,0
12,5
11,0
12,5
14,0
14,5
19,0
14,5
13,5
13,5
14,5
12,0
16,5
16,5
16,5
14,5
15,0
17,5
17,5
16,5
12,0
12,0
12,5
10,5
16,0
15,5
14,0
15,5
17,0
17,5
22,0
17,5
16,5
16,5
17,5
15,0
19,5
19,5
19,5
17,5
18,0
20,5
20,5
19,5
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
Mode 8K
FER
FFT
Guard
InDepth
Constellation
Code Rate
Doppler 5
Doppler 6
Doppler 7
Doppler 8
C/N 5
C/N 6
C/N 7
C/N 8
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/32
1/4
1/32
1/4
1/16
1/32
1/4
1/8
1/32
1/4
native
native
native
native
native
native
native
native
native
native
native
native
native
native
native
native
native
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
1/2
1/2
1/2
1/2
2/3
2/3
2/3
3/4
3/4
1/2
1/2
1/2
2/3
2/3
2/3
3/4
3/4
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
17,5
15,0
17,5
17,5
17,5
15,0
17,5
17,5
15,0
17,5
15,0
5,0
17,5
12,5
12,5
7,5
5,0
30,0
25,0
30,0
30,0
30,0
25,0
25,0
25,0
20,0
30,0
25,0
5,0
25,0
15,0
20,0
10,0
10,0
35,0
30,0
35,0
35,0
35,0
30,0
35,0
35,0
30,0
35,0
30,0
10,0
35,0
25,0
25,0
15,0
10,0
10,0
10,5
11,0
9,5
13,0
13,0
22,0
16,0
16,0
15,0
13,5
27,0
14,5
19,0
19,5
22,0
21,0
13,0
12,5
11,5
12,5
15,5
15,5
22,5
18,5
17,0
18,0
19,0
30,0
21,0
21,0
21,5
23,0
21,5
16,0
15,5
14,5
15,5
18,5
18,5
25,5
21,5
20,0
21,0
22,0
33,0
24,0
24,0
24,5
26,0
24,5
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
FFT
Guard
InDepth
Constellation
Code Rate
Doppler 1
Doppler 2
Doppler 3
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
8K
1/32
1/4
1/8
1/16
1/32
1/4
1/8
1/32
1/4
1/32
1/4
1/16
1/32
1/4
1/8
1/32
1/4
native
native
native
native
native
native
native
native
native
native
native
native
native
native
native
native
native
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
QPSK
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
16 QAM
1/2
1/2
1/2
1/2
2/3
2/3
2/3
3/4
3/4
1/2
1/2
1/2
2/3
2/3
2/3
3/4
3/4
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
2,0
20,0
17,5
20,0
20,0
20,0
17,5
20,0
20,0
17,5
20,0
17,5
7,5
20,0
17,5
17,5
20,0
17,5
30,0
30,0
30,0
30,0
30,0
30,0
30,0
30,0
30,0
30,0
30,0
10,0
30,0
25,0
30,0
30,0
30,0
MFER
Doppler 4
C/N 1
40,0
35,0
40,0
40,0
40,0
35,0
40,0
40,0
35,0
40,0
35,0
15,0
40,0
35,0
35,0
40,0
35,0
8,0
7,5
6,5
5,5
11,0
11,5
8,5
12,0
11,5
14,0
12,5
14,0
11,0
15,5
16,5
15,5
19,0
C/N 2
C/N 3
C/N 4
9,0
10,5
10,5
7,5
12,5
14,0
14,0
14,5
16,0
12,0
13,5
25,5
15,0
13,5
16,5
19,0
20,5
12,0
13,5
13,5
10,5
15,5
17,0
17,0
17,5
19,0
15,0
16,5
28,5
18,0
16,5
19,5
22,0
23,5
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
40,0
5.4
Analysis of the DVB-H in 1.75MHz bandwidth
5.4.1
Comparison with the 7MHz bandwidth
All measurements are been done with the automatic test bench. Then, some results should no
accuracy, but it is possible to extract some general trends.
The maximum Doppler is approximately divided by 4 in comparison to the 7MHz due to the intercarrier spacing is also divided by 4.
page 41 (41)
In the same line, the guard interval duration is multiply by 4 in comparison to the 7MHz. Then, the
echo delay acceptance is also multiply by 4.
Regarding the Doppler criteria and the SFN criteria (cell size), 2K in BW 1.75MHz seems
equivalent to 8K in 7MHz
One interesting point concerns the power consumption of the receiver. The power consumption is
nearly divided by 4 in comparison to the 7MHz (measured without time slicing – continuous power)
•
With 7MHz bandwidth : 200mW
•
With 1.75 bandwidth : 60mW
5.4.2
Comparison with DAB standard
The following table shows that the mode 2K in DVB-H is similar to the mode I of DAB: the carrier
spacing is approximately 1kHz.
Table 19: Frequency domain parameters for DVB-H in 1.75MHz bandwidth and for DAB
DVB – T/H
DAB
Frequency Domain
Parameters
8K
4K
2K
I
IV
II
III
Channel Bandwidth
1,712 MHz
1,712 MHz
1,712 MHz
1,712 MHz
1,712 MHz
1,712 MHz
1,712 MHz
6 817
3 409
1 705
1536
768
384
192
Number of carriers
K
Value of carrier number
Kmin
Value of carrier number
Kmax
Carrier Spacing
1/Tu
Spacing between carriers
Kmin and Kmax
0
0
0
0
0
0
0
6 816
3 408
1 704
1535
767
383
191
244 Hz
488 Hz
977 Hz
1 000 Hz
2 000 Hz
4 000 Hz
8 000 Hz
1,66 MHz
1,66 MHz
1,67 MHz
1,536 MHz
1,536 MHz
1,536 MHz
1,536 MHz
The two following tables show that the mode EEP 1/2 of DAB should be compared with the mode
QPSK 2/3 of DVB.
Table 20: DVB-H bitrate in 1.75MHz bandwidth
without MPE-FEC
DVB-H bitrate
with MPE-FEC = 3/4
Guard interval
Guard interval
Code rate
1/4
QPSK
16-QAM
64-QAM
1/8
1/16
1/32
1/4
1/8
1/16
1/32
1/2
1,09 Mbps
1,21 Mbps
1,28 Mbps
1,32 Mbps
0,82 Mbps 0,91 Mbps
2/3
1,45 Mbps
1,61 Mbps
1,71 Mbps
1,76 Mbps
1,09 Mbps 1,21 Mbps
1,28 Mbps
1,32 Mbps
3/4
1,63 Mbps
1,81 Mbps
1,92 Mbps
1,98 Mbps
1,22 Mbps
1,36 Mbps
1,44 Mbps
1,48 Mbps
5/6
1,81 Mbps
2,02 Mbps
2,13 Mbps
2,20 Mbps
1,36 Mbps
1,51 Mbps
1,60 Mbps
1,65 Mbps
7/8
1,91 Mbps
2,12 Mbps
2,24 Mbps
2,31 Mbps
1,43 Mbps
1,59 Mbps
1,68 Mbps
1,73 Mbps
1/2
2,18 Mbps
2,42 Mbps
2,56 Mbps
2,64 Mbps
1,63 Mbps 1,81 Mbps
1,92 Mbps
1,98 Mbps
2/3
2,90 Mbps
3,23 Mbps
3,42 Mbps
3,52 Mbps
2,18 Mbps 2,42 Mbps
2,56 Mbps
2,64 Mbps
3/4
3,27 Mbps
3,63 Mbps
3,84 Mbps
3,96 Mbps
2,45 Mbps
2,72 Mbps
2,88 Mbps
2,97 Mbps
5/6
3,63 Mbps
4,03 Mbps
4,27 Mbps
4,40 Mbps
2,72 Mbps
3,02 Mbps
3,20 Mbps
3,30 Mbps
7/8
3,81 Mbps
4,23 Mbps
4,48 Mbps
4,62 Mbps
2,86 Mbps
3,18 Mbps
3,36 Mbps
3,46 Mbps
1/2
3,27 Mbps
3,63 Mbps
3,84 Mbps
3,96 Mbps
2,45 Mbps
2,72 Mbps
2,88 Mbps
2,97 Mbps
2/3
4,35 Mbps
4,84 Mbps
5,12 Mbps
5,28 Mbps
3,27 Mbps
3,63 Mbps
3,84 Mbps
3,96 Mbps
3/4
4,90 Mbps
5,44 Mbps
5,76 Mbps
5,94 Mbps
3,67 Mbps
4,08 Mbps
4,32 Mbps
4,45 Mbps
5/6
5,44 Mbps
6,05 Mbps
6,40 Mbps
6,60 Mbps
4,08 Mbps
4,54 Mbps
4,80 Mbps
4,95 Mbps
7/8
5,72 Mbps
6,35 Mbps
6,72 Mbps
6,93 Mbps
4,29 Mbps
4,76 Mbps
5,04 Mbps
5,20 Mbps
Table 21: DAB bitrate according to mode
0,96 Mbps
0,99 Mbps
page 42 (42)
DAB bitrate
Data
DAB MODE
Code rate
with EEP
I
1/4
DATA with EEP
FIC
TOTAL
MULTIPLEX
IV
0,80 Mbps
II
0,80 Mbps
III
0,80 Mbps
0,80 Mbps
3/8
0,98 Mbps
0,98 Mbps
0,98 Mbps
0,98 Mbps
1/2
1,15 Mbps
1,15 Mbps
1,15 Mbps
1,15 Mbps
5/8
1,44 Mbps
1,44 Mbps
1,44 Mbps
1,44 Mbps
3/4
1,73 Mbps
1,73 Mbps
1,73 Mbps
1,73 Mbps
1/3
0,032 Mbps
0,032 Mbps
0,032 Mbps
0,043 Mbps
1/4
0,83 Mbps
0,83 Mbps
0,83 Mbps
0,84 Mbps
3/8
1,01 Mbps
1,01 Mbps
1,01 Mbps
1,02 Mbps
1/2
1,18 Mbps
1,18 Mbps
1,18 Mbps
1,19 Mbps
5/8
1,47 Mbps
1,47 Mbps
1,47 Mbps
1,48 Mbps
3/4
1,76 Mbps
1,76 Mbps
1,76 Mbps
1,77 Mbps
Moreover, the DVB-H standard offers multitude of combinations of modes regarding the DAB
standard. The guard interval should be reduced to increase the bit rate. The constellation should be
rise too (QPSK to 16QAM).
To finalise this analyse, the mode EEP 1/2 of DAB should be measured with the same condition:
TU6 with the MFER5% criteria.
Table 22: Analyse DVB-H in comparison to DAB in TU6
Standard
DVB
(2K)
GI
1/4
1/8
MFER
QPSK
1/2
10Hz
60 Hz
110 Hz
120 Hz
8 dB
9 dB
12 dB
40dB
QPSK
2/3
10Hz
60 Hz
100 Hz
120 Hz
11 dB
12 dB
15 dB
40dB
16QAM
1/2
10Hz
55 Hz
90 Hz
110 Hz
13 dB
14 dB
17 dB
40dB
16QAM
2/3
10Hz
50 Hz
90 Hz
100 Hz
16 dB
18 dB
21 dB
40dB
QPSK
1/2
10Hz
65 Hz
120 Hz
130 Hz
8 dB
9 dB
12 dB
40dB
QPSK
2/3
10Hz
65 Hz
110 Hz
130 Hz
11 dB
12 dB
15 dB
40dB
16QAM
1/2
10Hz
60 Hz
100 Hz
120 Hz
13 dB
14 dB
17 dB
40dB
10Hz
55 Hz
100 Hz
110 Hz
16 dB
18 dB
21 dB
40dB
10Hz
TBD
TBD
TBD
TBD
TBD
TBD
40dB
16QAM
2/3
DAB
(mode I)
1/4
EEP
1/2
6
page 43 (43)
DiBcom lab tests
DiBcom would like to test and validate the DIB7000 receiver, in pure SFN environment. These
measurements have been performed using NIM7000-SD2 ref. board and the software SDK5.0. The
tuner used is the MC44CD02 Freescale. It gives the performances for various modulation
parameters (constellation, size FFT, code rate Viterbi and guard interval).
6.1
Pure SFN : 2 paths channel
6.1.1
General
In SFN network, signal transporting useful information can come several sources. In order to have
a simple model of this kind of network, the channel used is a 2 paths channel defined by : the delay
and the loss between these 2 paths.
Two kind of tests have been performed, using the same failure criteria : Quasi Error Free. QEF is
the limit, for which BER Ratio is lower than 2e-4 (after Viterbi treatment), and lower than 10e-11 at
the input of the MPEG-2 multiplexer (after Reed-Solomon treatment).
6.1.2
Paths channel without added noise
In this test, we measure for one delay the minimum loss between the 2 paths in order to reach the
QEF.
For DVB-T/H modulation, it provide maximum delay without loss between the two paths, allow to
correctly demodulate the signal (QEF) which is in general, equal or near (superior) to TIG Time
duration of the Guard Interval. In addition, this test allows to know the maximum echo amplitude
possible for a desired delay. The figure below explains this principle.
Figure 28: 2 paths channel curve
In fact, the measurement consists to measure the maximum delay between the 2 paths channel for
a loss of 0dB, while QEF reception is reached. Then, the attenuation between 2 paths channel
increases up to QEF reception is reached again. And so on.
page 44 (44)
6.1.3
Paths channel with added noise
The same measure and methodology is applied for a second test, which describes performances of
receiver when noise added. The added noise is carrier-to-noise used (C/Nused) which is defined
as follow :
•
We measure the C/Nrequired which the maximum C/N value for QEF reception when the
delay is included between 1.95µs and 0.95*guard interval and echo attenuation is 0dB.
•
C/Nused = C/Nrequired + 3 dB.
6.2
DiBcom results
6.2.1
8K FFT mode
Legend :
•
Blue : Guard intarval : 1/4 => 896*1/4 = 224µs
•
Pink : Guard intarval : 1/8 => 896*1/8 = 112µs
•
Cyan : Guard intarval : 1/16 => 896*1/16 = 56µs
•
Black : Guard intarval : 1/32 => 896*1/32 = 28µs
Modulation Mode : 8k, QPSK, CR 1/2
Figure 29: Curve without noise added in 8K, QPSK, CR 1/2
page 45 (45)
Figure 30: Curve within noise added, C/N = 9.5 dB, in 8K, QPSK, CR 1/2
Modulation Mode : 8k, 16-QAM, CR 2/3
Figure 31: Curve without noise added in 8K, 16QAM, CR 2/3
page 46 (46)
Figure 32: Curve within noise added, C/N = 19.7 dB, in 8K, 16QAM, CR 2/3
6.2.2
2K FFT mode
Legend :
•
Blue : Guard intarval : 1/4 => 224*1/4 = 56µs
•
Red : Guard intarval : 1/8 => 224*1/8 = 28µs
•
Green : Guard intarval : 1/16 => 224*1/16 = 14µs
•
Pink : Guard intarval : 1/32 => 224*1/32 = 7µs
Modulation Mode : 2k, QPSK, CR 1/2
page 47 (47)
Figure 33: Curve without noise added in 2K, QPSK, CR ½
Figure 34: Curve within noise added, C/N = 9.7 dB, in 2K, QPSK, CR 1/2
Modulation Mode : 2k, 16-QAM, CR 2/3
page 48 (48)
Figure 35: Curve without noise added in 2K, 16QAM, CR 2/3
Figure 36: Curve within noise added, C/N = 20.4 dB, in 2K, 16QAM, CR 2/3
6.2.3
page 49 (49)
Comparison between Guard Interval
Figure 37: Curve standardizing (delay / each IG) for 8k 16QAM CR 1/2 all IG
While standardizing delay by guard time, we can observe that the curves are not superimposed.
This below figure shows the authorized going beyond of time, in percentage by ratio to symbol
duration time.
With :
Equalisation time limit where the modulator to operate correctly :
ETimeLimit = 7 / 24 * TU
Out time interval equal :
Out time [% TU] = ( ETimeLimit − IG ) / TU
page 50 (50)
Figure 38: Curve standardizing for going beyond of guard time (8k 16-QAM CR 1/2 all IG)
With following curve, we notice that there are not effect between 8k and 2k FFT modes. We can
think that the 4k FFT mode do not should also pose effects. Effect for various bandwidths is also no
one.
page 51 (51)
Figure 39: Curve standardizing for going beyond of guard time (2k & 8k 16-QAM CR 1/2 all
IG)
6.2.4
Code rate influence for SFN mode
However, the difference between the C/Nrequired (in QEF reception) for 2 successive values of
code rate is larger and not constant when we measure in echo mode that in gaussian mode.
Finally, for large values of code rate, the modulator has a lot of difficulty to operate. With echoes
operating, the code rate does not have to exceed 3/4.
The below DiBcom results realized in 8MHz 8k/2k 16-QAM IG 1/4 various CR show that.
Table 23: DVB-H Performance according to pure echoes and modulations
6.3
Conclusion
The C/Nrequired (in QEF reception) for weak code rate is strong. Use of code rate 3/4 and 5/6 is
not adapted in pure SFN mode.
This receiver is not sensitive to the 2 paths channel if the delay between these 2 paths is less than
the guard interval.
There are not notion of pre-echo or post-echo in these results (the curve a symmetrical around 0).
Even with a delay higher than the guard interval the demodulation is possible. For instance in 8K
FFT mode 16-QAM CR 1/2 & GI 1/16 => GI = 56µs the QEF is reach for 100µs delay between the
2 paths with the same loss (0dB attenuation), and a delay of 150µs is possible with a loss of 3dB
between the 2 paths.
page 52 (52)
Figure 40: 8k mode 16-QAM 1/2 all IG
Figure 41: 8k mode 16-QAM 1/2 IG 1/16
We observe that the absolute value for which the maximum attenuation echo is reach is the same
(+/- 300µs). Then, the echo is regarded as a constant noise.
The break-point corresponding at 7/24 * TU can be noticed.
page 53 (53)
These curves show that the conclusion points is correct, like the curve symmetrical or the
possibility of leaving of the guard interval while keeping a correct operation.
The 4K mode has not been measured because of our test equipment which can not generate 4K
mode + SFN channel. However, in 4k mode FFT, the expected value is proportional with a ratio 2
between 2k-4k and 4k-8k FFT modes. Thus, we can easily to found the limit guard interval values,
like the following instance, with 16-QAM CR 1/2 & GI 1/16 :
•
•
•
In 8k FFT mode => the QEF is reach for 100µs delay between the 2 paths with the same
loss (0dB attenuation).
In 2k FFT mode => the QEF is reach for 25µs delay between the 2 paths with the same
loss (0dB attenuation), corresponding to a ratio 4 (2*2).
In 4k FFT mode => the expected value is 50µs.
page 54 (54)
7
Mier Comunicaciones laboratory tests
Mier Comunicaciones laboratory tests are focused on the analysis of on-channel repeaters (also
known as gapfillers) as source of DVB-H signal. The results obtained on these tests are used on
the work done at Task 5 of Work Package 2: The usability of repeaters in DVB-H networks at
reference document [1].
This SFN repeater, featuring as an easier deployment and lower cost equipment, has the problem
then of transmitting at the same frequency in which the signal is received, so a feed back from the
transmitted signal is induced at the input of the repeater.
B
Feedback Path
(i.e. Antennae Coupling)
RF
Received Signal
A
Delay
RF
Output Signal
Gap-Filler
Figure 42: On-channel Repeater diagram
This situation limits the operation gain (i.e. output power) depending on the installation conditions
(antennae coupling). The operation and limits of on-channel repeaters with DVB-H technology are
one part of the work done on these laboratory tests.
An improvement to the limitation of on-channel repeaters has been the use of echo canceller
devices. The use of this solution is proved and already working for DVB-T on-channel repeaters,
allowing the deployment of this repeater on DVB-T networks.
But repeaters for DVB-H networks will move from the typical broadcast sites on rural or mountain
areas with static environments to urban sites with dynamic environments. Tests simulating these
new situations have been performed to on-channel repeaters with standard echo canceller device.
Previously, and in order to simulate these different coupling profiles for an on-channel repeater on
laboratory, a RF multipath generator device have been designed.
So on the work done has been split and is presented with the following points:
1. WingTV laboratory trails at RAI: a conventional on-channel repeater is simulated and
tested with a DVB-H receiver.
2. Design of a multipath generator RF input / RF output: in order to simulate different
coupling conditions for on-channel repeaters on DVB-H networks.
3. On-channel repeater with standard echo canceller laboratory tests: This equipment is
tested for different conditions expected for repeaters on DVB-H networks.
7.1
WingTV laboratory trails at RAI
7.1.1
On-channel repeaters test at RAI
A specific test for on-channel repeaters was defined at laboratory trails at RAI, with the objective to
test the performance of a DVB-H receiver with this transmitting equipment.
The operation point for the on-channel repeater was set at the maximum safe point specified on the
DVB-H Implementation Guidelines [7]. So on it was simulated an on-channel repeater with an
operation gain 10 dB below the feedback coupling (the operation gain is defined as the difference
between desired emitted power and power received by the repeater).
The performance of the DVB-H receiver with this signal is compared from other different SFN
profiles signals defined (situations where more than one signal is received).
page 55 (55)
The configuration for the lab test bench in order to simulate the on-channel repeater system (i.e. a
feedback of the transmitted signal to the receiving) is detailed hereunder, based on the use of the
Elektrobit Dual Channel Simulator.
ELEKTROBIT
CHANNEL SIMULATOR
COFDM
Modulator
IF/RF
CONVERTER
CH 1
TU6 PROFILE
EFA Test receiver
DVB-H
RECEIVER
CH 2
10 dB ATTEN.
5 us DELAY
Figure 43: Lab test for on-channel repeater simulation
The Channel #1 of the Elektrobit channel simulator is used for the TU6 generation (i.e. the
degradations from the mobile channel model), and the channel #2 is used for the on-channel
repeater model (i.e. a delay and a relative attenuation for the fed back signal at the repeater).
The feed back impulse response of the on-channel repeater is shown on next figure, where
multiple replica signals are transmitted.
Figure 44: Simulated On-channel repeater response
A 5 us delay has been used as typical delay value (see DVB-H Implementation Guidelines [7]).
One screen shot of the global impulse response for this system is shown below. The receiver sees
multiple replicas of the TU6 profile with the convolution of the multiple signals transmitted for the
on-channel repeater.
Figure 45: Receiver input impulse response
The test has been performed with one modulation parameters.
•
FFT: 8K
•
Guard interval: ¼
•
Modulation: 16QAM
page 56 (56)
•
Code error rate: 1/2
•
MPE-FEC: ¾
•
Rows: 512
•
Burst length: 250 ms
Further work will be done with different modulation parameters.
7.1.2
Results
The results obtained are attached on the following table.
Table 24: Comparison of performance for different SFN profiles (1 to 3) and the on-channel
repeater (gapfiller)
SFN-Profile1
SFN-Profile2
SFN-Profile3
SFN-Profile4:
Gapfiller
Rel. Level
-15,5 dB
0 dB
0 dB
Delay
179,2
179,2
6 us
Fd Max
120 Hz
130 Hz
125 Hz
-10 dB
5 us
120 Hz
MFER 5%
Fd Max / 2
3 dB
13,0 dB
120 Hz
12,0 dB
130 Hz
12,0 dB
125 Hz
13,5 dB
120 Hz
10 Hz
13,0 dB
12,5 dB
12,0 dB
13,0 dB
The SFN profiles from 1 to 3 refer to the reception of two signals with the relative level and delay
specified. The SFN profile 4 is the profile defined by the special lab test bench for on-channel
repeater (gapfiller).
The receiver performance is checked in these profiles with the MFER 5% error criteria with the
following situations: C/N need at Fdmax, C/N need at Fdmax/2, Fd at 3dB, C/N need at 10 Hz
It can be concluded that from the table of results that:
•
Similar values for the different situations are obtained, so receiving from an on-channel
repeater is not different from receiving to other SFN conditions.
•
The safe operation margin recommended for a conventional on-channel repeater (this is
with non echo canceller device) at DVB-H Implementation Guidelines [7], where an
operation gain 10 dB below the isolation is specified, is checked as a value not to be
surpassed.
7.2
Design of a RF multipath generator
A RF COFDM Multipath Generator has been designed and mounted. This equipment is able to
emulate, in laboratory conditions, of any coupling channel between the transmitting and receiving
antennas that could be seen by an On-Channel Repeater (OCR) in a real site. In other words, it
emulates the echoes generated by the OCR when “passed” through the site-scenario. Therefore, it
is a very useful tool for evaluating the performance and limitations of the device under test, being it
a conventional or an OCR with an echo canceller device.
The following figure shows the block diagram of the COFDM Multipath Generator.
page 57 (57)
COFDM MULTIPATH GENERATOR
Down-converter
Up-converter
DSP
RF
RF
RF Input
RF Output
Figure 46: Block diagram of the COFDM MultiPath Generator
A very low phase-noise, high stability reference oscillator is used for the internal synthesis of the
Local Oscillator, which is used both for the down-conversion and up-conversion. Therefore, the
output RF frequency is exactly the same as the input frequency.
The COFDM Multipath Generator designed takes a DVB-T/DVB-H input signal and generates
multiple time-delayed replicas of such signal at exactly the same frequency.
Path #n
Path #(n-1)
Useful
signal
Paths
#1 to #n
Path #1
Rx
COFDM Path
Generator
Tx
On-Channel
Repeater
OCR in a real site: n coupling paths
between Tx and Rx antennas
Useful
signal
On-Channel
Repeater
(D.U.T.)
RF
Coupler
Dummy
Load
OCR in Laboratory: emulation of n
coupling paths between Output-Input
Figure 47: Using the COFDM MultiPath Generator to emulate real-site echoes
Each path can be configured individually to be static (constant amplitude) or dynamic (time-variable
amplitude). When configured as dynamic, the speed with which the amplitude changes can also be
programmed. There is also the possibility to emulate signal fading, that is, the speed with which all
the paths change their amplitude can be programmed.
The following figure shows two impulse response examples that can be generated (the impulse
response shown is the one of the MultiPath Generator)
(A)
(B)
Figure 48: An example of some of the impulse responses that can be generated
In the first profile (A), the MultiPath Generator acts as a single path. This could be used to emulate
only the direct coupling between the transmitting and receiving antennas in a OCR.
page 58 (58)
The second profile (B) is a combination of paths with different amplitudes and delays. It emulates
the antenna coupling channel that could take place in urban sites, where there are multiple echoes
caused by the buildings around the OCR.
The COFDM Multipath Generator can also be used to emulate the signal that would see any
receiver in a real SFN network. In SFN networks, the received signal is a combination of several
time-delayed components with different amplitudes each one (one component from each
transmitter):
Reference
signal
COFDM Path
Generator
D.U.T.
Figure 49: Using the COFDM MultiPath Generator to emulate SFN reception components
The following figure shows two examples of the kind of receiving signals that can be generated.
Figure 50: Two examples of a receiving signal, generated with the MultiPath Generator
7.3
DVB-H On-channel repeaters with echo cancellers
laboratory tests
As it has been seen on the laboratory with the on-channel repeaters tests at RAI, the safe
operation point (gain 10 dB below the isolation) should be respected on DVB-H technology. This
restriction affects the deployment of this equipment on field as the output power is limited
depending on the installation conditions.
In order to improve this restriction, echo cancellers have been used in DVB-T On-channel
repeaters, allowing the deployment of this equipment to extend coverage on DVB-T networks.
But the use of this solution on DVB-H networks has to deal different environment conditions from
the DVB-T ones, as repeaters will be mostly moved to urban areas.
So on the following profiles are proposed to simulate different environment conditions to test
standard echo canceller devices for on-channel repeaters performance:
•
Profile #1: Rural or mountain profile. A single echo that simulates the direct coupling
between transmitting and receiving antennas. This profile is corresponding to the typical
broadcast site on a mountain area.
•
Profile #2: Static urban profile. Multiple echoes with different amplitudes and delays.
This profile would be corresponding to urban areas, where objects (buildings…) around the
on-channel repeater reflect the transmitted signal, and therefore multiple coupled signals
are present at the input of the on-channel repeater.
•
Profile #3 Dynamic urban profile. Multiple dynamic echoes with varying amplitudes. It’s
the same as profile #2, but non static reflection objects (vehicles…) are added. Therefore
multiple coupled signals are present at the input of the on-channel repeater with varying
amplitudes.
The results for an on-channel repeater without (left part) and with (right part) standard echo
canceller device are attached for the different profiles defined.
7.3.1
page 59 (59)
Profile 1
This profile simulates a single echo.
1st
2nd
3rd
4th
Figure 51: Standard echo canceller performance with profile #1 (Impulse response and
spectrum, without (left) and with (right) echo canceller device)
It can be seen on the measurements the cancellation of the echo on the time domain (upper
measurements) and the elimination of the ripple on the spectrum (below). This performance allows
improving the restriction on the gain (i.e. output power) for on-channel repeaters.
7.3.2
Profile 2
In this profile multiple echoes are considered, and they are transmitted multiple times as they are
feedback through the on-channel repeater.
1st
2nd
3rd
page 60 (60)
Figure 52: Standard echo canceller performance with profile #2 (Impulse response and
spectrum, without (left) and with (right) echo canceller device)
In this situation it can be seen on time domain measurements (above) that standard echo canceller
only eliminates part of the echoes. Specifically, it cancels the first part that corresponds to the
direct path signal from the transmitting antenna, but it doesn’t cancel longer distance (time) signals.
On the spectrum domain (below measurements), with this profile peaks on the signal appear added
to the ripple, but doesn’t disappear also with the use of a standard echo canceller.
So on with the performance observed on this profile the use of a standard echo canceller
practically doesn’t improve the limitation of the on-channel repeater and the use of this equipment
will hardly depend on the environment conditions of each site. Enhanced Echo Cancellation
techniques should deal with this multiple echoes environment.
7.3.3
Profile 3
Respect to the previous profile the multiple echoes are now amplitude varying.
Figure 53: Standard echo canceller performance with profile #3 (Impulse response without
(left) and with (right) echo canceller device)
Practically there’s no difference with the operation of a standard echo canceller with this profile (as
even the first part of echoes is not cancelled). The processing time that the echo canceller takes to
calculate the cancelling values is too large, and the conditions have changed then. Enhanced Echo
Cancellation techniques should deal with this time variable multiple echoes environment.
7.4
Conclusions
From the work done on these laboratory tests with on-channel repeaters and standard echo
cancellers devices for on-channel repeaters it can be concluded the following:
•
DVB-H conventional on-channel repeater (this is with non echo canceller device) will be
output power limited depending on the installation conditions, as it happened on DVB-T
networks.
page 61 (61)
•
In general DVB-H repeaters will be installed under different environments than DVB-T
repeaters, as they will move closer to urban sites. This provokes that conditions for onchannel repeaters will be harder, and so on advanced profiles to simulate these conditions
should be considered on laboratory to test this transmitting equipment performance.
•
The improve with standard echo cancellers have been tested under different simulated
conditions for on-channel repeaters on a DVB-H network, showing limitations due the
multiple objects and variant conditions of urban areas.
•
Advanced echo cancellers should be considered to manage these harder conditions.
page 62 (62)
8
SIDSA SuperLode automatic test tool
8.1
Introduction
SIDSA has developed the SuperLode application to perform automatic lab measures.
The SuperLode is an automated measuring tool coded in Borland Builder C++ for the R&S SFQ
and DVB-T/H receivers. Its functionality encompasses both static and mobile channel situations.
-4
For static channels, SuperLode’s objective is to obtain the minimum C/N for a QEF BER of 2 x 10 .
Channels supported under this category are Gaussian, Rayleigh, and Ricean. On the other hand,
for mobile channels the criteria changes to MFER 5% and the objective is to find Fdmax, C/Nmin at
Fd3dB, and C/Nmin at 2Hz, 10 Hz, and at Fdmax/2, printing the final results and drawing the MFER 5%
curve. The figure below shows a running static test.
Figure 54: SuperLode user interface
8.2
Setup
The SuperLode Setup is based in the SFQ. The interface with it is done through Serial port.
Commands are sent to change the channel characteristics, as well as the modulation parameters.
The SuperLode program also monitors the SIDSA receiver console output to check the reception
quality.
The setup is as follows:
page 63 (63)
Figure 55: Test synoptic
8.3
Static Channel Tests
SuperLode is very versatile and open to several configuration options for static channel tests. The
bisection algorithm for finding the minimum C/N is very fast and straight forward, although several
configuration parameters have to be set. For example, capture time lets you define the time in
seconds you want SuperLode to read C/N statistics before deciding to go to the next step. Normal
capture times are between 5 seconds and 10 seconds. The Delta variable giver the tolerance of the
search algorithm. Within that tolerance, the result is taken as good enough and printed.
The main test variables are set on an .ini file and has the following structure:
[Main]
Title="Loren 3.0 (Internal ADC)"
Std=DVB-H
[Test1]
Title="8K 16QAM 2/3 1/4"
Type=Static
Freq=666000000
Level=-30
Mod=16QAM
FFT=8k
Guard=-100
HP_Rate=2_3
LP_Rate=2_3
BW=8
Channel=awgn
Alpha=0
IQ=normal
The first section called [Main] gives general information about the tests to be performed. This is just
for reporting purposes. The next section called [Test1] shows the test variables to be used for this
page 64 (64)
test: frequency, power level, modulation, FFT, guard interval, high and low priority code rates,
bandwidth, channel type, hierarchy and IQ swap. If a sweep needs to be done on any of these
variables, a -100 has to be placed as shown for the guard interval case.
8.4
Dynamic Channel Tests
Dynamic tests also have several configuration options to take care of. For example, the user has
the choice of entering the PID service to be used for statistics, the maximum doppler starting point,
the number of frames to get statistics from, and the MFER delta which sets the tolerance of the
error to be taken.
The procedure is similar as with the static channel tests, and only the ‘Type’ variable in the .ini file
has to be changed to ‘Dynamic’.
The algorithm starts with a bisection to find the maximum doppler for C/N of 50 dB (maximum SFQ
C/N), then it moves down to get the minimum C/N for 2 Hz, 10 Hz, and Fdmax/2, and finally it looks
for the doppler at Fd3dB.
8.5
Output Report
SuperLode is able to report in three different formats:
•
Straight text
•
RTF
•
HTML
The HTML output is shown below for a dynamic channel test. It shows the main test parameters,
then gives a small table of results, and finally shows the MFER 5% figure. For static channels, the
output is set to table format.
page 65 (65)
Figure 56: Report page sample
8.6
WingTV measure results:
Static:
The static results are:
CN for post Viterbi BER=2e-4 (dB)
BW
GI
Mode
Modulation
QPSK
8MHz
1/4
1/8
1/16
1/32
2k
4k
8k
16QAM
64QAM
CR
Gaussian channel
FPGA SNR 25 dB
Theory
MBRAI
Loss
1/2
3,9
3,5
5,6
0,4
2/3
5,7
5,3
7,4
0,4
3/4
6,7
6,3
8,4
0,4
1/2
9,7
9,3
11,3
0,4
2/3
12
11,4
13,7
0,6
3/4
13,3
12,6
15,1
0,7
1/2
14,7
13,8
17,0
0,9
2/3
18,1
16,7
19,2
1,4
20,2
19,4
20,8
0,8
3/4
page 66 (66)
CN for post Viterbi BER=2e-4 (dB)
BW
GI
Mode
Modulation
QPSK
8MHz
1/4
1/8
1/16
1/32
2k
4k
8k
16QAM
64QAM
Gauusian channel
CR
FPGA SNR 25
dB
Theory P
MBRAI
Theory
Loss with
Loss with P
P6
P6
1/2
7,2
5,9
7,9
5,9
1,3
1,3
2/3
10,4
9,6
10,9
9,9
0,8
0,5
3/4
14,3
12,4
13,2
13,6
1,9
0,7
1/2
13,4
11,8
13,8
11,8
1,6
1,6
2/3
16,8
15,3
16,8
15,8
1,5
1,0
3/4
19,8
18,1
19,4
19,4
1,7
0,4
1/2
18
16,4
18,7
16,7
1,6
1,3
2/3
23
20,3
22,1
20,9
2,7
2,1
3/4
27
23
24,8
24,4
4,0
2,6
Doppler:
FFT
GI
Modulation Rate
Burst
MPE
Rows Length
-FEC
(ms)
"Top
modes"
MFER 5%
AWGN
Rayleigh
Fd Max
Fd Max / 2
3 dB
10 Hz
6,7 dB
107 Hz
9,0 dB
105 Hz
9,0 dB
8K
1/4
QPSK
1/2
1/2
512
250
3,2 dB
8K
1/4
QPSK
1/2
2/3
512
250
3,2 dB
6,8 dB
107 Hz
9,0 dB
100 Hz
9,0 dB
8K
1/4
QPSK
1/2
3/4
512
250
3,3 dB
6,9 dB
99 Hz
9,0 dB
100 Hz
9,0 dB
8K
1/4
QPSK
1/2
5/6
512
250
3,3 dB
6,9 dB
99 Hz
10,0 dB
95 Hz
10,0 dB
8K
1/4
QPSK
1/2
7/8
512
250
3,4 dB
7,0 dB
98 Hz
11,0 dB
93 Hz
11,0 dB
8K
1/4
QPSK
2/3
2/3
512
250
5,2 dB
9,9 dB
97 Hz
11,5 dB
93 Hz
11,5 dB
8K
1/4
QPSK
2/3
3/4
512
250
5,2 dB
9,9 dB
97 Hz
12,0 dB
93 Hz
12,0 dB
8K
1/4
QPSK
2/3
5/6
512
250
5,2 dB
10,0 dB
97 Hz
13,0 dB
90 Hz
13,0 dB
8K
1/4
QPSK
2/3
7/8
512
250
5,3 dB
10,1 dB
96 Hz
13,5 dB
90 Hz
13,5 dB
8K
1/4
16QAM
1/2
2/3
512
250
9,1 dB
11,3 dB
96 Hz
15,0 dB
90 Hz
15,0 dB
8K
1/4
16QAM
1/2
3/4
512
250
9,1 dB
11,4 dB
96 Hz
15,5 dB
90 Hz
15,5 dB
8K
1/4
16QAM
1/2
5/6
512
250
9,2 dB
11,4 dB
94 Hz
16,0 dB
87 Hz
16,0 dB
8K
1/4
16QAM
1/2
7/8
512
250
9,2 dB
11,5 dB
92 Hz
17,0 dB
85 Hz
17,0 dB
8K
1/4
16QAM
2/3
2/3
512
250
11,3 dB
15,2 dB
87 Hz
18,0 dB
83 Hz
18,0 dB
8K
1/4
16QAM
2/3
3/4
512
250
11,3 dB
15,2 dB
86 Hz
18,5 dB
80 Hz
18,5 dB
8K
1/4
16QAM
2/3
5/6
512
250
11,4 dB
15,3 dB
82 Hz
19,5 dB
78 Hz
19,5 dB
8K
1/4
16QAM
2/3
7/8
512
250
11,5 dB
15,3 dB
75 Hz
21,0 dB
75 Hz
21,0 dB
8K
1/4
64QAM
1/2
5/6
512
250
14,1 dB
17,5 dB
70 Hz
22,0 dB
70 Hz
22,0 dB
8K
1/4
64QAM
1/2
7/8
512
250
14,1 dB
17,6 dB
70 Hz
23,0 dB
65 Hz
23,0 dB
8K
1/4
64QAM
2/3
2/3
512
250
17,4 dB
22,5 dB
60 Hz
25,0 dB
50 Hz
25,0 dB
8K
1/4
64QAM
2/3
3/4
512
250
17,5 dB
22,5 dB
50 Hz
26,0 dB
50 Hz
26,0 dB
8K
1/4
64QAM
2/3
5/6
512
250
17,5 dB
22,6 dB
45 Hz
29,0 dB
40 Hz
29,0 dB
9
page 67 (67)
Conclusions
In the Wing-TV project, the work package 4 has been in charge to coordinate all laboratories tests,
common and individual. Several partners of this project have performed many laboratories tests
which have confirmed DVB-H receiver performances during the three common laboratories tests.
But, these tests have allowed to investigate others parameters of DVB-H receiver in mobile
environment.
The same measurement methodology has been used in all laboratories tests (common and
individual) based on the document “Validation task force report” and Wing TV D8 report. Thank to
the same methodologies, all results should be compared and analysed.
The RAI and NOKIA tests are the continuity of the common laboratories tests in order to complete
and to confirm the DVB-H receiver performances (C/N and Frequency Doppler) with the TU6 profile
in the different DVB-H modes. An analysis has been realized to determine the influence of each
DVB-H modulation parameters and, then, to help the user to choose the best DVB-H mode
according to this application.
The Ericsson tests have permit to evaluate a new ICI canceller algorithm at low complexity and
achieve good performance by using a window function and cancellation of ICI on one sub-carrier
from two adjacent sub-carriers. The functionality of the ICI cancelling algorithm has been proved
using both frequency offset and a time-variant channel causing Doppler spread, with good results,
very close to the theoretical improvement.
The TeamCast tests are a first approach of the bandwidth shrinkage effect in a context no DVB-H
standardized but compatible of the T-DMB (or DAB) standard. The performances are consistent
between 1.75MHz and 7MHz: Maximum Frequency Doppler four times lower, receiver
consumption four times less important. The remaining work consists to evaluate DAB modulation
with the same methodology.
The DiBcom tests have consisted to validate their DVB-H receiver in environment combining SFN
and mobility. This receiver is not sensitive to the 2 paths channel if the delay between these 2
paths is less than the guard interval. There are not notion of pre-echo or post-echo in these results
(the curve a symmetrical around 0). Even with a delay higher than the guard interval the
demodulation is possible.
The Mier tests have permit to evaluate the effect of a mobile channel profile on a gap-filler including
a standard echo canceller. This research shows the mobile transmissions are difficult to the echo
canceller. But, the improve with standard echo cancellers have been tested under different
simulated conditions for on-channel repeaters on a DVB-H network, showing limitations due the
multiple objects and variant conditions of urban areas. Advanced echo cancellers should be
considered to manage these harder conditions.
page 68 (68)
References
[1] Wing TV: “Wing TV Network Issues”. D11. June 2006.
[2] CEPT-EBU: "Report on planning and Introduction of Terrestrial Digital Television in
Europe", December 1997.
[3] G. Faria, J. A. Henriksson, E. Stare, and P. Talmola, ”DVB-H: Digital Broadcast Services to
Handheld Devices,” Proc. of the IEEE, Vol. 94, No. 1, pp. 194-209, Jan 2006.
[4] M. Faulkner, L. Wilhelmsson, and J. Svensson, “Low-complex ICI cancellation for
improving Doppler performance in OFDM systems,” Accepted for presentation at IEEE
Vehicular Technology Conference 2006 Fall, 25-28 Sept. 2006, Montreal Canada.
[5] ETSI EN 300 744: Framing structure, cannel coding and modulation for digital terrestrial
television.
[6] ETSI TR 102 401 v1.1.1: Validation Task Force report.
[7] ETSI TR 102 377 v1.2.1: DVB-H Implementation Guidelines

Wing TV-WP4-D5 Compa..

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