Ampli Norton - Philippe Roux

AMPLIFICATEUR OPERATIONNEL DE TYPE " NORTON "
( LM3900 de National Semiconductor ) 1
Le circuit intégré LM3900 dont la documentation est donnée en annexe comporte quatre
amplificateurs différentiels de transconductance dont la tension de sortie est proportionnelle à la
différence des courants appliqués aux deux entrées. On se propose d’analyser le fonctionnement
d’un amplificateur de ce type.
1°PARTIE : ÉTUDE DE L'AMPLIFICATEUR DIFFÉRENTIEL DE COURANT
La tension d’alimentation VCC est fixée à 30 V et la température à 25°C où |VBE | = 0,6 V
pour tous les transistors.
On considère l'étage amplificateur représenté figure 1 où le transistor T1 (gain en courant
!n de 100) est chargé par un générateur de courant idéal IC1 = 150 µA et excité par un générateur
de courant : IB1 = IC1 / !n pour que le transistor soit polarisé dans sa zone linéaire.
+V CC
1) Pour une variation faible du courant de base ib1 autour de sa valeur
de repos IB1, déterminer en tenant compte de la résistance interne rce1
(500 K! ) du transistor T1 :
IC1
T1
a) Le gain en tension A1 = v2 / v1 du montage.
IB1
v1
v2
b) Les résistances d'entrée Re1et de sortie Rs1. Faire l’A.N.
Figure 1
On réalise un amplificateur différentiel de courant en associant au transistor amplificateur T1,
deux transistors rigoureusement identiques T2 et T3 (!n = 100) formant un miroir de courant simple
selon la figure 2.
+V CC
IC1
I B1 =
B1
IC 1
"n
T1
IC3
I1
B2
!
T3
IC2
I2
v1
v2
T2 VBE3
VBE2
Figure 2 : amplificateur différentiel de courant.
1
Philippe ROUX © 2009
http://rouxphi3.perso.cegetel.net
1
2) Calculer l’expression du courant IC1 de T1 en fonction des courants continus d’entrées I1 et I2
et du gain en courant !n des transistors.
On applique respectivement sur les entrées B1 et B2 des petites variations de courant i1 et i2
autour des valeurs de repos I1 et I2.
3) A l’aide du schéma équivalent aux petites variations et aux fréquences moyennes (négliger
la résistance rce de T2 et T3 devant rbe), montrer que la tension de sortie v2 est sensiblement
proportionnelle à la différence (i2-i1). Déterminer le coefficient de transconductance Rm de
cet amplificateur différentiel de courant.
2° PARTIE : RÉALISATION ET ÉTUDE DE L'AMPLIFICATEUR OPERATIONNEL
Le montage précédent possède un grand gain en tension mais sa résistance de sortie est
trop grande. Afin de diminuer celle-ci, on associe au transistor T1, le montage donné en figure 3.
Les transistors T4 et T5 sont polarisés par des générateurs de courant continus idéaux I4 et I5 (on a
toujours IC1 = 150 µA).
Le transistor PNP possède un gain en courant faible ! p = 5 (du aux procédures d’intégration).
+V CC
I4
T5
IC1
T4
B1
T1
IC3
I1
B2
T3
I2
v1
v2
I5
vs
T2
Figure 3
1) Etablir la relation liant les courants I4 et I5 au courant IC1 et au gain en courant !p de T4 et !n de
T5; On donne I5 = 1,2 mA, calculer I4. En déduire la valeur des résistances d'entrées rbe4 et rbe5 de
T4 et T5.
2) Dessiner le schéma aux petites variations équivalent au montage T4 et T5 de la figure 3,
chaque transistor étant remplacé par son schéma en « ! ib » (on négligera seulement la résistance
rce de T4 et on prendra pour T5 : rce = 50 K").
3) Calculer la résistance d’entrée Re4 du montage constitué par T4 et T5, vue par le transistor T1
entre C1 et la masse. Donner son expression approchée et faire l’application numérique. Le gain
en tension A1 est-il modifié ?
4) En déduire l’expression du gain en tension A2 = vs / v2 puis calculer A3 = vs / v1.
2
5) En utilisant la méthode de l’ohmmètre, déterminer la résistance de sortie Rs du montage complet
entre la sortie et la masse. Donner son expression approchée et faire l’A.N.
6) Etablir la relation liant la tension de sortie vs aux courant d'entrées i1 et i2.
3° PARTIE : UTILISATION
On réalise un amplificateur de tension en utilisant cet amplificateur opérationnel
« Norton » dans le montage de la figure 4. Les courants I1 et I2 définis sur la figure 2, sont fournis
respectivement par les résistances R1 et R2.
1) Dessiner le schéma en continu du montage de la figure 5 en mettant en évidence les courants I1 et
I2 .
2) En utilisant les résultats de la 1° partie, montrer que la tension continue de sortie VS est
proche de VCC / 2.
3) Dessiner le schéma du montage de la figure 4 pour les petites variations. On supposera que C1 et
C2 ont une impédance négligeable à la fréquence d’utilisation.
4) En utilisant les propriétés de l’amplificateur « Norton », calculer en régime des petits signaux, le
gain en tension A = vs / ve du montage de la figure 5 (les résistances d’entrées en B1 et B2 sont
supposées suffisamment faibles devant les autres résistances).
C2
+V CC = 30 V
R
R
47 K!
47 K!
R2
C1
100 K!
1 M!
B2
Norton
R3
B1
ve
R1
1 M!
Figure 4
3
vs
CORRECTION 2
1°PARTIE : ÉTUDE DE L'AMPLIFICATEUR DIFFÉRENTIEL DE COURANT
1. Schéma aux petites variations :
v1 vbe1rbe1
v2
ib1
rce1
gm1vbe1
a) Le gain en tension du montage : A1 =
Transconductance de T1 : gm1 =
b) Résistance d’entrée Re1!= "n
v2
= "gm1rce1 = "3000 (1)
v1
I C1
= 6mS .
UT
UT
= 16, 7K#
I C1
Rs1 = rce1 = 500K" (2)
!
2. Expression du courant IC1 de T1 en fonction des courants continus d’entrées I1 et I2 et du gain
!
en courant !n des transistors.
!
+V CC
IC1
IC1
B1 " n
IC3
I1
!
B2
T3
IC2
I2
T1
v1
v2
T2 V
BE3
VBE2
!
!
Les transistors T2 et T3 ont la même tension VBE.
V
V
I C 2 = I SBC exp( BE 2 )
I C 3 = I SBC exp( BE 3 )
soit : IC2 = IC3 et IB2 = IB3
UT
UT
I
1
I 2 = I C 2 + I B2 + I B3 = I C 2 + 2 C 2
IC2 = IC 3 = I2
2
"n
1+
"n
!
"n
I C1
I C1 = "n (I1 # I 2 ) (3)
I B1 =
= I1 # I 2
$ I1 # I 2
"n
2 + "n
!
2
! Philippe ROUX © 2009
!
http://rouxphi3.perso.cegetel.net
4
3. Schéma équivalent du montage.
B2 B3 C2
vbe2
vbe3
i2
B1 C3
rbe
2
C1
rbe1
vbe1
i1
gm2vbe2
v2
rce1
gm1vbe1
gm3vbe3
!
Nœud B2 : i2 " gm2 v be2 "
2vbe2
=0
rbe
soit : vbe2 = vbe3 =
i2
2
gm2 +
rbe
(4)
D’autre part : vbe1 = rbe1 (i1 " gm 3vbe3 ) (5)
!
v2 = "gm1vbe1rce1 (6)
!
!
#
&
%
(
i2
!
En exploitant
les relations (4) et 5) : v2 = "gm1rbe1rce1%(i1 " gm 3
)(
2 (
%
gm2 +
%$
rbe ('
Les transconductances gm2 et gm3 sont égales (IC2 = IC3) et d’autre part : gm2 >>
2
.
rbe
!
v2 " #gm1rbe1rce1 [i1 # i2 ] = Rm [i2 # i1 ] (7)
Transrésistance de l’amplificateur : Rm = "n rce1 = 50M# (8)
!
!
2° PARTIE : RÉALISATION ET ÉTUDE DE L'AMPLIFICATEUR OPERATIONNEL
!
1) Relation liant les courants I4 et I5 au courant IC1 et au gain en courant !p de T4 et !n de T5.
+ V CC
I4
"n
I 4 # (" p +1)I C1
!
IC1
T4
T5
(" n + 1)[ I4 # ( " p + 1)IC1 ]
!
! "p
" p I C1
!
!
!
1,2 mA
I5
! Courants dans les transistors T et T .
4
5
Expression du courant I5 : I 5 = " p I C1!+ ("n +1)[ I 4 # ("n +1)I C1 ]
5
!
I4 =
I 5 + I C1 ["n (" p +1) +1]
("n +1)
(9)
Application numérique :
!
I4
rbe4
IC5
rbe5
904 µA 167 ! 450 µA 5,55 k!
2) Schéma aux petites variations équivalent au montage T4, T5.
! pib4
ib4
rbe4
ib4
ib5
v2
rce
rbe5
vs
! nib5
3) Expression de la tension v2.
v2 = rbe4 ib4 + rbe5ib5 + rce [ib 4 + "nib5 ] (10)
Avec : ib5 = (" p +1)ib4
v
Résistance d’entrée : Re4 = 2 = rbe4 + (" p +1)rbe5 + rce [1+ "n (" p +1)]
ib 4
!
!
Re4 " rce [#n (# p +1)] = 30M$ (11)
!
Le gain A1 qui devient : A1 = "gm1 (rce1 // Re4 ) n’est pas modifié en effet : Re4 >> rce1.
4) Gain en tension A2. !
!
vs = rceib4 [1+ "n (" p +1)]
!
On en déduit alors : A2 =
v2 = Re4 ib 4
vs
= 1 (12)
v2
Aussi le gain en tension global est inchangé : A3= A1.
!
!
5) Résistance de sortie Rs du montage.
La méthode de l’ohmmètre impose d’annuler l’excitation des entrées soit i1 et i2 nuls. Dans
ces conditions le transistor T1 est seulement simulé par sa résistance interne rce1. Le schéma
d’analyse est le suivant :
! pib4
ib4
rbe4
rce1
ib4
i’
i
ib5
rce
rbe5
! nib5
6
u
u
u
= rce //
i"
i
i" = #ib 4 [1+ $n ($ p +1)]
u = "rbe5ib5 " (rce1 + rbe4 )ib 4
Rs =
!
!
!
Rs = rce //
rce1 + rbe4 + rbe5 (" p +1)
[1+ " ("
n
p
+1)]
#
rce1
= 833$ (13)
"n (" p +1)
6) Relation liant la tension de sortie vs aux courant d'entrées i1 et i2 : vs = Rm (i2 " i1 ) (14)
!
3° PARTIE : UTILISATION
!
1. Schéma en continu du montage.
+VCC = 30 V
R
47 K!
R
47 K!
R2
1 M!
I2
B2
Norton
B1
Vs
I1
R1
1 M!
2. Tension de sortie Vs en régime continu. Transformation du montage constitué par VCC, R et
R, à l’aide de Thévenin :
R/2
R2
1 M!
I2
B2
+VCC /2
Norton
B1
VBE2 VBE1
Vs
I1
R1
1 M!
On peut négliger la résistance de Thévenin R/2 devant R2.
7
!
VCC
"VBE 2
I2 = 2
(15)
Vs = R1 I1 +VBE1 (16)
R2
Compte tenu des propriétés de l’entrée B1 : I1 = I 2 + I B1 où IB1 représente le courant de base
de T1 (IB1 = 1,5 µA).
:
Sachant que R1 =R2 !
V
V
Vs = CC + R1 I!B1 + (VBE1 "VBE 2 ) # CC + R1 I B1 = 16, 5V
2
2
Vs est sensiblement égale à VCC/2.
3. Schéma du!montage pour les petites variations.
100 K!
i2
B2
Norton
R3
B1
R2
1 M!
ve
vs
i1
R1
1 M!
ve
sachant que R2 >> résistance d’entrée en B2.
R3
v
Courant i1 : i1 = s
R1
! la relation (14) : vs = Rm ( ve " vs )
Avec
R3 R1
v
Rm R1
A!
= s=
ve R3 (R1 + Rm )
v
R
A = s " 1 = 10
Sachant que !
: Rm >>R1
ve R3
4. Courant i2 : i2 =
!
!
8
LM2900/LM3900/LM3301 Quad Amplifiers
General Description
Features
The LM2900 series consists of four independent, dual input,
internally compensated amplifiers which were designed
specifically to operate off of a single power supply voltage
and to provide a large output voltage swing. These amplifiers make use of a current mirror to achieve the non-inverting input function. Application areas include: ac amplifiers,
RC active filters, low frequency triangle, squarewave and
pulse waveform generation circuits, tachometers and low
speed, high voltage digital logic gates.
Y
Y
Y
Y
Y
Y
Y
Y
Wide single supply voltage
4 VDC to 32 VDC
g 2 VDC to g 16 VDC
Range or dual supplies
Supply current drain independent of supply voltage
Low input biasing current
30 nA
High open-loop gain
70 dB
Wide bandwidth
2.5 MHz (unity gain)
a
Large output voltage swing
(V b 1) Vp-p
Internally frequency compensated for unity gain
Output short-circuit protection
Schematic and Connection Diagrams
Dual-In-Line and S.O.
TL/H/7936 – 2
Top View
TL/H/7936 – 1
C1995 National Semiconductor Corporation
TL/H/7936
Order Number LM2900N, LM3900M, LM3900N or LM3301N
See NS Package Number M14A or N14A
RRD-B30M115/Printed in U. S. A.
LM2900/LM3900/LM3301 Quad Amplifiers
February 1995
Absolute Maximum Ratings
If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
LM2900/LM3900
LM3301
Supply Voltage
32 VDC
28 VDC
g 16 VDC
g 14 VDC
Power Dissipation (TA e 25§ C) (Note 1)
Molded DIP
1080 mW
1080 mW
S.O. Package
765 mW
a
b
Input Currents, IIN or IIN
20 mADC
20 mADC
Output Short-Circuit DurationÐOne Amplifier
Continuous
Continuous
TA e 25§ C (See Application Hints)
b 40§ C to a 85§ C
Operating Temperature Range
b 40§ C to a 85§ C
LM2900
LM3900
0§ C to a 70§ C
b 65§ C to a 150§ C
b 65§ C to a 150§ C
Storage Temperature Range
260§ C
Lead Temperature (Soldering, 10 sec.)
260§ C
Soldering Information
Dual-In-Line Package
Soldering (10 sec.)
260§ C
260§ C
Small Outline Package
Vapor Phase (60 sec.)
215§ C
215§ C
Infrared (15 sec.)
220§ C
220§ C
See AN-450 ‘‘Surface Mounting Methods and Their Effect on Product Reliability’’ for other methods of soldering surface mount
devices.
ESD tolerance (Note 7)
2000V
2000V
Electrical Characteristics TA e 25§ C, V a e 15 VDC, unless otherwise stated
LM2900
Parameter
Open
Loop
Voltage Gain
Voltage Gain
LM3900
LM3301
Conditions
Units
Over Temp.
DVO e 10 VDC
Inverting Input
Min
Typ Max Min
Typ Max Min
Typ Max
1.2
2.8
2.8
2.8
1.2
1.2
V/mV
Input Resistance
1
1
1
Output Resistance
8
8
9
kX
2.5
2.5
2.5
MHz
Unity Gain Bandwidth
Inverting Input
Input Bias Current
Inverting Input, V
Inverting Input
Slew Rate
Positive Output Swing
Negative Output Swing
0.5
20
Supply Current
RL e % On All Amplifiers
6.2
Output
Voltage
Swing
RL e 2k,
a
V e 15.0 VDC
VOUT High
a
Output
Source
Current
Sink
Capability
ISINK
30
b
IIN e 0,
IIN a e 0
a
V e Absolute
Maximum Ratings
200
30
200
10
6.2
0.2
300
0.5
20
10
13.5
0.09
30
0.5
20
13.5
b
IIN e 10 mA,
a
IIN e 0
VOUT Low
VOUT High
e 5 VDC
MX
nA
V/ms
6.2
10
0.09
0.2
mADC
13.5
0.09
0.2
VDC
b
IIN e 0,
a
IIN e 0
RL e % ,
29.5
(Note 2)
b
VOL e 1V, IIN e 5 mA
29.5
6
18
6
10
5
18
0.5
1.3
0.5
1.3
0.5
1.3
5
2
26.0
5
5
mADC
Electrical Characteristics (Note 6), V a e 15 VDC, unless otherwise stated (Continued)
LM2900
Parameter
LM3900
LM3301
Conditions
Power Supply Rejection
TA e 25§ C, f e 100 Hz
Mirror Gain
@
@
20 mA (Note 3)
200 mA (Note 3)
20 mA to 200 mA (Note 3)
Units
Min
Typ
0.90
0.90
1.0
1.0
Max
Min
Typ
1.1
1.1
0.90
0.90
1.0
1.0
70
Max
Min
1.1
1.1
0.90
0.90
70
Typ
Max
70
1
1
dB
1.10
1.10
mA/mA
DMirror Gain
@
2
5
2
5
2
5
%
Mirror Current
(Note 4)
10
500
10
500
10
500
mADC
Negative Input Current
TA e 25§ C (Note 5)
1.0
1.0
Input Bias Current
Inverting Input
300
300
1.0
mADC
nA
Note 1: For operating at high temperatures, the device must be derated based on a 125§ C maximum junction temperature and a thermal resistance of 92§ C/W
which applies for the device soldered in a printed circuit board, operating in a still air ambient. Thermal resistance for the S.O. package is 131§ C/W.
Note 2: The output current sink capability can be increased for large signal conditions by overdriving the inverting input. This is shown in the section on Typical
Characteristics.
Note 3: This spec indicates the current gain of the current mirror which is used as the non-inverting input.
Note 4: Input VBE match between the non-inverting and the inverting inputs occurs for a mirror current (non-inverting input current) of approximately 10 mA. This is
therefore a typical design center for many of the application circuits.
Note 5: Clamp transistors are included on the IC to prevent the input voltages from swinging below ground more than approximately b 0.3 VDC. The negative input
currents which may result from large signal overdrive with capacitance input coupling need to be externally limited to values of approximately 1 mA. Negative input
currents in excess of 4 mA will cause the output voltage to drop to a low voltage. This maximum current applies to any one of the input terminals. If more than one
of the input terminals are simultaneously driven negative smaller maximum currents are allowed. Common-mode current biasing can be used to prevent negative
input voltages; see for example, the ‘‘Differentiator Circuit’’ in the applications section.
Note 6: These specs apply for b 40§ C s TA s a 85§ C, unless otherwise stated.
Note 7: Human body model, 1.5 kX in series with 100 pF.
Application Hints
Unintentional signal coupling from the output to the non-inverting input can cause oscillations. This is likely only in
breadboard hook-ups with long component leads and can
be prevented by a more careful lead dress or by locating the
non-inverting input biasing resistor close to the IC. A quick
check of this condition is to bypass the non-inverting input
to ground with a capacitor. High impedance biasing resistors used in the non-inverting input circuit make this input
lead highly susceptible to unintentional AC signal pickup.
Operation of this amplifier can be best understood by noticing that input currents are differenced at the inverting-input
terminal and this difference current then flows through the
external feedback resistor to produce the output voltage.
Common-mode current biasing is generally useful to allow
operating with signal levels near ground or even negative as
this maintains the inputs biased at a VBE. Internal clamp
transistors (see note 5) catch-negative input voltages at approximately b0.3 VDC but the magnitude of current flow has
to be limited by the external input network. For operation at
high temperature, this limit should be approximately 100 mA.
This new ‘‘Norton’’ current-differencing amplifier can be
used in most of the applications of a standard IC op amp.
Performance as a DC amplifier using only a single supply is
not as precise as a standard IC op amp operating with split
supplies but is adequate in many less critical applications.
New functions are made possible with this amplifier which
are useful in single power supply systems. For example,
biasing can be designed separately from the AC gain as was
shown in the ‘‘inverting amplifier,’’ the ‘‘difference integrator’’ allows controlling the charging and the discharging of
the integrating capacitor with positive voltages, and the ‘‘frequency doubling tachometer’’ provides a simple circuit
which reduces the ripple voltage on a tachometer output DC
voltage.
When driving either input from a low-impedance source, a
limiting resistor should be placed in series with the input
lead to limit the peak input current. Currents as large as
20 mA will not damage the device, but the current mirror on
the non-inverting input will saturate and cause a loss of mirror gain at mA current levelsÐespecially at high operating
temperatures.
Precautions should be taken to insure that the power supply
for the integrated circuit never becomes reversed in polarity
or that the unit is not inadvertently installed backwards in a
test socket as an unlimited current surge through the resulting forward diode within the IC could cause fusing of the
internal conductors and result in a destroyed unit.
Output short circuits either to ground or to the positive power supply should be of short time duration. Units can be
destroyed, not as a result of the short circuit current causing
metal fusing, but rather due to the large increase in IC chip
dissipation which will cause eventual failure due to excessive junction temperatures. For example, when operating
from a well-regulated a 5 VDC power supply at TA e 25§ C
with a 100 kX shunt-feedback resistor (from the output to
the inverting input) a short directly to the power supply will
not cause catastrophic failure but the current magnitude will
be approximately 50 mA and the junction temperature will
be above TJ max. Larger feedback resistors will reduce the
current, 11 MX provides approximately 30 mA, an open circuit provides 1.3 mA, and a direct connection from the output to the non-inverting input will result in catastrophic faila
ure when the output is shorted to V as this then places the
base-emitter junction of the input transistor directly across
the power supply. Short-circuits to ground will have magnitudes of approximately 30 mA and will not cause catastrophic failure at TA e 25§ C.
3
Typical Performance Characteristics
Open Loop Gain
Voltage Gain
Voltage Gain
Input Current
Supply Current
Large Signal Frequency
Response
Output Sink Current
Output Class-A Bias Current
Output Source Current
Supply Rejection
Mirror Gain
Maximum Mirror Current
TL/H/7936 – 9
4
Typical Applications (V a e 15 VDC)
Inverting Amplifier
Triangle/Square Generator
a
VODC e
V
2
AV j b
R2
R1
TL/H/7936– 3
TL/H/7936 – 4
Frequency-Doubling Tachometer
Low VIN b VOUT Voltage Regulator
TL/H/7936 – 5
TL/H/7936 – 6
Non-Inverting Amplifier
Negative Supply Biasing
a
VODC e
AV j
V
2
VODC e
R2
R1
AV j
TL/H/7936 – 7
5
R2 b
V
R3
R2
R1
TL/H/7936 – 8
Typical Applications (V a e 15 VDC) (Continued)
Low-Drift Ramp and Hold Circuit
TL/H/7936 – 10
Bi-Quad Active Filter
(2nd Degree State-Variable Network)
Q e 50
fO e 1 kHz
TL/H/7936 – 11
6
Typical Applications (V a e 15 VDC) (Continued)
Voltage-Controlled Current Source
(Transconductance Amplifier)
TL/H/7936 – 12
Hi VIN , Lo (VIN b VO) Self-Regulator
Q1 & Q2 absorb Hi VIN
TL/H/7936 – 13
Ground-Referencing a Differential Input Signal
TL/H/7936 – 14
7
Typical Applications (V a e 15 VDC) (Continued)
Voltage Regulator
Fixed Current Sources
(VO e VZ a VBE)
TL/H/7936–15
I2 e
Voltage-Controlled Current Sink
(Transconductance Amplifier)
R1
I1
R2
TL/H/7936 – 16
Buffer Amplifier
VIN t VBE
TL/H/7936 – 18
TL/H/7936 – 17
Tachometer
VODC e A fIN
TL/H/7936 – 19
8
*Allows VO to go to zero.
Typical Applications (V a e 15 VDC) (Continued)
Low-Voltage Comparator
Power Comparator
No negative voltage limit if
properly biased.
TL/H/7936 – 21
TL/H/7936 – 20
Comparator
Schmitt-Trigger
TL/H/7936 – 22
TL/H/7936 – 23
Square-Wave Oscillator
Pulse Generator
TL/H/7936 – 24
TL/H/7936 – 25
Frequency Differencing Tachometer
VODC e A (f1 b f2)
TL/H/7936 – 26
9
Typical Applications (V a e 15 VDC) (Continued)
Frequency Averaging Tachometer
VODC e A (f1 a f2)
TL/H/7936 – 27
Squaring Amplifier (W/Hysteresis)
Bi-Stable Multivibrator
TL/H/7936 – 29
TL/H/7936–28
Differentiator (Common-Mode
Biasing Keeps Input at a VBE)
‘‘OR’’ Gate
feAaBaC
TL/H/7936 – 31
AV e
1
2
TL/H/7936 – 30
‘‘AND’’ Gate
Difference Integrator
feA#B#C
TL/H/7936–32
TL/H/7936 – 33
10
Typical Applications (V a e 15 VDC) (Continued)
Low Pass Active Filter
fO e 1 kHz
TL/H/7936 – 34
Staircase Generator
VBE Biasing
AV j b
R2
R1
TL/H/7936 – 35
TL/H/7936 – 36
Bandpass Active Filter
fo e 1 kHz
Q e 25
TL/H/7936 – 37
11
Typical Applications (V a e 15 VDC) (Continued)
Low-Frequency Mixer
TL/H/7936 – 38
Free-Running Staircase Generator/Pulse Counter
TL/H/7936 – 39
12
Typical Applications (V a e 15 VDC) (Continued)
Supplying IIN with Aux. Amp
(to Allow Hi-Z Feedback Networks)
TL/H/7936 – 40
One-Shot Multivibrator
PW j 2 c 106C
*Speeds recovery.
TL/H/7936 – 41
Non-Inverting DC Gain to (0,0)
TL/H/7936 – 42
13
Typical Applications (V a e 15 VDC) (Continued)
Channel Selection by DC Control (or Audio Mixer)
TL/H/7936 – 43
14
Typical Applications (V a e 15 VDC) (Continued)
Power Amplifier
TL/H/7936 – 44
One-Shot with DC Input Comparator
a
Trips at VIN j 0.8 V
VIN must fall 0.8 V
a
prior to t2
TL/H/7936 – 45
High Pass Active Filter
TL/H/7936 – 46
15
Typical Applications (V a e 15 VDC) (Continued)
Sample-Hold and Compare with New a VIN
TL/H/7936 – 47
Sawtooth Generator
TL/H/7936 – 48
16
Typical Applications (V a e 15 VDC) (Continued)
Phase-Locked Loop
TL/H/7936 – 49
Boosting to 300 mA Loads
TL/H/7936 – 50
17
Split-Supply Applications (V a e a 15 VDC & Vb e b15 VDC)
Non-Inverting DC Gain
TL/H/7936 – 51
AC Amplifier
TL/H/7936 – 52
18