Development Of Omnidirectional Collinear Arrays With Beam

DEVELOPMENT OF OMNIDIRECTIONAL COLLINEAR ARRAYS WITH BEAM STABILITY FOR BASE STATION AND MOBILE APPLICATIONS A thesis submitted in fulfilment of the requirements for The degree of Master of Engineering Robert A. Daly School of Electrical and Computer Engineering College of Science Engineering and Health RMIT University November 2013 RMIT UNIVERSITY Development of omnidirectional collinear arrays with beam stability for base station and mobile applications II
Robert A. Daly RMIT University Development of omnidirectional Collinear arrays with beam stability For base station and Mobile applications III
A thesis submitted in fulfilment of the requirements for The degree of Master of Engineering Robert A. Daly School of Electrical and Computer Engineering College of Science Engineering and Health RMIT University 2014 IV
Declaration ere due acknnowledgmen
nt has been made; the work is tha
at of the I certify that except whe
uthor alone; the work has not beenn submitted previously in whole or iin part to qu
ualify for au
an
ny other acaademic awarrd; the conttent of the thesis t
is the result of w
work which has h been caarried out siince the official commeencement daate of the approved ressearch progrram; any ed
ditorial workk, paid or unp
paid, carriedd out by a third party is accknowledgedd. obert Andrew
w Daly Ro
211/11/2013 V
ACKNOWLEDGEMENTS The master degree has been very challenging and rewarding, whilst being full time employed in a role as Senior Electrical Engineer in antenna design. My day to day activity and after hours commitments to my family including my elderly parents have made this degree that much harder. Without the encouragement and support of my family and work colleagues, this degree may not have been possible. I would therefore like to thank my parents, for their encouragement and good wishes along the way. I would like to acknowledge my wife Lisa, and daughter Annie for their continued support and encouragement. Annie has helped with typing up some of the thesis chapters. Acknowledgements go to My Supervisors Doctor Wayne Rowe, and Professor Kamran Ghorbani, for guidance towards the thesis and review of papers and chapters during my candidature. Acknowledgements go to my fellow researchers for introducing me to university life, and the advice and friendly support they have given along the way. Acknowledgements to Mr. Mark Mezzapica and Mr. Steve Jacques, RF Industries Pty Ltd for allowing the time off work to attend the crucial meetings and for funding of the materials, the use of the far field antenna test range facilities and network analyser to verify the experimental results and for lodging a provisional patent to help protect the I.P of the invention during the project. Acknowledgements to Mr. Neville Brown, for proof reading of the chapters VI
GLOSSARY AUT Antenna under test CDMA Code division multiple access: mode of transmission CPW Coplanar waveguide CST Computer Simulation Technology: Company name COCO Coaxial collinear BER Bit error rate: data integrity parameter dBd Decibel power ratio relative to a dipole used to qualify antenna gain dBi Decibel power ratio relative to an isotropic radiator used to qualify directivity dBc Decibel power ratio below carrier level used to qualify PIM D.C Direct Current Ɛr Dielectric constant real EM Electromagnetic GCPW Grounded coplanar waveguide MIMO Multiple input multiple output MMIC Monolithic microwave integrated circuit PIM Passive intermodulation GRP Glass reinforced plastic PIP Peak Instantaneous Power PET Polyethylene terephthalate: A polymer used for flexible printed circuit substrate RFID Radio frequency identification TEM Transverse Electromagnetic TDMA Time division multiple access: transmission UHF Ultra high frequency 300‐1000 MHz VSWR Voltage Standing Wave Ratio WIFI wireless delivered internet service VII
of Contents
Table o
GLLOSSARY………
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……………………
…….VII LIST OF TABLESS ................................................................................................................................... X LIST OF FIGUREES ................................................................................................................................ XI CH
HAPTER 1 ..................................................................................................................................... 1‐16 1.1 INTRODUCTION ...................................................................................................................................... 1‐17 1.2 PROBLEM STATEEMENT ............................................................................................................................. 1‐19 1.22.1 Beam tilt.................................................................................................................................... 1‐19 1.22.2 PIM “Pa
assive intermo
odulation” ................................................................................................ 1‐27 1.22.3 PIP “Peak Instantane
eous Power” ............................................................................................. 1‐29 1.3 AIMS OF THE THHESIS ................................................................................................................................ 1‐31 1.4 THESIS OBJECTIVVES .................................................................................................................................. 1‐32 1.5 SPECIFICATIONSS ..................................................................................................................................... 1‐34 1.55.1 Limitatiions ................................................................................................................................ 1‐35 1.6 LITERATURE REVVIEW ................................................................................................................................ 1‐36 1.66.1 Self‐cho
oking related to dipole arraay antennas ........................................................................ 1‐36 1.7 ANTENNAS FOR BASE STATION AAPPLICATIONS .............................................................................................. 1‐39 1.77.1 Antenna specification
ns regulation .......................................................................................... 1‐39 1.77.2 Charactteristics of the
e collinear arrray antenna ........................................................................ 1‐40 1.77.3 Series ffed arrays ........................................................................................................................ 1‐41 1.77.4 Centre ffed array ante
enna ......................................................................................................... 1‐47 1.77.5 Corpora
ate fed colline
ear arrays ................................................................................................. 1‐52 1.77.6 Planar o
omnidirection
nal arrays.................................................................................................. 1‐56 1.8 DESIGN LIMITATTIONS IN ALL BASE STATION ANTE NNAS ................................................................................. 1‐58 CH
HAPTER 2 “COPLANAR
R WAVEGUID E (CPW)” FEED NETWORK WITH CENTREE FEED ............ 2‐61 2.1 STRUCTURE OF TTHE PLANAR CEN
NTRE FEED NETWO
ORK .................................................................................... 2‐61 2.11.1 The feeed network com
mponent impeedance contro
ol .................................................................. 2‐63 2.11.2 Feed neetwork compo
onent phase coontrol ................................................................................ 2‐64 2.11.3 Feed neetwork current distribution .......................................................................................... 2‐64 2.11.4 Relation
nship to a coa
axially fed archhitecture ............................................................................ 2‐65 2.2 REQUIREMENTSS AND LIMITATION
NS FOR THE PLAN
NAR CENTRE FEED
D NETWORK .................................................... 2‐66 2.3 FEED NETWORK DESIGN FOR A PLANAR COLLINEAAR ARRAY ............................................................................. 2‐67 2.33.1 Theorettical design off the feed netw
work .................................................................................. 2‐67 2.33.2 Circuit m
model results ................................................................................................................ 2‐70 2.33.3 The finiite ground “Co
oplanar wavegguide (CPW)”” and secondarry slot transm
mission line ........ 2‐73 V
VIII
2.33.4 Close prroximity of the transmissionn lines ................................................................................ 2‐75 2.33.5 Antenna port T‐juncttions ......................................................................................................... 2‐84 2.33.6 Phase ccompensation slotline term ination ............................................................................... 2‐89 2.33.7 Parameetric sweep off the slotline trrack width ......................................................................... 2‐91 2.4 ELECTROMAGNEETIC SIMULATION
N OF THE PLANARR CENTRE FEED N
NETWORK ........................................................ 2‐93 2.44.1 Electrom
magnetic mod
del results ................................................................................................. 2‐94 2.5 SUMMARY ............................................................................................................................................ 2‐96 CH
HAPTER 3 A BROADBA
AND PRINTED
D DIPOLE ELEM
MENT FOR A C
COLLINEAR ARRRAY ............... 3‐98 3.1 INTRODUCTION ...................................................................................................................................... 3‐98 3.1 A REVIEW OF EXXISTING BROADBAAND AND PLANA R DIPOLES ............................................................................ 3‐99 3.2 DESIGN CONSIDERATIONS FOR AAN OMNIDIRECTIOONAL COLLINEARR ARRAY DIPOLE EELEMENT ................................ 3‐103 3.22.1 The transition from ccylindrical to pplanar topolog
gy ............................................................... 3‐104 3.3 THE PROPOSED PLANAR DIPOLE ............................................................................................................... 3‐107 3.33.1 Proposeed planar dipo
ole structure ........................................................................................... 3‐107 3.33.2 Proposeed planar dipo
ole characteris
istics ................................................................................. 3‐109 3.33.3 The pla
anar dipole optimisation meethod ................................................................................ 3‐110 3.33.4 Planar d
dipole inside rradome .................................................................................................. 3‐113 3.33.5 Dipole p
performance iin the array ............................................................................................ 3‐116 3.33.6 Radiatio
on performan
nce of the plannar dipole arra
ay elements ................................................ 3‐116 3.4 SUMMARY .......................................................................................................................................... 3‐117 CH
HAPTER 4 ROBUST CO
OPLANAR WAV
VE GUIDE TO MICROSTRIP TRANSITION .................... 4‐118 4.1 INTRODUCTION .................................................................................................................................... 4‐118 4.2 LITERATURE REVVIEW OF BROAD BBAND “COPLANAAR WAVEGUIDE (CPW)”
(
TO MICROSTRIP TRANSITTIONS .............. 4‐119 4.22.1 Perform
mance goals fo
or the “Coplannar waveguide (CPW)” to M
Microstrip trannsition ............ 4‐126 4.3 THE STRUCTUREE AND OPERATION
N OF THE PASSIV
VE COUPLING CIRC
CUIT ............................................................ 4‐126 4.4 BROADBAND PAASSIVE TRANSITIO
ON PERFORMANCCE ..................................................................................... 4‐129 4.5 BACK TO BACK TTRANSITION MEAASUREMENT .............................................................................................. 4‐132 4.6 THE PASSIVE COUPLING ASSEMBLY .......................................................................................................... 4‐134 4.7 SUMMARY .......................................................................................................................................... 4‐137 CH
HAPTER 5 “COPLANAR
R WAVEGUID E(CPW)” FED OMNIDIRECT
TIONAL COLLI NEAR ARRAY
Y WITH FR
REQUENCY IN
NDEPENDENT BEAM DIRECTTION ............................................................................ 5‐138 5.1 STRUCTURE OF TTHE OMNIDIRECTTIONAL ARRAY W
WITH BEAM STABILITY ............................................................ 5‐139 5.2 IMAGES OF THE SIX ELEMENT “C
COPLANAR WAVEEGUIDE (CPW)” ARRAY COMPON
NENTS. .................................... 5‐141 5.22.1 Array co
onstruction .................................................................................................................. 5‐144 5.3 PERFORMANCE EXPECTATIONS ................................................................................................................ 5‐147 IX
X
5.4 ARRAY ANTENNA PERFORMANCE ............................................................................................................ 5‐148 5.44.1 Radiatio
on pattern me
easurement teechnique .......................................................................... 5‐149 5.44.2 Simulatted and measu
ured radiationns patterns ....................................................................... 5‐154 5.44.3 Gain ch
haracteristics ................................................................................................................ 5‐157 5.44.4 Peak instantaneous p
power perform
mance measu
ured ............................................................. 5‐159 5.44.5 Passivee intermodulattion performaance measured
d ................................................................. 5‐161 5.5 POWER HANDLING PERFORMANCE .......................................................................................................... 5‐161 5.55.1 Power m
measurementt method ................................................................................................ 5‐162 5.55.2 Power ttest results ................................................................................................................... 5‐162 5.6 MECHANICAL CHARACTERISTICS ............................................................................................................. 5‐164 5.7 FIXED BEAM TILTT ARRAY ......................................................................................................................... 5‐165 5.77.1 Down ttilt performancce ........................................................................................................... 5‐167 5.8 SUMMARY .......................................................................................................................................... 5‐171 HAPTER 6 CH
THESIS OVE
ERVIEW AND PPROPOSED FU
URTHER WOR
RK....................................... 6‐172 6.1 CHAPTER 1 SUM
MMARY ‐ BACKGROUND INFORMA
ATION ............................................................................... 6‐172 6.2 CHAPTER 2 SUM
MMARY ‐ FEED NETWORK ................................................................................................. 6‐173 6.3 CHAPTER 3 SUM
MMARY ‐ PLANAR
R DIPOLE .................................................................................................. 6‐174 6.4 CHAPTER 4 SUM
MMARY ‐ “COPLA
ANAR WAVEGUID
DE (CPW)” TO MICROSTRIP PAS
SSIVE COUPLER ......................... 6‐174 6.5 CHAPTER 5 SUM
MMARY ‐ COMPLLETE ARRAY PERFFORMANCE EVALUATION ....................................................... 6‐176 6.6 CONCLUSION ....................................................................................................................................... 6‐177 6.7 PROPOSED FUTU
URE WORK ...................................................................................................................... 6‐177 6.77.1 Furtherr developmentt to improve rradio site efficciency .......................................................... 6‐177 6.77.2 Beam tilt control usin
ng new technoology. .............................................................................. 6‐178 6.77.3 Beam d
direction contrrol to allow foor frequency reeuse ............................................................ 6‐178 List of Tables
Ta
able 1-1 Characcteristics of the three
t
convention
nal array feed ty
ypes ............................................................................. 1-18 Ta
able 1-2 Propossed electrical specifications of th
he collinear arra
ay prototype ................................................................. 1-34 Ta
able 1-3 Propossed mechanical specifications o
of the collinear array
a
prototype .........................
.
................................... 1-35 Ta
able 1-4 typical base station antenna specifica
ations ..................................................................................................... 1-60 Ta
able 4-1 The pe
erformance spec
cifications for th
he passive coup
pler .............................................................................. 4-126 Ta
able 4-2 Compa
arison the propo
osed passive co upling transition
n to references (measured
(
backk-to-back results)..... 4-134 Ta
able 5-1 Electriccal specification
ns (from Chapte
er 1) .................................................................................................... 5-147 Ta
able 5-2 Beam tilt
t characteristic
cs (in degrees) . ........................................................................................................... 5-157 Ta
able 5-3 Half po
ower beam width
hs (in degrees) ........................................................................................................... 5-157 Ta
able 5-4 Pattern
n characteristics
s for the six elem
ment array with frequency independent beam direction .................. 5-157 Ta
able 5-5 Power measurement with
w thermal mo nitoring............................................................................................... 5-163 Ta
able 5-6 Mecha
anical characteristics ................ ........................................................................................................... 5-165 Ta
able 5-7 Tilt sta
ability characteristics of omnidirrectional array fe
eed technologie
es....................................................... 5-166 X
List of Figures
Fig 1-1 Frequency dependent beam tilt in series fed array ........................................................................................... 1-20 Fig 1-2 Zero tilt array vs. series fed array when invert mounted ................................................................................... 1-21 Fig 1-3 A base antenna that has overshot the fringe area ............................................................................................ 1-22 Fig 1-4 The series fed base antenna corrected using beam tilt to cover the fringe area .............................................. 1-22 Fig 1-5 Beam tilt in base station arrays to allow for frequency reuse............................................................................ 1-23 Fig 1-6 Mechanical beam tilt as used with unidirectional arrays ................................................................................... 1-24 Fig 1-7 Electrical beam tilt in omnidirectional and unidirectional antenna arrays.......................................................... 1-25 Fig 1-8 Field coverage prediction map with no beam tilt [9] .......................................................................................... 1-26 °
Fig 1-9 Coverage map with 5 beam tilt [9] ................................................................................................................... 1-26 Fig 1-10 Intermodulation products ................................................................................................................................ 1-27 Fig 1-11 PIM test setup ................................................................................................................................................ 1-29 Fig 1-12 Non-destructive ionisation test ....................................................................................................................... 1-30 Fig 1-13 The typical response for inter element isolation............................................................................................. 1-37 Fig 1-14(a) Cylinder and (b) planar choke analogue showing the self-choke function ................................................. 1-38 Fig 1-15 Back to back choke impedance to odd mode on two-element conventional sleeve dipole ........................... 1-38 Fig 1-16 Schematic of a series fed array ...................................................................................................................... 1-41 Fig 1-17 Meander type collinear array .......................................................................................................................... 1-42 Fig 1-18 Transposed cable antenna array .................................................................................................................... 1-43 Fig 1-19 Series fed suppressor collinear....................................................................................................................... 1-43 Fig 1-20 Series fed annular slot fed collinear array: section view ................................................................................. 1-45 Fig 1-21 Planar double-sided PCB series fed array [24] ............................................................................................... 1-46 Fig 1-22 Series fed arrays configured for bidirectional coverage ................................................................................. 1-46 Fig 1-23 Pattern behaviour of an omnidirectional centre fed dipole array.................................................................... 1-47 Fig 1-24 Schematic of unidirectional centre fed array with a side feed ......................................................................... 1-49 Fig 1-25 Schematic of an omnidirectional centre fed array with a routed feed ............................................................. 1-49 Fig 1-26 The Marconi- Franklin antenna [27] ............................................................................................................... 1-50 Fig 1-27 Centre fed suppressor collinear antenna ........................................................................................................ 1-50 Fig 1-28 Section view of centre fed collinear array antenna [7] ................................................................................... 1-52 Fig 1-29 Schematic of a corporate fed array ................................................................................................................. 1-53 Fig 1-30 A cluster of 120° panel array antennas .......................................................................................................... 1-53 Fig 1-31 Tri-sector arrangement of corporate fed panel arrays in a cellular network ................................................... 1-54 Fig 1-32 Corporate fed folded dipole array (RFI BA80-67) [2] ..................................................................................... 1-54 Fig 1-33 Corporate fed printed sleeve dipole array [2] ................................................................................................. 1-55 Fig 1-34 Corporate fed sleeve dipole array with custom feed network [31] .................................................................. 1-55 Fig 1-35 “Coplanar waveguide (CPW)” based planar centre fed array [34] ................................................................. 1-57 Fig 1-36 CPW based planar series fed array [36] ........................................................................................................ 1-58 Fig 2-1 The basic architecture of the planar array feed network .................................................................................. 2-62 Fig 2-2 CPW input with slot lines feeding the antenna ports ........................................................................................ 2-62 Fig 2-3 The centre feed point comprising of a CPW with series impedance transformer ............................................ 2-63 Fig 2-4 The end of the feed network showing the slot lines and the return to virtual ground ....................................... 2-64 Fig 2-5 Surface current vectors ..................................................................................................................................... 2-65 Fig 2-6 The line and terminal impedances of the planar centre feed network .............................................................. 2-69 Fig 2-7 the theoretical feed network schematic............................................................................................................ 2-69 Fig 2-8 Input reflection coefficient magnitude (circuit model) ........................................................................................ 2-70 Fig 2-9 Transmission loss characteristics of ports 2 to 7 (circuit model)...................................................................... 2-71 Fig 2-10 phase response for ports 2 to 7 (circuit model)............................................................................................... 2-71 Fig 2-11 Transmission loss characteristics of ports 8 to 13 (circuit model)................................................................... 2-72 Fig 2-12 Phase response for ports 8 to 13 (circuit model) ............................................................................................ 2-72 Fig 2-13 Geometry of CPW transmission line model .................................................................................................... 2-73 Fig 2-14 Port Impedance of the 85Ω CPW (Smith chart normalised to 85Ω) .............................................................. 2-74 Fig 2-15 Geometry for 127Ω slot line model ................................................................................................................. 2-74 Fig 2-16 Impedance of the 127Ω slot line (Smith chart normalised to 127Ω) .............................................................. 2-75 Fig 2-17 Schematic layout for the 5 port feed network evaluation ............................................................................... 2-76 Fig 2-18 CPW termination stub length reference .......................................................................................................... 2-77 Fig 2-19 Maximum port phase difference with variable CPW stub length (in mm) ....................................................... 2-78 Fig 2-20 Reflection coefficient magnitude with variable CPW stub length (mm).......................................................... 2-78 Fig 2-21 Maximum port magnitude difference with variable CPW stub length (mm) .................................................... 2-79 Fig 2-22 CPW ground track geometry reference .......................................................................................................... 2-80 Fig 2-23 Maximum port phase offset with change in CPW ground track width ............................................................. 2-80 Fig 2-24 Port level with change in CPW ground track width (in mm) ............................................................................ 2-81 Fig 2-25 |S11| vs. frequency with change in CPW ground track width (in mm) ............................................................. 2-81 Fig 2-26 CPW discontinuity planar power divider ................................................................................................... 2-82 Fig 2-27 Port phase with change in CPW discontinuity gap width (in mm) .................................................................. 2-83 Fig 2-28 Port level with change in CPW discontinuity gap width (in mm) ..................................................................... 2-83 Fig 2-29 Reflection coefficient magnitude of feed network with change in CPW discontinuity gap width (in mm) ...... 2-84 Fig 2-30 Schematic of the three element slotline sub-array feed network with variable parameters ........................... 2-85 Fig 2-31 Port characteristics where the dipole port gap width is 0.2mm (EM) model ................................................... 2-86 Fig 2-32 Port characteristics where the dipole port gap width is 0.5mm (EM) model ................................................... 2-87 Fig 2-33 Port characteristics where the dipole port gap width is 0.6mm (EM) model ................................................... 2-87 Fig 2-34 Port characteristics where the dipole port gap width is 0.7mm (EM) model ................................................... 2-88 Fig 2-35 Port characteristics where the slot line stub length is 1 mm (EM) model........................................................ 2-89 XI
Fig 2-36 Port characteristics where the slotline stub length is 9 mm (EM) model......................................................... 2-90 Fig 2-37 Port characteristics where the slotline stub length is 17 mm (EM) model....................................................... 2-90 Fig 2-38 Port characteristics where the slotline track is 4mm wide (EM) model ........................................................... 2-91 Fig 2-39 Port characteristics where the slotline track is 3.2mm wide (EM) model ........................................................ 2-92 Fig 2-40 Port characteristics where the slotline track is 2.4 mm wide (EM) model ....................................................... 2-92 Fig 2-41 The CPW array feed network port configuration ............................................................................................ 2-93 Fig 2-42 Phase and amplitude of the lower array ports (EM model) ............................................................................ 2-94 Fig 2-43 Phase and amplitude of the upper array ports (EM model) ............................................................................ 2-95 Fig 2-44 |S11| of the planar feed network (EM model) ................................................................................................... 2-96 Fig 3-1 The broad banding technique used for [44] .................................................................................................... 3-100 Fig 3-2 (a) Biconical dipole (b) Bowtie dipole [16] ...................................................................................................... 3-101 Fig 3-3 Sleeve dipole with virtual high frequency source: .......................................................................................... 3-102 Fig 3-4 Cylindrical to planar element evolution .......................................................................................................... 3-105 Fig 3-5 A cylindrical sleeve dipole and the analogue planar dipole ........................................................................... 3-106 Fig 3-6 |S11| for the conventional cylindrical dipole compared to the planar dipole ..................................................... 3-106 Fig 3-7 Proposed planar dipole schematic and interface to feed network (with parameters) .................................... 3-108 Fig 3-8 Current distribution for dipoles with different CPW feed methods .................................................................. 3-109 Fig 3-9 Final dimensions of the proposed planar dipole (in mm) ............................................................................... 3-110 Fig 3-10 The comparison models used (a) slots and notches filled in (b) no filling..................................................... 3-111 Fig 3-11 |S11| of the proposed dipole with and without notch and slot elements ........................................................ 3-112 Fig 3-12 The |S11| of a proposed planar dipole as the notch depth is adjusted.......................................................... 3-113 Fig 3-13 Single element planar dipole with radome loading ...................................................................................... 3-113 Fig 3-14 Feed point impedance of a planar dipole mounted inside 76mm diameter radome .................................... 3-114 Fig 3-15 Normalised planar dipole reflection coefficient magnitude impedance (reference is 129Ω) ........................ 3-114 Fig 3-16 E field current distribution at lowest frequency= 850MHz ............................................................................. 3-115 Fig 3-17 E field current distribution at the highest frequency = 960MHz .................................................................... 3-115 Fig 3-18 The typical (a) elevation and (b) azimuth radiation pattern of dipole ........................................................... 3-117 Fig 4-1 Capacitive coupled CPW to microstrip transition [36]. .................................................................................... 4-120 Fig 4-2 A transition with common source track and electromagnetically coupled ground [37] .................................. 4-121 Fig 4-3 A transition with common source track and electromagnetically coupled ground [38] ................................... 4-122 Fig 4-4 A transition similar to Fig 4-2 but with tapered conductors [39] ..................................................................... 4-123 Fig 4-5 An electromagnetically coupled CPW to Microstrip transition, common ground [40]...................................... 4-124 Fig 4-6 These are examples of state of the art performance coupler circuits [50] ...................................................... 4-125 Fig 4-7 Example of directly connected CPW to Microstrip transition [42] .................................................................. 4-125 Fig 4-8 Schematic layout of the proposed passive CPW to Microstrip transition ....................................................... 4-127 Fig 4-9 The layout of the coupling transition bottom layer showing coupling pad alignment ...................................... 4-128 Fig 4-10 low pim edge connector application ............................................................................................................. 4-129 Fig 4-11 Back to back coupler test fixture ................................................................................................................... 4-130 Fig 4-12 Back to back coupling test with parametric slot depth ................................................................................. 4-131 Fig 4-13 A parametric sweep of the passive coupler with various Microstrip probe lengths ...................................... 4-132 Fig 4-14 Modelled and measured S parameters of transition characterised back to back ......................................... 4-133 Fig 4-15 Side view of passive transition attached to the PET CPW feed PCB ........................................................... 4-135 Fig4-16 Passive coupler alignment test configuration................................................................................................ 4-136 Fig4-17 Parametric sweep of the passive coupler substrate alignment ..................................................................... 4-136 Fig 5-1 A schematic of the overall array components ................................................................................................ 5-140 Fig 5-2 Components of the “Coplanar waveguide (CPW)” array with an antenna using the same radome ............... 5-141 Fig 5-3 The foam spacers used to support the array PCB .......................................................................................... 5-142 Fig 5-4 The passive coupler circuit attached to the array .......................................................................................... 5-142 Fig 5-5 The full length of the “Coplanar waveguide (CPW)” array .............................................................................. 5-143 Fig 5-6 close-up of “Coplanar waveguide (CPW)” transmission line........................................................................... 5-143 Fig 5-7 close-up of upper dipole .................................................................................................................................. 5-144 Fig 5-8 close-up of top element ................................................................................................................................... 5-144 Fig 5-9 Planar dipole structure .................................................................................................................................... 5-145 Fig 5-10 Cut away representation of the array ........................................................................................................... 5-146 Fig 5-11 Measured and simulated reflection coefficient magnitude for the proposed collinear array ........................ 5-149 Fig 5-12 range test control and logging equipment ..................................................................................................... 5-150 Fig 5-13 setting up antenna for testing........................................................................................................................ 5-151 Fig 5-14 Ground reflection range ............................................................................................................................... 5-152 Fig 5-15 Cartesian plot of array elevation pattern ...................................................................................................... 5-152 Fig 5-16 Polar axis plot of elevation pattern shown in fig 5.15 ................................................................................... 5-153 Fig 5-17 Model and measured elevation patterns 850 MHz ....................................................................................... 5-154 Fig 5-18 Model and measured elevation patterns 870 MHz ....................................................................................... 5-155 Fig 5-19 Model and measured elevation patterns 910 MHz ....................................................................................... 5-155 Fig 5-20 Model and measured elevation patterns 930 MHz ....................................................................................... 5-156 Fig 5-21 Model and measured elevation patterns 960 MHz ....................................................................................... 5-156 Fig 5-22 Measured gain and modelled realized gain for the six element array........................................................... 5-159 Fig 5-23 Non-destructive ionisation test of the passive coupler ................................................................................. 5-160 Fig 5-24 Power testing configuration .......................................................................................................................... 5-163 Fig 5-25 Adjustment of the “Coplanar waveguide (CPW)” feed point position to control tilt ................................... 5-167 Fig 5-26 Vertical radiation pattern for the zero tilt collinear array............................................................................... 5-168 Fig 5-27 Vertical pattern for a 4° fixed beam tilt collinear array ................................................................................. 5-169 Fig 5-28 Reflection coefficient magnitude of array with 4°beam tilt compared to array with 0°beam tilt .................. 5-170 Fig 5-29 Gain comparison (pattern maximum) 4°tilted and 0° tilt array .................................................................... 5-170 XII
List of Appendices
APPENDIX A The topology of the lower coupling and second dipole of the planar collinear array .......................... 6-185 APPENDIX B The topology of the central feed point and inner two dipoles of the planar collinear array ................. 6-186 APPENDIX C The topology of the second last and last dipole in the planar collinear array ..................................... 6-187 APPENDIX D The complete planar collinear array topology ..................................................................................... 6-188 APPENDIX E First model trial of a ten element array ............................................................................................... 6-189 APPENDIX F The broad band gain response of a ten element array ....................................................................... 6-189 APPENDIX G The ten element “Coplanar waveguide (CPW)” array at 840 MHz ..................................................... 6-190 APPENDIX H The ten element “Coplanar waveguide (CPW)” array at 850 MHz ..................................................... 6-190 APPENDIX I The ten element “Coplanar waveguide (CPW)” array at 860 MHz ....................................................... 6-191 APPENDIX J The ten element “Coplanar waveguide (CPW)” array at 870 MHz ...................................................... 6-191 APPENDIX K The ten element “Coplanar waveguide (CPW)” array at 880 MHz ..................................................... 6-192 APPENDIX L The ten element “Coplanar waveguide (CPW)” array at 890 MHz ...................................................... 6-192 APPENDIX M The “Coplanar waveguide (CPW)” panel array (850-960 MHz, 15 dBi) ........................................... 6-192 APPENDIX N The reflection coefficient for the “Coplanar waveguide (CPW)” six element panel array .................... 6-193 APPENDIX O The azimuth pattern for the “Coplanar waveguide (CPW)” six element panel array .......................... 6-193 APPENDIX P The elevation pattern for the “Coplanar waveguide (CPW)” six element panel array ........................ 6-194 APPENDIX Q The broad band directivity response for the trough reflector array ..................................................... 6-194 APPENDIX R The spacing from the reflector to the array for the trough reflector array ............................................ 6-194 APPENDIX S The angle of the trough reflector ......................................................................................................... 6-195 APPENDIX T The reflection coefficient for omnidirectional and unidirectional array ................................................ 6-195 1-13
Abstract: Since the 1800’s, inventors have considered electromagnetic radiation as a medium for communications between ship and shore and base to mobile. Generally, these were narrow band services, supported by simple series fed arrays. These arrays are limited in bandwidth by an inherent tapered phase profile which exhibits an undesirable vertical beam angle shift with excessive change in frequency. This performance is not suitable for the modern broad band networks. Antennas that are suitable for these networks are the centre fed and corporate fed arrays. These antennas are relatively expensive and complex with a coaxial architecture to allow for the feed network to be routed up the core away from the dipoles. With a symmetrical tapered or linear phase profile, the radiation pattern remains on the horizon over at least 10% bandwidth. This thesis aims to presents the development of an antenna which has a unique compact planar architecture, and has achieved the performance required for broad band communications channels in the 900 MHz band. With low cost; volume manufacture, PIM and PIP considered. A major achievement in this work has been the development of a planar passive coupled dipole array with coplanar feed network presented on a single sided low cost flexible PCB. In this format, the array can be re produced accurately at a fraction of the cost of the conventional sleeve dipole arrays. Major challenges in the implementation of such a solution, are in the substitution of a coaxial feed network with planar alternatives and still maintain symmetrical phase profile stability and a reasonable power rating. Due to PCB processing limits for track spacing, the array has been designed with a main transmission line track spacing of 0.638 mm resulting in a higher impedance of 86 Ohms. At this track spacing, the main transmission line can withstand a 2.1 kV ionisation test allowing for a PIP value of 1 Kw. To compliment this planar array, a specially designed planar dipole; matched to the new feed network scheme. With slots to augment the radiation pattern for improved directivity and reduced azimuth ripple, and notches to stagger tune the dipole to cover a broader than otherwise band and compensate for the reactance associated with close proximity to the slotline tracks. This dipole meets broad bandwidth and omnidirectional pattern specifications similar to cylindrical sleeve dipoles, while still maintaining inter‐stage isolation. To further compliment the array attributes, a robust coaxial cable connection to the thin and flexible substrate employs a novel passive coupler transition circuit. The proposed transition is 1-14
electromagnetically coupled to source and ground terminals of the array, eliminating the need for soldering, resulting in low loss, low PIM “Passive intermodulation” typically <‐148 dBc, This coupler also provides the impedance transformation from 70 Ohms to 50 Ohms An omnidirectional six element “coplanar waveguide (CPW)” fed collinear array was assembled from the planar feed network, dipole elements and passive coupler components. Most research in the area of printed antennas is focused on small low gain antennas. This research presents a high gain antenna printed array supporting six elements with a centre fed network, having a total length approximately 1.8 meters. This large array has the unique attributes of wide band, less than 1 degree beam tilt over a frequency range of 850 to 960 MHz (for a measured reflection coefficient magnitude greater than 14 dB), with low measured passive intermodulation below ‐140dBc “Decibel power ratio relative to transmit carrier”. The prototype collinear array can also withstand a wind loading of 240 km/h. The resulting planar collinear antenna array meets all of the electrical specifications for low power base station applications, including operation in the desired frequency band, high gain, and extremely stable radiation pattern. A gain of 10 dBi has been achieved with omnidirectional radiation. 1-15
Chapter 1
Since the disccovery of ele
ectricity and electromagn
netic radiatio
on in the 18800’s, inventtors have his phenome non as a me
edium for com
mmunicationns. The radio
o stations beeen actively exploiting th
ussed in modern networks need reliabble infrastruccture which enables conttact to all sttations in orrder to transsfer data and telephonyy. An antenn
na with this capability iss the colline
ear array. Deepending on
n its feed arrrangement, tthe radiation
n pattern rem
mains stablee in shape w
within the baand of operation. This thesis t
preseents an ante
enna which meets thesee requireme
ents in a un
nique planarr architecturre. The chal lenge in this design is the substituution of coaxial feed neetwork with planar alterrnative, and still maintains effective
e inter‐stage e isolation an
nd phase stability. It is designed to be low costt, with low p
passive interrmodulation and provide
es a zero es are norm
mally dominaated by complex and beeam tilt patttern. In the industry theese attribute
exxpensive pro
oducts. This cost reductioon has been achieved byy creating ann array of dip
poles and feeed networkss on a single sided flexiblle printed cirrcuit. Both ““coplanar waaveguide (CP
PW)” and slo
ot line feedeers have bee
en successfuully integrate
ed on this architecture. Some rese
earch has beeen active in
n the area off printed anttennas, mostt focus on sm
mall low gainn antennas. T
This new reesearch is intto a high gain antenna w
which is able to support six elemennts with a ce
entre fed neetwork. All components
c
s in this dessign have be
een optimized to providde a highly efficient peerformance w
with structurral integrity.
1--16
1..1
Introd
duction
W
With radio networks expa
anding and thhe introducttion of multicarrier digitaal radio syste
ems with increased den
nsity some new probleems arise, namely n
passsive inter‐m odulation and peak e more soph
histicated, thhey call for a higher instantaneouss power. As radio netwoorks become
offered by the new p
standard in antenna consstruction to deliver the enhanced performance syystems [1]. Th
he omnidirecctional array antenna hass three majo
or methods o
of feeding as s listed Table 1‐1. The sim
mplest is thee series fed a
array where each eleme
ent feeds thrrough the reesidual current to the neext element until a pointt of diminish ing return is reached. Se
eries feed neetworks are limited in op
perational baandwidth by frequency ddependent beam tilt. A more comp
plex approacch is the ceentre feed network, n
wh
hich is actuaally two serries feed neetworks backk‐to‐back witth 180° offseet in phase. The differential phase reelationship, rresults in almost compleete cancellattion of beam
m tilt in the ffar field. To prevent apeerture blocka
age, feed mally restricts the designn to heavierr tubular caables are shielded from the dipoless. This norm
co
onstructions that allow the cables too rout internally. The bro
oadest band feed network of the th
hree is corpo
orate feed. In this connfiguration, the same power level aand phase profile p
is ap
pplied to eacch element. The corporrate feed haas the advantage of broaadband perfformance an
nd a high po
ower rating as each ele ment sharess the load of o the applieed power. Losses L
in co
orporate feed networks are usually higher than for a centre fed array due to extra
a cabling ussed in the deesign. This lo
oss can direcctly affect the antenna effficiency andd result in the gain of th
he antenna b
being lower relative to itss overall lenggth. Hiistorically, arrray antenna
as for cellulaar base station applicatio
ons are handd assembled
d. This is to
o ensure reliiable constru
uction to meeet the requ
uirements of the new ccellular syste
ems, and en
nables indep
pendent low
w maintenancce service. Conventional centre fedd and corpo
orate fed om
mnidirection
nal arrays ca
an be compplex in desiggn and have
e many fastteners and soldered junctions. Theese attribute
es not only m
make manuaal assembly a
a lengthy an d expensive process, ut also are likely l
to con
ntribute to tthe generatio
on of PIM which w
is reallised as noisse in the bu
reeceivers passs band. PIM affects both analogue an
nd digital sysstems equallyy [2]. Th
he multi‐carrrier environm
ment of moddern cellular systems brin
ngs to light a nother phen
nomenon kn
nown as PIP “peak instantaneous poower”. PIP is a function of multipl e carriers co
ombining into one antenna and can
n result in prremature equipment failures. The isssues of PIM
M and PIP oints of conccern for currrent base station s
arrayy antennas aas the availa
ability of arre crucial po
to
ower spacee is limite
ed. The economicaal alternative is to combine multiple 1--17
transmitters/services into one antenna, resulting in higher incidences and consequences of PIP and PIM. Table 1‐1 Characteristics of the three conventional array feed types Feed type Advantage Disadvantage 
Low cost 
Frequency dependent beam tilt Series fed array 
Simple construction 
Narrow band 
Light weight 
Sharp gain response 
Lower wind loading 
No invert mounting due to tilt 
Axis omnidirectional 
Max gain limited by mechanical stability and progressive phase errors 
Unpredictable coverage symmetric 
Zero beam tilt 
Heavy weight 
Can be invert mounted 
Relatively expensive, 
Broad band 

Beam tilt can be adjusted Prone to vibration fatigue over time 
Flat gain response 
High wind loading 
Can support transmitters 
Large radome diameter 
Complex construction 
Feed network must be routed inside antenna 
High parts count 
High labour costs Centre fed array combined 
High continuous power rating 
Highly predictable coverage 1-18

Zero beam tilt 
Heavy weight 
Can be invert mounted 
High wind loading 
Broad band 
Corporate fed array 
Beam tilt can be adjustable Relatively more expensive than centre feed 
Supports combined transmitter’s high continuous power rating 
Large radome diameter 
Complex construction 
High feeder losses 
Feed network must be routed inside antenna 
High parts count 
High labour costs 
Flat gain response 
High gain arrays are possible over a wide band 
Highly predictable coverage 1..2
Probllem statem
ment
Th
he advances in communications equuipment have resulted in
n a need forr more band
dwidth to allow for high
h speed data
a, digitally enncoded cellu
ular networkks such as CD
DMA system
ms. The ntennas neeeded for the
ese networkks must be capable of supporting g broad ban
ndwidths. an
M
Maintaining p
phase stabilitty is the cha llenge as ph
hase instability manifestss itself as shift in the veertical beam angle. This is known as bbeam tilt and
d will be disccussed in secction 1.2.1 Th
here are so
ome operatiional challe nges for baase station antennas iin this new
w age of co
ommunicatio
ons. The challenges relatte to the inte
egration of services into a finite specctrum. At so
ome point th
he frequencies combinaations allocated can mixx together aand generate
e a third orrder productt of 2A‐B an
nd is known as PIM “Paassive interm
modulation”. This subjecct will be discussed in section 1.2.2 yet an other problem related
d to bandw
width is PIP “Peak instantaneouss Power. PIP
P is generateed as two orr more transsmitters aree combined to t utilize ne antenna. The more tra
ansmitter poower sourcess involved th
he higher thee probability of signal on
peeaks combin
ning and ge
enerating a potentially catastrophicc power su rge known as Peak instantaneouss Power whiich results inn voltage sp
pike. This ph
henomenon will be disccussed in seection 1.2.3 1.2.1
Beam
m tilt W
When mobile radio system
ms were firstt introduced
d 50 years ag
go, the systeems were qu
uite basic an
nd narrow band b
by toda
ay’s standardds. Most haad single fre
equency requuirements th
hat were eaasily supportted by simple
e series fed ccollinear arraay designs. Later, networrks required the base stations to support s
mo
ore traffic aand broade
er bandwidtths, and too allow forr duplex ommunicatio
ons and multiple transm
mitters into one o antenna
a. The frequuency depen
ndent tilt co
th
hat is experieenced by seriies fed antennna arrays to
o these broad
der bandwiddth systems iis seen in Figg 1‐1. Zero
o tilt is seen at the designn frequency;; however sig
gnificant tilt may be exp
perienced att the upper and lower band edgess. This led to the intro
oduction of higher perfformance an
ntenna systems, such as the corporatte fed and ce
entre fed arrrays. These aarrays are ca
apable of caancelling out the frequen
ncy depende nt tilt. 1--19
Low
west freq
quency Deesign
freequency
Highest
frequency
Figg 1‐1 Frequeency dependent beam tiltt in series fed array Beeam tilt can be a significa
ant problem for high gain series fed arrays. As t he gain of an
n array is increased (i.ee. increasingg the numbeer of array elements), the effect of the beam tilt is w
the maximum ttilt exceeds the 3dB beam b
width,, causing th
he tilted prronounced where freequency beaam to appea
ar in a null oof the un‐tillted frequen
ncy pattern. A tilt value
e of 7° is tyypical for a seeries fed arra
ay at the low
wer edge of aan 8% bandw
width. Clearly, the series s
fed array a
is onlyy suitable fo
or narrow band b
base station servvices [3]. M
Mounting anttennas invertted to one aanother is an option to conserve toower space and stop seeries fed arraay tilting out of service. Series fed antennas when w
mounteed in this wa
ay, tilt in op
pposing direections at the band edgee frequencie
es, whereas as zero tilt antenna remains in seervice over a broad band as shown inn Fig 1‐2. 1--20
Opp
posing
seeries
fed array stayss in
Zero tilt arrays alw
ways remain
serv
vice
in service
Seriees fed arrayy at
loweest freque
ency
mayy tilt out of
servvice eries fed arraay when inve
ert mounted
d Figg 1‐2 Zero tilt array vs. se
It is apparent that series fed antennaas when ope
erated awayy from the ddesign freque
ency can prrovide a narrrow band off useable doown tilt. This may be dessirable in a nnarrow band
d service, where a singlee channel tra
ansmitter is location on high terrain
n and may ovver shoot the fringes 4 exhibits ho
ow a slight ttilt has impro
oved the off the coveragge area as shown in Fig 1‐3. Fig 1‐4
co
overage for aan elevated tterrain site. 1--21
Figg 1‐3 A base antenna tha
at has overshhot the fringe area e antenna coorrected using beam tilt tto cover the ffringe area Figg 1‐4 The series fed base
1--22
perating a seeries fed antenna array ooffset from th
he design fre
equency tiltss the main lobe. Op
Th
he level of tillt determined by the mannipulation off Eqn. (1.1)
cos
cos
c
⇒
cos
(1.1) [44]
Progresssive
θ
90°
= beam tilt aangle Alternatively down d
tilt ca
an be set too a particular value for a number oof reasons in
n a radio neetwork, for example, in
nterference suppression
n or simply to enhancee the covera
age in a reestricted area as seen above. Dow n tilt can also be used as a metho d used to attenuate crrosstalk from
m adjacent ce
ells allowing ffor frequenccy reuse [5], [3] as shownn in Fig 1‐5. Figg 1‐5 Beam ttilt in base sttation arrayss to allow forr frequency rreuse In a cellular neetworks, pattern shapingg including beam tilt is an
n option to ffirstly limit th
he size of he cell and seecondly incrrease the carrrier to noise C/N ratio by up to 10 dB [3], [6]. In some th
ap
pplications, m
mechanical b
beam tilt is uused [3]. Fig 1‐6 shows h
how mechannical down tilt is used 1--23
independentlyy of electrical tilt. Thee rear lobe tilts t
up, resulting in a bbeneficial null being directed close in behind th
he antenna. In Fig 1‐7 it iss shown thatt electrical tiilt is effectivve when used with omniidirectional a
arrays as th
he entire aziimuth pattern is tilted ddown to the
e required angle. a
For uunidirectional arrays where down ttilt is used to
o fine‐tune thhe cell coverrage, electriccal tilt will allso down tiltt the rear lobe which may m not be desirable. d
EElectrical tilt can be con
ntrolled rem
motely by me
echanical are in the acctuators attaached to phase shifting eelements witthin the arrayy [1]. Otherrs methods a
fo
orm of a rheo
ostat, an arc shaped con ductor element with a contactor to ttap at variab
ble phase lengths on thee transmissio
on line. Pateents protect m
most of these tilt mecha nisms [7], [8
8]. Fig 1‐6 M
Mechanical b
beam tilt as used with un
nidirectional arrays 1--24
Omnidirectional
Unidirectionnal
unidirectiona
al antenna a rrays Figg 1‐7 Electriccal beam tilt in omnidire ctional and u
he characterristics of the antenna pattern in conjunction with terraiin path losss can be Th
em
mployed to p
predict coverage. Fig 1‐88 and Fig 1‐9
9 illustrate how effectivee this mappin
ng is, the fo
ormer showin
ng a coveragge mapping ffor Newland Hill SA for a zero tilt basee station [9], and the lattter with a 5
5° down tilt applied. An i ncreased are
ea for freque
ency reuse iss indicated in
n Fig 1‐9. 1--25
overage pred
diction map with no beam tilt [9] Figg 1‐8 Field co
Reuse of a channel could be
n this region allowed in
Fig 1‐9 C
Coverage maap with 5° bea
am tilt [9] 1--26
1.2.2
PIM
M “Passive in
ntermodulattion” Paassive interm
modulation (PIM) (
is cau sed by the mixing of tw
wo or more carriers ressulting in higher order products. PIM P can occuur in solder joints and loosely braidded copper [10] [11] [112]. Ferromaggnetic materials such as ferrites also
o generate PIM. Any platting containing nickel is likely to cau
use PIM. Loo
osely fitting ttubes that are only partially solderedd can cause PIM. The within the most concerning mixes arre third ordeer products as they are most likely tto appear w
perational frrequency band of the syystem, no am
mount of filtering will erradicate PIM
M. The 3rd op
orrder emission
ns are at higher amplitudde than the o
other products (such as tthe 5th and 7
7th order) ass shown in Fig F 1‐10. Th
he 3rd order products arre generated
d under the conditions 2a‐b 2
and 2b
b+a, fall closee to the mix frequencies associated w
with the systtem TX channnels denoted
d a and b. If the carrier leve
els are stronnger than the
e normal exp
pected receivve level, the 3rd order ugh the recceiver noise threshold, as indicatedd in Fig 1‐1
10 This prroduct can break throu
intermodulation “noise” will increasee bit errorss in a digital system reeducing the network x system. caapacity. It will desensitise the receiveer channels iin a conventional duplex a b f2
f1 3
3
5
7
2
Receiver noise thresh
hold
5
7
3
2
4
3
Figg 1‐10 Interm
modulation p
products A conventionaal soldered ccoaxial cable interface to
o the dipole is historicallyy most susce
eptible to hat is knownn as PIM mitigation m
PIM in an arrray design. Designing this interfaace with wh
1--27
techniques should result in minimizing PIM. PIM mitigation can take the form of semi rigid coaxial lines, keeping the assembly clean and free of metal particles. The measurement of devices immunity to the generation of PIM is done using a test set that excites the potential for PIM in a device. To be effective the test set itself must have high PIM immunity. Shown in Fig 1‐11 is such a test set. It comprises of all the components necessary to couple two transmitters into one output that produces the intermodulation carrier. A receive band pass filter is used to detect only this product, and reject the fundamental carriers from the transmitters. Some notch reject filters are added to the transmit output which further suppress the fundamental carriers from reaching the receiver. To measure PIM levels of ‐150 dBc, the test set must be capable of measuring ‐160 dBc. The intermodulation product is such a low level; a low noise amplifier is needed to increase the resolution. 1-28
INTERMOD
ANALYSER 35 dB gain LNA DEVICE
Connect to
855 TX 2 SPECTRUM CABLE CABLE
LOW PIM 10 dB hybrid coupler
BAND PASS RX
RX
REJECT
BAND
TX1 TX2
Diplexer
850 TX 1 Figg 1‐11 PIM ttest setup 1.2.3
PIP ““Peak Instan
ntaneous Pow
wer” W
With advanced radio netw
works using m
multi‐carrierr digital schemes, there iis a statistica
al chance th
hat multiple transmitter carrier pow
wers can combine in‐ph
hase to geneerate a high
h voltage grradient withiin an array ccombiner/feeed network [2]. This volttage is capabble of destro
oying the insulation com
mponents in an array, inccluding coaxial cable diele
ectrics and innsulators. Fo
or all base sttation arrayss that suppoort multiple ccarrier systems, electricaal acceptance testing sh
hould also incclude a non‐destructive high pot testt at a peak le
evel of approoximately 2.1
1 kV [2]. 1--29
This test involves applying a high voltage to the components in order to detect and highlight faulty solder joints, intermittent contacts, moisture and contaminated dielectrics. Failure to detect these quality points can result in early system failure. For non‐destructive device testing, the applied voltage is gradually increased towards the peak voltage whilst monitoring the leakage current as shown in Fig 1‐12. If the current exceeds a pre‐set threshold or an arc occurs before reaching the test voltage the device has failed and the applied voltage resets. The voltage spike associated with PIP is not readily detected by conventional power meters. It is normally seen as noise affecting the bit error rate (BER). The measurement of PIP is not practical, and prevention is preferable. Eqn. 12 can be used to calculate the potential value for PIP based on the power, modulation index and the number of transmitters in use at a transmitter site. (12) M = modulation Peak to Average ratio dB Pca = Power per channel at antenna port in watts N = Number of channels in use Fig 1‐12 Non‐destructive ionisation test 1-30
1..3
Aims of the thessis
Th
his thesis willl present ressearch into aa new cost e
effective and
d environmenntally friendly option fo
or base statiion antennas that will not only eliiminate radiiation beam
m tilt with ch
hange in freequency, but also substa
antially reduuce the PIM potential by using a singgle copper la
ayer on a prrinted flexible substrate. A passive cooupling technique is emp
ployed to disstribute pow
wer to the an
ntenna elem
ments, allowing for a m
more compacct architectu
ure and exppansion to a a greater nu
umber of eleements for a a higher gainn if required
d. Other pla
anar omnidirrectional arrrays with zeero beam tilt found in the
e literature aare not as ad
daptable. A comprehen
nsive set of electromaggnetic simulaations and optimisationns are prese
ented to ormance posssible from the t printed planar p
array.. Another feature is acchieve the highest perfo
th
he entirely paassive couple
ed input, meeaning the base station a
array has no DC continuiity to the interfacing connector. Ele
ectromagnet ic coupling of o the groun
nd and sourcce lines of the t input onnector is eemployed for improved mechanical stability, and
d to providee a gradual transition co
fro
om the coaxial mode of o the conn ector to the
e “Coplanar waveguide (CPW)” Qu
uasi TEM “TTransverse electromagne
e
etic". The design has less environmental im
mpact than antennas a
deesigned usingg aluminium or brass tubbe sections w
with many so
older connecctions. Th
he outcomess from this thesis are eexpected to spawn furth
her researchh in the area
a of high sp
pecification antennas ussing printedd circuit bassed construcction. This novel type of feed neetwork is not restricted tto omnidirecctional base station ante
ennas, and m
may find app
plications fo
or smart anteennas and pa
atch arrays w
with the implementation of spatial divversity. 1--31
1..4
es
Thesis objective
Th
his thesis invvestigates a solution too the problem of freque
ency dependdent beam tilt t in an om
mnidirection
nal array, in a
a form that is suitable fo
or high volume manufaccturing and low cost. Th
he array will provide a VSSWR and raddiation pattern performance, which m
meets or excceeds the cu
urrently avaailable centrre fed anteennas. The
e feed netw
work will m
make use of o planar traansmission lines to form
m a compacct and efficiient centre feed netwo rk with gable phase prrofile [13] su
uitable for a a centre fed omnidirectional array. The feed nnetwork willl be well iso
olated from the radiatio
on element. The array will be flexible in such that the addition of more antennaa elements to
o the array iss possible without excee
eding a fixed radome diameter of p
suffficient mechanical stability with enhhanced PIM and PIP 633 mm. The array will provide peerformance by using passive couppling to the input, as opposed o
to a directly soldered co
onnection. TThe outcome
es of this ne w research w
will contribu
ute to the deevelopment of highly effficient planaar feed netw
works for coollinear arrayy antennas for f current aand next ge
eneration co
ommunicatio
ons systems. 1.1.1
Chaapter overvie
ew CH
HAPTER 2 outlines o
the design and performancce of a plan
nar feed nettwork for a zero tilt co
ollinear arrayy. The implem
mentation oof a “Coplanaar waveguide
e (CPW)” maain line with compact paarallel slot transmission lines to connnect to the
e dipoles is described. The motivation and teechniques ussed in deve
eloping thiss feed netw
work are exxplained, witth the obje
ective of maintaining a minimal wid
dth to the arrray. HAPTER 3 details the design of a nnew broadbaand planar dipole d
to su it the coplanar feed CH
neetwork desccribed in Ch
hapter 2. The planar dipole is shown s
to hhave the broadband peerformance o
of a cylindriccal dipole of similar width
h, enabled by the use of edge slots and notch feeatures in thee printed ele
ement. 1--32
CHAPTER 4 introduces a passive coupled “Coplanar waveguide (CPW)” to microstrip transition circuit that provides a rigid interface to the input connector of the collinear array. The coupler covers the 850 – 960 MHz band and provides a gradual transition from the “Coplanar waveguide (CPW)” modes to the microstrip mode. Mechanically, this transition is novel as it allows a flexible substrate to interface with a rigid laminated PCB providing an efficient coupling to the antenna without DC continuity. It also offers enhanced mechanical stability suitable for the direct connection of a coaxial cable or connector. In conjunction with the solder free printed circuit based array, the DC isolation helps provide exceptional PIM and PIP performance. A special low PIM cable transition is also developed and used to interface to the attached cable. CHAPTER 5 explains the integration of the sub‐assemblies to realise the complete printed collinear array with frequency independent beam direction, and presents a discussion on the simulated and experimental results. The power handling and mechanical performance of the array are also examined. Methods of providing a fixed beam tilt in the array are also investigated. CHAPTER 6 provides a summary of the key conclusions from the research, and suggests avenues for future investigation. 1-33
1..5
cifications
Spec
Th
he specificattions in Tablle 1‐2 and TTable 1‐3 are based on the expecteed performa
ance of a co
onventional collinear arrray antennaa with conssideration giiven to the lower temperature su
ubstrate of th
he prototype
e and reduceed power ratting. Taable 1‐2 Prop
posed electriical specificaations of the collinear arrray prototypee ELECTTRICAL SPECIIFICATIONS
Frequency band 850‐960
0 MHz HPBW 10.5 de
egree typical
Gain 10 dBi
Side lobe ssuppression typically: ‐10dB
Azimuth paattern stabillity +/‐ 0.8 dB over 360˚˚ 10% bandw
width Tilt 0 degre
ee +/‐ 1 over 10% bandw
width With the o
option to offe
er up to 4° down tilt. Power rating: 100 Waatts Maximum at 26˚ C VSWR: <1.5:1 o
over 10% bandwidth Connector interface: 7/16 Diin jack PIM: ‐150 dB
Bc test two carrier @ +477dBm Peak Instan
ntaneous Po
ower: 1 KW 1--34
Taable 1‐3 Prop
posed mecha
anical specifiications of th
he collinear a
array prototyype The wiind loading ccharacteristiics based on
n array length Projected aarea 1270
1
Lateral thrust @160 km
m/h 160 N 1
Wind gust rating >240 km/h >
Torque @ 160 km/h 73 Nm 7
Length: 1850mm 1
Radome OD: 76mm 7
Mounting: 89 mm diame
8
eter Weight: 2 kg 2
1.5.1
Limiitations There are several com
mmercial lim
mitations for the printed collinear arrray for base
e station ns. The majo
or ones are liisted below: application
1.
The proposed array a
will bee suitable fo
or low powe
er telemetryy and cell extension e
ns where the
e applied pow
wer <100 waatts, based o
on the printeed materials used for application
the feed neetwork and a
antenna elem
ments. 22.
It is restricted in
n diameter too a standard
d off the shellf radome wiith inner diameter of 63 mm. 33.
pacing is gooverned by the t PCB pro
ocessing toleerance, whicch sets a Printed track sp
minimum cconductor trrack spacing of 0.5 mm.
1--35
1.6
Liiterature re
eview
In section
n 1.6.1, discu
ussions on seelf‐choking ffunction of ssleeve dipolees. This funcction, related to
o dipole array antennaas. The nexxt sections 1.6.2 1
to 1.66.8 describe
e the applicatio
ons, the regu
ulations and the details of the three
e different feeed methods for the collinear array an
ntenna. In seection 1.7 the limitation
n related too all base sta
ation antennas.. 1.6.1
Self‐chokingg related to ddipole array antennas Each dipo
ole in an arrray is relatedd to the nexxt through mutual m
coup ling. The mu
utual coupling eeffects can b
be minimisedd by suppre
essing the ind
duced curre nts between
n the elements.. The effectiveness of tthis choking can be see
en in the raadiation pattterns where an improvement is pattern side lobes aare reduced.
The dipolee isolation iss tested by ffeeding one dipole whilstt measuring the level off that signal across the term
minals of anoother dipole
e in the arrayy. Mutual cooupling is alw
ways n an array of dipoles. Its eeffects are re
educed by in
nter elementt choking present in
(“High im
mpedance barriers that acct to cancel ssurface curre
ents betweenn the dipoles”) The use o
of a cylindrical dipole in aan array hass an infamou
us function oof “self‐chokking”. This functtion helps to
o prevents oout of phase
e radiation from f
distortiing the radia
ation pattern. A typical diipole to dipoole isolation
n > 25dB is achievable w
with an elem
ment spacing off 0.81λ as shown in Fig 1 ‐13. 1-36
Fig 1‐13 TThe typical re
esponse for inter elemen
nt isolation
t dipoles iis controlled
d by the inte
ernal λ/4 lonng cylinder cavity The chokee action of the where thee low imped
dance short ccircuited bulkhead is refflected and iinverted to form very high impedance which is preesented at th
he open end
d of the chokke [7], creating a quiet zone. as shown in Fig 1‐14((a). The plaanar dipole also a providess a narrow band w
is sligghtly less efffective than
n the cylinddrical version
n, as self‐chokee function, which shown in Fig 1‐14 (b). This functioon appears in a planar fo
orm as the d ipole arms rreach make their waay on the insside arms off the choke aare cancelled
d out λg/4. Currents that m
dipole as shoown back to back by the higgh impedance presented at the open end of the d
in Fig 1‐15
5 , where the
e impedancee and currentt relationship
p is found. 1-37
(aa) Quiet zone
e (b
b)
Fig 1‐14(aa) Cylinder an
nd (b) planarr choke analo
ogue showing the self‐chhoke function
n Current level
l
CCurrent level after cho
oking
bbefore chokin
ng Equivalent ccircuit
Non‐
λ/4
High Z
Lo
ow Z
λ/4
ng radiatin
High Z
H
Low
wZ
Fig 1‐15 Back to back choke imppedance to odd o mode on two‐elemeent conventional sleeve dip
pole 1-38
1.7
A
Antennas
fo
or base sta
ation applications
Most of the t literature for this tyype of anten
nna is in the form of ppatents from
m the 1980’s wh
hen there wa
as fierce com
mpetition to supply ante
ennas for thee utilities and
d car phones applications in i the VHF aand UHF bands. These patents werre filed by major m
commerciial entities of o that timee. Most havve now specialised in tthe cellular base station arrays. Some have mergedd, and some are no longe
er trading. H
Hence, the public enna literature is scarce and there has been very litttle development of thiss type of ante
ennas manuffactured today are mosttly adaptatioons of the ea
arlier since this time. Ante
designs. nt of The omnidirectional collinear arrray antenna has been the critica l componen
ectional patttern with narrow mobile raadio since the 1930’s [144]. The broad omnidire
elevation beam width
h is most suuitable for mobile m
comm
munications base station
ns. In plications, the azimuth p attern is shaaped to suit specific requuirements of the some app
network [3]. The co
ollinear arrayy antenna finds applications for m ilitary comm
mand one and eme
ergency beaacons, and public posts, airccraft traffic control, marrine sea pho
mobile raadio. Howevver, they aree most widely used for cellular m
mobile teleph
hone services and Wi‐Fi bro
oadband inteernet services. 1.7.1
Antenna spe
ecifications rregulation As this is a performance sells maarket, antenn
nas gain and
d power ratiings are in many m
or example soome manufaacturers claim
m a two‐elem
ment array h
has a cases over stated. Fo
dBd and a four‐element
f
t antenna has h a gain off 6dBd. Thiis is incorrect as gain of 3d
aperture ssize, internal losses suchh as phase d
delay error, V
VSWR and reesistive losse
es all restrict th
he realised ggain. To helpp address th
his problem, a standard TIA329 [15] is in place to regulate th
he industry. It standardises how pattern meeasurementss are performed
d, and prese
ents a standaard gain ante
enna design that can eassily be fabriccated and used
d as a comm
mon referennce for all antenna specifications linked to a a set formulae. At freque
encies abovee 600 MHz, a pyramida
al horn is uused as the gain overing eachh segment of the standard. Below 600 MHz, dual d ipole flat paanel arrays co
omoting their products w
with referencce to band are used. Many companies are now pro
TIA329 gaain standardss and conditiions. 1-39
nother perceeived indicator of superriority in thee industry. These T
Power raatings are an
figures arre usually estimates e
baased on the
e power han
ndling of th e coaxial ca
ables considereed the mostt at risk paart of the design. d
Anttennas withh highly reactive matching circuits are likely to prroduce damaaging standing wave hoot spots resu
ulting melting well bbefore reach
hing the pow
wer limit of thhe cable. dielectric stress and m
1.7.2
Characteristtics of the coollinear arrayy antenna The main objective off the collineaar dipole arraay antenna is to providee an advantage in the form o
of gain over that of a sinngle dipole byy compressin
ng the availaable power in
nto a narrow beeam directed
d into the cooverage area. The total p
power radiatting from a single dipole is ccongruent w
with that of aa high gain aarray. Collinear array anntennas classsified as broadside have patttern maxim um occurring at right angles to the ccentre axis o
of the mnidirection
nal collinear arrays distrribute this energy radiallly. A maxim
mum array. Om
lobe occu
urs at θ = 90
0 degrees w hen each dipole elemen
nt is fed on the same phase p
front (in phase), and has a syymmetricallyy tapered or o equal am
mplitude currrent on applied. distributio
The omnid
directional collinear arraay in its practtical form is an array of ddipoles which
h are positioned
d in a linear pattern mosst commonlyy in the vertiical plane at a spacing off less than a guided wavelength. At thiis spacing, th
he resultant radiation paattern provid
des a m gain at zenith. At largeer spacing’s, extra lobes appear (gratting lobes) w
which maximum
reduce the amount off power availlable at the m
main lobe [13]. Symmetrical aperture
e tapering m
methods such
h as binomial and Dolf‐TTchebyschefff [13] can reducce the side lobe level frrom collineaar arrays to below 20 ddB; howeverr this results in a reduction in directivityy. Specific ap
pplications such as monoo pulse radarr and adaptive beam anten
nnas requiree high side lobe l
suppression to redduce backscatter alize such loow side lob
bes requires additional attenuation and ghosting [16]. To rea
p
shifterrs at each feeed source. An acceptable side lobbe / gain balance variable phase can be acchieved when an array hhas an eleme
ent spacing of 3/4 λ or less. A colliinear array eneergised by a a travelling w
wave, must consider so
ome fundam
mental rules and constraintt, including a a consistentt element sp
pacing of less than a waavelength an
nd an element length l
equal to or beloow λ/2. A non‐complian
n
nt design in most casess will 1-40
u
le performannce and sevvere pattern distortion. Self‐radiatio
on of result in unpredictabl
the feed n
network is a common cauuse for patte
ern distortion
n [3]. The advan
ntage of the
e increased ggain is delive
ering intellig
gible commuunications ovver a wide coveerage area, especially e
w
where repeatter stations are a used. RRepeater stattions rely on high performance antennnas to receivve the weakkest signals and rebroad
dcast these sign
nals to the frringe of the radio horizo
on, which ma
aybe 125 km
m radius from
m the repeater d
depending o
on frequencyy. The repeaaters are norrmally duplexx with frequency splits of up u to 10 MHz. It is m ost important in these systems to cover the same s
geographic area with reception a nd transmission. Most repeaters w
will be locate
ed on bove the co verage area. A series fe
ed array mayy not be able to elevated sites high ab
his level of performance due to beam
m tilt with ch
hange in freqquency, as sh
hown provide th
in Section
n 1.3. 1.7.3
Series fed arrrays The seriees fed array is a very simple desiggn. In mosst cases thee same elem
ment geometryy is repeated throughoutt the array, leading to lower manufaacturing costt. An omnidirecctional seriess fed collineear array sch
hematic show
wn in Fig 1‐ 16. The grad
dient colouring of the arrow
ws representts the curren
nt amplitude taper over tthe length o
of the M
series fed arrays have very high input impedance, as impeda
ances array. Most combine iin series at tthe input. Seeries fed arrrays also havve a progresssive phase e
error, which limits the pattern stability oover frequen
ncy. An
ntenna ele
ements Series fed f
array
Fig 1‐16 Schematic of a series fed array 1-41
ollinear array antenna caan be realise
ed using seve
eral sections of resonant wire A series co
lengths off λ/2 to up to
o 5λ/8 long sseparated byy a phase shifting devicees such as coils or meanderss, as seen in Fig 1‐17. T he meanderring sections of this arra y create a p
phase shift such that the seccond half cyccle of the waave is absorb
bed, leaving only the possitive e [17] whilsst the non‐radiated energy is disssipated as heat. h
half cyclee to radiate
Applicatio
ons for these
e meander ttype collineaar arrays include narrow
w band VHF base station an
ntennas and mobile mouunted antennas for military and utiliities. Analyssis of this type of array ha
as only everr been apprroximated [1
17]; the com
mplexity of such ntenna elem
ments and phasing se
ections are interdependent modellingg is the an
impedancce transformation elemeents. That is, the radiation function aand the matcching functions are insepara
able. Raddiating
Noon-radiating on
o first cyclee
Meander type
e collinear a rray Fig 1‐17 M
A series feed array can also be real ized using se
ections of coaxial cable innterconnected in transposittion as show
wn in Fig 1‐118 [18]. Thiis array funcctions by rouuting the applied energy inside the coa
axial cable nnon‐radiatingg sections in each half ccycle, leavingg the s
to radiate on the positive
e half cycle. This ensurees that onlyy the exposed sections positive p
part of the cyycle is able too radiate, ressulting in a ssingle major lobe and sm
maller side lobess. There is extensive lite rature on th
he numerical analysis of tthis type of a
array antenna [19]. The feed point imppedance for this antenna
a increases w
with the add
dition he design is ssuitable for n
narrow band
d commerciaal products in
n the of more eelements. Th
UHF 400 M
MHz band. 1-42
Radiaating
Non- radiating
r
Fig 1‐18 TTransposed cable antennaa array Another coaxial c
cable
e based anteenna known as a suppre
ession collinnear array [2
20] is shown in Fig 1‐19. The T suppresssion collinear array cre
eates an alteernating radiator element/ssuppressor structure consisting of resonant/low impeedance radiator sections, aand non‐reso
onant/high i mpedance suppressors a
at the designn frequency. This array feed point imp
pedance is high (typicaally > 300 Ω) Ω and requuires a com
mplex h as dual stuub tuning att the input. As the imppedance is highly matching circuit such
ound reactive, tthese antennas have a vvery narrow impedance bandwidth, typically aro
3% [21]. The antenna
a finds appliccation as a n
narrow band
d base statio n antenna in
n the 400‐1000 MHz band. Non-radiating
Radiating
Fig 1‐19 Series ffed suppresssor collinear
he most com
mmon series fed antennas is the annular slot feed coaxial sleeve One of th
dipole arrray, also kno
own as COC O (“Coaxial Collinear”), as depictedd in Fig 1‐20. It comprisess of a numbe
er of weldedd dipole assemblies moun
nted one aboove the othe
er. A coaxial caable with outter shield re moved is insserted, form
ming an arrayy of annular slots between the dipole bulkheads oof each asse
embly which
h feed each adjacent sleeve he elementss are field isoolated from one anotherr by the integgral λ/4 choke in dipole. Th
the dipolee sleeve. Ass the feed linnes between
n dipole elem
ments are cooaxial cabless, the 1-43
element spacing is governed by the velocity of propagation as calculated in equation (1.2) [22]. This propagation delay determines the electrical length of the cable used in the antenna impedance matching circuits as well as the element spacing. A spacing of 0.7λ is ideal for low side lobes [13]. The metal element components provide a high degree of thermal dissipation making this design suitable for higher power applications. As the element diameter is much greater than that of the coaxial cable, this antenna can provide a broadband VSWR response. The feed point impedance is also lower than previous examples mentioned above. √
where: c = velocity of light = 3 ×108 m/s μ = permeability of dielectric ε = permittivity of dielectric 1-44
(1.2) Dip
pole bulkhea d Dielectric λ/4 Coaxial Annularr slot feed Fig 1‐20 Series fed ann
nular slot fedd collinear arrray: section view The use of prin
nted circuitss for collineear array allows for mo
ore degrees of freedom
m for e for impeedance mattching, as exxtra open ccircuit stubs can be inttegrated to compensate
exceessive reacttance [23]. The basicc structure shown in Fig 1‐21 iss a typical PCB reprresentation of a coaxial dipole typee collinear array a
using microstrip m
trransmission lines [24] [25]. An exxample of a ccommerciallyy available series fed arrray antenna iis the COL85
5‐870 na is a mean
nder phase s hifted array of five radia
ating elemennts on a low cost [9]. This antenn
orm radiating elements aand the meander flexiible substratte. The substrate carries tthe planar fo
elem
ments, manu
ufactured witth the precisse positioning offered by PCB technoology. The pllanar anteenna elemen
nts are attached to thee choke element, which
h also housees the matcching circu
uit. The anttennas are re
elatively lighhtweight, bro
oadband (with VSWR < 11.5:1 over an 8% band
dwidth), and
d have a gain
n of 6 dBd. The antennaa is specifically designedd to minimise
e the geneeration of PIM typically < ‐150 dBc.. In Fig 1‐22
2 a typical application foor this seriess fed array is shown, ggenerating a bidirectionaal pattern. 1-45
Upper layer copper 0.75λ Lower layer copper Dielectric layer Fig 1‐21 Planar double‐sided PCB series fed array [24] Fig 1‐22 Series fed arrays configured for bidirectional coverage 1-46
1.7.4
Centre fed a
array antennna The canceellation of beam b
tilt inccurred by se
eries fed arra
ays is the m
main objectivve of implemen
nting a centrre fed array. A centre fe
ed array is fundamental ly two seriess fed arrays mo
ounted backk to back aand fed in phase, p
producing a diffferential current distributio
on in each series sub arrray. This arrrangement effectively e
caancels the beam b
tilt with change c
in fre
equency. Tiltt occurs in both b
main beams simulttaneously, but in the oppossing direction
n, forming a “gable phasse profile” [1
13] as shownn in Fig 1‐23. The main beam of the ce
entre fed arrray now remains stable with channge in frequency rather thaan abruptly tilting. Uppper sub arraay
+ =
Lowest
+
L
Lower sub arrray
=
Complete arrray
Centre frrequency
+
=
Higghest frequen
ncy
Fig 1‐23 P
Pattern beha
aviour of an oomnidirectio
onal centre fe
ed dipole arrray The majo
ority of convventional cenntre fed dip
pole arrays for f base stattion applicattions have a paattern bandw
width of 8% ffor six radiating elements. As the caables in a ce
entre fed array are in the sa
ame plane ass the dipoless, they are likely to be soource of spurious 1-47
radiation unless isolated from the radiating surfaces. There should also be good isolation between each sub array. If not properly isolated, out of phase energy radiating from the feed or the mounting tube can lead to the distortion of the radiation pattern. To satisfy the requirement of feed isolation, array design is often restricted to using the more expensive coaxial sleeve dipole with a hollow core [7]. The most complex part of this array is the power divider and impedance matching circuit. Impedance matching for the array must be located as close as possible to the feed point, in order to provide balanced broadband matching and minimize the phase errors. If the feed point is at the centre of the array with differential current distribution to the sub arrays, the amplitude remains constructive over a larger number of elements. Mutual coupling and electrical spacing’s exceeding λg limit the bandwidth of centre fed arrays. Outside of the operating band the end lobe levels significantly increase and the main lobe starts to tilt, as out of phase currents become dominant. Commonly, two routing configurations are used for centre fed arrays. In the case of the unidirectional array in Fig 1‐24, the series arrays are fed from the side. For the omnidirectional case in Fig 1‐25, the feed is routed inside the shielding. Coaxial choke elements at the base of the array prevent coupling back to the tower and feed cable. The Marconi‐Franklin array [26] as shown in Fig 1‐26 was one of the first collinear array antennas, which was re‐designed to be centre fed array [27]. The antenna is formed by partially stripping the braid off a coaxial cable, forming a coaxial dipole. By bending periodic meanders in the line the current distribution truncates and forms resonant and non‐resonant elements over the length of the array at the design frequency. The array relies on cable surface currents for the excitation. 1-48
Series fed
d sub arrays
Feed nettwork shield Fig 1‐24 Schematic of unidirectionnal centre fed
d array with a side feed λ //4 current ch
hoke Series fed sub arrays
Feed netw
work shield Fig 1‐25 Schematic of an omnidireectional centre fed array with a routeed feed 1-49
Radiating
sections
λ/4 currentt choke
Coax
xial section
Fig 1‐26 TThe Marconi‐ Franklin anntenna [27]
Another form of centrre fed array is the suppre
essor colline
ear antenna aas shown in [20]. Although this is a very early dessign (1955), it is still the
e concept ussed for man
ny of ducts Fig 1‐2
27. Its operaation is similar to today’s ceentre fed basse station anntennas prod
the seriess fed suppresssor collinea r antenna, e
except it take
es advantagee of the stan
nding waves gen
nerated either side of thee feed point. R
Radiating sections
Coaxiaal choke
Non‐radiating
N
g
Fig 1‐27 C
Centre fed suppressor colllinear anten
nna Many current antennas are basedd on these e
earlier welde
ed metal tubbe designs, w
which o meet the more stringgent base sta
ation are at higgh risk of causing PIM aand failing to
specifications. A typ
pical commeercially prod
duced centre
e fed array with adjusttable 1-50
beam tilt is shown in Fig 1‐28 [7], [28]. The cylindrical dipoles sleeves are welded to a series of support tubes and an air dielectric coaxial transmission line runs to the centre feed point. This antenna is an adaptation of the annular slot fed sleeve dipole array described in Fig 1‐20. In the centre fed version, the coaxial cable feed is isolated before reaching the mid‐point in the array. A tapping point from the inner conductor through to an insulated capacitive cylinder is coupled to the outer conductor of the secondary coaxial feed line. The inner conductor of the main feed line is then shunt fed onto the upper inner tube. The sub arrays are fed back to back; hence the phase relationship is 180˚ as required for differential phase distribution [13]. A shorting ring placed λ/4 from the feed point shunts the central dipole assembly tube back onto the outer of the main feed line in order to reduce the feed point impedance and provide static protection. 1-51
Sleeve dipoles
Imped
dance match
hing Shorrting ring Capacitive coupling rin
ng
w of centre feed collinear aarray antenn
na [7]
Fig 1‐28 SSection view
1.7.5
Corporate fe
ed collinear arrays Corporatee fed antennas selectted for ap
pplications such s
as M
MDS (Multi‐p
point distributio
on) Wi‐Fi [3] have speciffic pattern shaping. Som
me installatioons require a bi‐
directionaal radiation p
pattern, for example to cover narrow corridors such as tunnels, subways and roads [29]. Otherss require se
ectored coverage such as in a cellular network. orate In Fig 1‐29 a schemattic for a typiical corporatte fed array is presentedd. The corpo
power divide
ers which proovide a cascaded fed array feed networrk consists oof two‐way p
etwork. To be classed as corporate fed the feeed line secttions binary disstribution ne
must provvide the sam
me electrical phase length
h from the in
nput source to each antenna element rregardless off the mechannical position of the elem
ments in thee array [13]. This may lead to very long feed runs, eespecially wh
here the cable enters at tthe base. 1-52
S
Shielding
from radiating ssurfaces
Power divid
ders Fig 1‐29 Schematic of a corporate fed array Corporatee fed arrays are commoonly configu
ured as unid
directional inntegrated pa
anels [30]. This type of antenna arrray can app
proximate omnidirection
o
nal coverage by sectorial aarrangementt, shown in Fig 1‐30. A synthesised omnidirectiional coverage is achieved if three secctor arrays aare aligned such s
that th
he ‐10dB beaam width points d the arrays fed in phasee, as shown in Fig 1‐31. merge and
Fig 1‐30 A
A cluster of 1
120° panel arrray antennaas 1-53
33 x 120°
sspaced ppanel array
aantennas angement off corporate ffed panel arrrays in a celluular networkk Fig 1‐31 TTri‐sector arra
antenna elem
ments positio
oned An omnidirectional exxposed foldeed dipole arraay with the a
V
concentriccally [9] is shown in Fig 1‐32. This array provides brroadband VSWR performance (typically 26%) withh narrow elevation patte
erns and a n ominal gain of 6 dBd. Thee feed network for this array is com
mplex, consisting of an internal harrness made from
m low loss co
oaxial cable. Fig 1‐32 C
Corporate fed folded dip ole array (RFFI BA80‐67) [[2] One of the latest deve
elopments inn omnidirecttional corporate fed arraays is the CC
C807‐
pole radiatorrs are uniquee in that theyy are 11 series array [2] shown in Fig 1 ‐33. The dip
w
dipoless wrapped around meta
al forms creaating broadband PCB sheets of half wave ht cylindrical elements. The feed caable harness is routed innside the central lightweigh
support m
mast to preve
ent detuningg of radiatingg elements, a
and has an ooperating ban
nd of 746‐870 M
MHz. This array is desiggned for high
h power and low PIM, w
with a typical PIM value of << ‐150 dBc. 1-54
Fig 1‐3
33 Corporatte fed printedd sleeve dipo
ole array [2]
wn in Fig 1‐334 [31]. The feed Another fform of corporate fed sleeeve dipole aarray is show
network is i created ussing a custoom aluminium
m extrusion with indepeendent shaffts to form the air dielectrric multi‐tra nsmission line feed nettwork. A W
Wilkinson po
ower e corporate feed lines, and divider ussed to split the power equally to each of the
improve the t current balance leaading to low
wer side lobe
es and cleann patterns. The antenna has h relatively large diam
meter dipole
es, which incclude a matcching stub. The stub is in parallel with
h the feed ppoint, formed
d by folding back the grrounded elem
ment he centre of each dipolee. The who
ole assemblyy inserted innto a heavy‐‐duty end to th
radome and exhibits a
a bandwidth of 18%. Dipole end folded back forming a high impedance feed pooint Air dielecttric feed liness are A section off the feed tube with
part of the
e aluminium coaxial channnels Fig 1‐34 Corporrate fed sleevve dipole arrray with custom feed nettwork [31] 1-55
orate fed arrrays can prrovide beam
m tilt by usin
ng a delay oor line stretccher, The corpo
integrated
d in either the power diivider arms or as a phasse offset in tthe transmisssion line sectio
ons. A phasse shift distr ibuted at alll branches maintains m
low
w side lobess and impedancce matching in the tiltedd pattern. Another meth
hod is to useed phase shiifters and atten
nuators as bu
ulk componeents [32]. The phase shifters control the phase o
offset and the attenuators le
evel the ampplitude independent of tthe phase. TThis levellingg has the effectt of reducing the side lobbe levels. [16]. Although corporate fe
ed arrays cann be broad band, they lacck freedom iin designing for a particularr gain. Most noticeable i s extending from eight e
elements to ssixteen elem
ments in order to t increase gain from 111 dBi to 13
3 dBi respectively. The doubling off the aperture leads to exce
essive lossess in the ante
enna system due to the eextensive co
oaxial cable harn
ness. Printed circuit bassed array anttennas are n
now more w
widely use in base station ap
pplications [3]. The m
main advantaage comes from the h ighly repeattable geometryy and lower labour costs involved. Th
he physicallyy smaller sizee of compon
nents at higher frequenciess makes pri nted circuitss an obviou
us choice, ass with incre
eased onents need precise tole
erances thatt can renderr hand assembly frequencyy the compo
ineffectivee. 1.7.6
nidirectional arrays Planar omn
Available literature suggests thatt planar cen
ntre fed om
mnidirectionaal collinear array a
antennas do not app
pear to be aas popular as a series fed
d, perhaps ddue to the more m
ments. Som
me non‐coplaanar (two layyer printed ccircuit) centre
e fed complex ffeed requirem
arrays wh
here a coaxia
al cable is inncluded from
m the base o
of the array up to the ce
entre are moree prevalent. This exampple is prese
ented as a triple t
band [33]. So‐called coplanar centre fed collinear c
arraays of two elements e
ha
ave also beeen reported [34]. ains non‐cop lanar compo
onents that w
would need tto be hand ffitted This desiggn also conta
during asssembly, like
ely incurringg a significaant labour cost. c
“Copllanar wavegguide (“Coplanaar waveguide
e (CPW)”)” ffeeding geom
metries commonly requiire air bridge
es to equalise gground poten
ntial. These thin conducctors bridge tthe ground ttracks in order to cancel ou
ut higher ord
der modes aat discontinuities such as bends annd transition
ns to connectorrs [35]. 1-56
The anten
nnas in [34] are a logicaal embodiment for a tw
wo‐element centre fed array a
based on
n coplanar wave‐guide . However, issues asssociated w
with the currrent distributio
on and ampllitude were raised but n
not resolved. Also, the direct air brridge connectio
on from the inner conducctor of the “C
Coplanar waveguide (CPW
W)” to the active conductorr of the seccond dipole elements ass shown in Fig 1‐35 do es not allow
w for further exxpansion of tthe array dessign to more elements. Air bridges Air brid
dge to inner
conducttor
Fig 1‐35 ““Coplanar wa
aveguide (CPPW)” based p
planar centre
e fed array [334] A four eleement series fed uniplanaar collinear aarray is described in [36]], and is depicted in Fig 1‐36. The array uses elect romagneticaally coupled dipole elem
ments througgh an aperture in the groun
nd tracks. Thhis feed comprises of a ““Coplanar waaveguide (CP
PW)” h periodicall y slotted fin
nite ground tracks t
to feeed the dipoles in transmission line with
he distance b
between slotts appears to
o be at λg inttervals. The measured V
VSWR series. Th
bandwidth is quite na
arrow at 2%,, and impedaance matching is achieveed using a single Coplanar waveguide (CPPW)” feed line. series traansformer integrated i nto the “C
Expansion
n to more array elemennts is possib
ble due to passive coup led architecture. The anten
nna array was w fabricateed on a 200
0 micron thiick Rogers RR4003 substtrate, providing a low cost array implem
mentation forr the 10 GHz band. 1-57
Ground trrack slots feeed the Integrateed Series passive dipole strips
transform
mer Fig 1‐36 C
CPW based p
planar series fed array [36] 1.8
D
Design
limitations in a
all base station antennas
In order to t support subscribers s
w
with a high quality and reliable nettwork, the la
atest technologgical advance
es and systeem implementations succh as “Time division multiple access (TD
DMA)”, “code division m ultiple accesss (CDMA)” a
and “wide baand code divvision multiple access (WC
CDMA)” aree offered by telecomm
munication ccarriers, ma
aking nd frequencyy reuse [3]. The economiccal use of broad bandwiddths, spatiall diversity an
base station antenna
as deployed in these systems must not hinder the exceptional w technologiies. performance offered by these new
echniques, a a number oof challengess for With the introduction of these advanced te
d
arise. A highlly concentrated spectrum leads to nnew phenom
mena antenna designers such as PIM (Passive intermodulaation) and PIP (Peak instantaneous ppower). Man
ny of nt base station antenna designs have not consid
dered these pphenomena,, and the curren
as such may m not meet these neew specificattions. An antenna depployed in cellular networks must also b
be able to suustain relentless mechanical stressess of wind loa
ading and vibraation, and be impervioous to moisture. The combinattion of multiple transmitteers at a single base statiion site resu
ults in high peak powers.. Antennas used in these highly h
conce
entrated rad io sites are heavily scrutinized for cconstruction
n and power handling. ower A growingg trend in network deliveery is taking shape, wherre closer spacced lower po
base statiion sites are
e deployed [[3]. This is driven by en
nvironmentaal factors, ass the lower pow
wered sites w
will consumee less energy. Lower po
ower sites arre also less llikely 1-58
to generate PIM and PIP. This presents an ideal situation for printed circuit based antennas as high volume, lower power rated antennas at a fraction of the cost of the conventional base station arrays. There are numerous flexible substrates that may be employed into printed base station arrays. PET is low cost and is readily available substrate from PCB processing companies in Asia. The material is Bo PET (“Biaxial‐oriented polyethylene terephthalate (Bo PET)” has been available in several forms for some time. Originally by DuPont, development of advanced materials based on PET starting in the 1950s; NASA first used it in 1960 for the Echo 1 satellite project where it found applications in the materials used for manufacture of the inflated reflective sphere. These flexible PET substrates originated from the research into alternative low cost flexible PCB substrates and materials. It is a lower cost alternative to Kapton and Polyamide for flexible interconnects, although Kapton and Polyimide can operate at much higher temperature than PET [37]. PET is now widely uses in (“radio frequency identification (RFID)” tags and labels and in multi‐layer, low profile PCBs found in current electronic equipment. Despite its low cost, its RF properties of 125 micron thick, 35 micron copper clad PET are very good, with a permittivity εr = 2.8 – 3.0 and loss tangent of 0.0013 making it suitable for high frequency applications. Prior to PET substrates, the production of flexible substrates in lengths greater than 500 mm would have proven too costly for antenna array applications. There are only a small number of PCB manufacturers capable of processing these longer PET flexible substrates. New techniques have significantly improved the registration accuracy in processing long PCB’s. This was motivated by the need to commercially produce a longer than standard length PCB’s for other antenna products. Six‐meter long elements designed for some VHF high gain series fed collinear arrays antennas [9]. Another substrate that is becoming more available is LCP (liquid crystal polymer) [38] . It has better thermal properties than PET, but is more expensive and minimum order quantities are required to produce it economically. RFID tag applications may increase demand for LCP, making it a more cost effective proposition in the future. 1-59
Shown below in Table 1‐4 are typical specifications for base station antennas available to industry. Most of these designs are based on the earlier developments in antenna design. Table 1‐4 typical base station antenna specifications 1-60
C
Chapter 2
anar waveg
guide (CPW
W)” feed nettwork with centre fee
ed
“Copla
Trraditionally, centre fed array netwoorks consist of a number of ridgedd coaxial conductors arrranged to provide a feed to coaxiallyy mounted ccylindrical metal dipoles [7]. In this cchapter a no
ovel planar feed arch
hitecture is introduced, with perfformance eequivalent to t these co
ommercial co
oaxial system
ms in terms oof achieved b
bandwidth, p
phase stabilitty and system
m losses. Th
he planar feeed networkk has the addded advanttages of rela
atively low ffabrication cost, c
and ad
daptability to t support beam b
tilt. TThis feed ne
etwork will also be moore repeatab
ble when m
manufactured
d in large batches b
due to maturitty of printed circuit tecchnology. With no so
oldered conn
nections in th
he array, passsive intermo
odulation [10
0] will be siggnificantly low
wer than co
oaxial implem
mentations. As the feed network is fflexible, it is less likely too be compromised by m
mechanically induced vib
bration. An innovative “Coplanar waveguide w
((CPW)” disco
ontinuity similar to the geometry o
of [39] is useed to distribute symmetrical linear aamplitude an
nd phase prrofiles along printed slotlines to centtre feed the ccollinear anttenna array eelements. Th
he performaance of the array is num
merically characterized and validateed using com
mmercial ellectromagneetic solver CSST Microwavee studio [40]]. 2..1
Struc
cture of the
e planar ce
entre feed network
Th
his section overviews o
th
he structure and operatiion of the planar centree feed netwo
ork. The im
mplementatio
on of a plana
ar structuredd feed netwo
ork for a centre fed collinnear array (a
as seen in Fig 2‐1) uses aa finite ground “Coplanaar waveguide
e (CPW)” as the primaryy feeder. The
e second n
is a a symmetricaal set of slotlines, which
h feed the aarray elemen
nts. The layer of the network mplitude must have crritical requirrements for this type oof feeding syystem are phase and am
m
mirror symmeetry through
h the centre axis of the aarray. This iss known as aa linear phasse profile [113]. Even tho
ough the phase delay off the individual sub arrays changes w
with frequency as in seeries fed arraay, the mirro
or symmetry of the portss have an opp
posing phasee error which
h cancels m
mutual phase shift effectss in the arrayy. In the pro
oposed netw
work, the exciitation of the
e array is prrovided by aa novel “Coplanar wavegguide (CPW)”” discontinuiity [41]. Thiss simple yet effective circuit in conjunction with
h the secondd layer of slot lines create
es a series pparallel powe
er divider w
with balanced
d ports. Effecctively the arrray networkk consists off two dual poort series fee
ed arrays baack‐to‐back. 2--61
Fig 2‐1 The baasic architecture of the pplanar array ffeed networrk Th
he initial secction of the ffeed networkk shown in FFig 2‐2 utilise
es an input ““Coplanar wa
aveguide (C
CPW)” line th
hat extends tto the centree of the array. The slotline transmisssion lines are
e formed in
n close proxim
mity to the gground trackks of the “Coplanar wave
eguide (CPW))”, a novel te
echnique ussed to redu
uce the overall width oof the feed network when comparred to othe
er planar sttructured arrrays [36], [34
4]. Co
onventional “Coplanar w
waveguide (CCPW)” ground tracks are typically douuble the width of the so
ource tracks (or larger) b
based on thiss paper [35] [42]. The pro
oposed feed network has a more co
ompact finitee ground “C
Coplanar wavveguide (CPW
W)” transmission line. TThis transmisssion line w
will require evaluation, e
as a it no lonnger conform
ms to the standard s
cloosed form equations e
avvailable for cconventional “Coplanar waveguide (CPW)”. A sshort circuit slotline stub
b is used affter the final antenna ports to ensuree appropriate phasing of the antennaa elements. CPWinp
put
Antenn
na port Stub
Anteenna port
CPW
C
source
Slot liine
Antenna port
p
Antenna port
p
Fig 2‐2 CPW input with slo
ot lines feed ing the antenna ports 2--62
2..1.1
The feed netwo
ork componeent impedance control TThe impedance of “coplanar waveguiide (CPW)” ttransmission line is primaarily set by the width off the sourcee track and the spacingg between the ground and a the souurce track. For F finite grround “Coplaanar wavegu
uide (CPW)”, the ground track width also plays a role. Narrow
wer gaps an
nd wider sou
urce track ressults in a low
wer impedance line. Th
he impedance of a slottline is contrrolled by th
he gap spacing. The narrrower gap between co
onductors reesults in low
wer impeda nce. In the
e case of th
he proposedd feed netw
work, the an
ntenna ports are equal in impedannce to the fe
eeding slotlines to ensuure maximum
m power trransfer. A series transformer is ussed to transfform from th
he main line impedance to the seriess parallel co
ombination feed f
point im
mpedance oof approximaately 127 Ω. In the prooposed netw
work, the seeries impedaance matching transform
mer is creatted by expanding the ““Coplanar wa
aveguide (C
CPW)” ground track spacing, creatingg a section off relatively h
higher impeddance as shown in Fig 2‐‐3. The traansformer iss used to m
match the slotline s
discontinuity juunction to the main “C
Coplanar waaveguide (CPW)” line. The “Copllanar waveg
guide (CPW))” transmisssion line co
ontinues passt the slot disscontinuity aand terminattes in a shuntt stub to a viirtual ground
d. Imp
pedance traansformer
Sllot discontinu
uity
CPW term
mination
Fig 2‐3 The ceentre feed po
oint comprissing of a CPW
W with seriess impedance transformerr Th
he slotline feeds comprise of the “CCoplanar waaveguide (CP
PW)” groundd track and an extra trrack runningg in parallel which alsoo electrically supports the slotline ooutput portts to the diipoles. The sslotlines routte to the exttreme ends of the array as shown inn Fig 2‐4, wh
here they evventually retturn to virtua
al ground viaa a shorted stub. 2--63
Stub
Sloot line
nd of the fee
ed network sshowing the slot lines and
d the return to virtual ground Fig 2‐4 The en
2..1.2
Feed
d network co
omponent pphase contro
ol Th
his feed network is base
ed on the w
well‐establish
hed theory related r
to p hased arrays; where eaach port is inter conn
nected usin g λg delayy lines. The “Coplanar waveguide (CPW)” trransmission line is term
minated in aa λ/2 short circuit stub
b (“Coplanarr waveguide
e (CPW)” teermination aas seen in Figg 2‐3). The λ/2 stub is p
primarily use
ed to transfoorm the sho
ort circuit baack to the discontinuity
d
nce for the upper arrayy. Short , thereby prroviding a phase referen
circuit stubs aare also placed at the farr end of the slot line transmission linnes to termiinate the slot line to virtual ground. 2..1.3
Feed
d network cu
urrent distri bution Th
he finite gro
ound coplanar transmisssion line in the propose
ed configuraation has qu
uasi TEM (““Transverse electromagn
netic”) [43] mode propaagation until its arrival at the grou
und track diiscontinuities. This behavviour is evideent from anaalysis of the ssurface curreent vectors a
as shown in
n Fig 2‐5(a) the t symmettrical currentt vectors are seen on a a section of the main “Coplanar w
waveguide (CPW)” transm
mission line. A section of the feed network is prresented in FFig 2‐5(b) co
ontaining bo
oth the “Cop
planar waveeguide (CPW
W)” and the pair of parrallel slotline
es shows similar curren
nt vectors on
n the “Coplannar waveguid
de (CPW)” line, and oppposing vectorrs for the slotlines on eiither side. At the slot ddiscontinuityy in Fig 2‐5(c)), the curren t is transverse to the nge at this po
oint where t he energy iss coupled trransmission lline, indicating a distinctt mode chan
to
o the slot linees. 2--64
(b)
(a) (c)
Fig 2‐5 Surfacee current vecctors n CPW (b) CP
PW with a paair of slotline
es (c) the CPW
W discontinuuity (a) the main
2..1.4
Relaationship to a coaxially ffed architectture he proposed
d planar fee
ed network has a similar current distributionn and anten
nna port Th
co
oupling to th
he commonlyy implementted coaxial co
ounterparts.. The cross ssection of the coaxial feeed network in [7] is veryy similar to thhe planar fee
ed network. Co
oaxially consstructed networks are fuully enclosed
d in the bodyy of the arraay they feed; as such prrevent the feeed networkk cabling from
m de‐tuning the radiating sections. TThe propose
ed planar feeed network takes advan
ntage of “Cooplanar wave
eguide (CPW)” high crosss talk [35] to
o provide a similar leveel of protecction. A maajor advantaage over co
oaxial architeectures is the open co
onductor sch
heme allowin
ng for easy i nspection du
uring assembly. The plaanar structurre can be reealized on a single sided,, flexible, low
w cost printe
ed circuit ma
aterial. This added flexib
bility also reeduces the incidence of o stress frractures fou
und with co
onventional solid tube welded co
omponents. 2--65
2.2
Requirements and limitations for the planar centre feed network
This section highlights the important requirements for a planar centre feed network used in a collinear array, and the limitations imposed by the structural configuration. The substrate for the planar centre feed network is selected based on electrical performance and low cost. An extremely thin substrate is effective at reducing transmission losses. However as the substrate becomes thinner, a 50 Ω characteristic impedance “Coplanar waveguide (CPW)” normally used for a feed network would require a very close ground spacing or an exceptionally wide source track. A thicker substrate enables lower impedances to be achieved more readily; however it would introduce phase delay that reduces antenna element spacing and lowers the directivity. Using a wider “Coplanar waveguide (CPW)” source track width is not an option in the proposed array due to the structural and mechanical constraints that limit the width of the array. The final configuration of the collinear array formed with this feed network will utilize a radome with an internal diameter of 63 mm. This limits the track widths used within the design of the feed network, as the slotlines and antenna elements of the collinear array also need to fit within this diameter. A very small ground gap is also undesirable due to the PCB manufacturing process. As the printed feed network is 1850 mm long, a step and repeat fabrication process is employed. The tolerance of this process is 0.5 mm [37]; making very small gap spacing’s unrealisable. The planar feed network must be sufficiently supported to prevent fracture in the copper cladding that may occur under severe vibration. The prototype array has two “D” shaped foam mouldings that fit either side of the PCB sandwiching it concentrically within the radome, as explained in Section 5.1.2. The foam material used is high density polyethylene, which was tested for RF suitability and was found to be inert. Due to the cellular consistency, the effective dielectric constant of this material is εr = 1.04, which is close to that of air. The planar nature of “Coplanar waveguide (CPW)” does not offer the same level of thermal dissipation as metal coaxial components. This will impact on the power rating that can be specified for this antenna. The substrate selected for the printed feed network is a relatively low cost Polyester (PET) which is 0.125 mm thick, with 0.035 mm of copper cladding. This substrate material is 2-66
commonly used for other base station antenna products [9]. The material specifications: εr of 2.8 and loss tangent of 0.0013 at 1 GHz. The film substrate can withstand a temperature of 150°C, which is sufficient for the specified maximum power of 100 Watts as demonstrated in a live power test in chapter 5.6.2. 2.3
Feed network design for a planar collinear array
The calculations for the feed network are for fundamental line transformers with substrate loading. The dimensions for the transmission line track widths and slot lines have been initially calculated using conventional transmission line formulae (found in any basic transmission line text – and hence were omitted from the thesis), and these baseline values were used as a starting point for the EM model using CST Microwave studio and CST design studio [40]. As there are areas of the feed network where some cross coupling is evident, no conventional formula can accurately calculate the line widths. Optimisation with a goal of a specific impedance value has been an effective way to determine these parameters. The line lengths are governed by the radiating element spacing, and the series transformer lengths. The design of the planar feed network for a centre fed collinear array starts with the characterisation of the main feed line geometry. Then the realisation of the transmission and discontinuity components of the feed network can take place. The use of electromagnetic simulation software is essential to design an open architecture feed network of this type. Extensive parameter sweeps and optimization has been used to develop these interactive transmission lines. 2.3.1
Theoretical design of the feed network The highest realized gain for a collinear array occurs with very close spacing between the antenna elements on opposing sides of the array. Implementing the planar feed structure describe in Section 2.1 would require smaller transmission line gaps than what is allowed for in step and repeat PCB processing. To realize a 50 Ω “Coplanar waveguide (CPW)” for the main feed line requires a source to ground track spacing of 0.125 mm. The PCB processing limitation has a minimum track spacing of 0.5mm [37]. 2-67
anar waveguuide (CPW)” to allow n impedancee of 85Ω is sselected for the finite grround “Copla
An
fo
or wider physical track sp
pacing as disscussed in Se
ection 2.2. T
To find the ooptimum geo
ometry, a paarametric sim
mulation is performed uusing [40]. The variable
es for this siimulation arre source trrack width, and ground track gap spaacing with ground track w
width fixed. At this impe
edance, a trrack spacing of 0.638 mm
m is possible. he secondary slotline feeds are alsoo set to a minimum gap size of 0.5 m
mm. This ressults in a Th
lin
ne impedancce of 127Ω. An advantaage of this w
wider track sspacing (highher impedan
nce) is an in
ncrease in thee break overr voltage relaated to PIP (P
Peak Instantaneous Pow
wer) [10], as d
discussed in
n Section 1.2.3. Th
he design off the feed network n
hass to preserve
e a symmettrical distribuution of the
e applied en
nergy. Each of the line lengths are pphase match
hed to ensurre the antennna ports recceive the saame phase signal. An am
mplitude tapper is expected as the four sub arrayys are series fed, and ass such energgy is lost at each conseecutive anten
nna port. Each E
antennaa port mustt present baalanced portt impedance
es, the enerrgy is balancced in each sub array tto ensure a uniform diistribution. Line length
hs are seleccted which are a related to a waveleength at the design frequency of 903 MHz. A
An impedancce schematicc of the feed
d network is shown in Fiig 2‐7. It ndicates the impedance and line leength of each branch of o the feed network to
o ensure in
m
maximum effiiciency. Teermination sttubs: 127Ω sloot line, lengtth λg
Slot line S
Im
mpedance 12
27Ω; length U
Upper section
n Low
wer section 85 Ω CPW
Slot lin
ne Typical Terminatio
on stubs: Port Impedancce 127Ω; mpedance im
length 0.0
053λ.
129Ω 2--68
0.0053λ. ne and termin
nal impedannces of the pllanar centre feed networrk Fig 2‐6 The lin
he 13 port feed f
networrk (one inpuut and 12 ou
utputs) consists of twoo sets of parrallel sub Th
arrrays in mirro
or symmetryy as shown inn Fig 2‐7, eacch sub array has 3 antennna ports represented ass a complex impedance block with aa real compo
onent of approximately 1129 Ω. The p
ports are lin
nked in seriees using line sections L1 with a lengtth of λg, and line sectionn L2 with len
ngth λg/2. Bo
oth L1 and L2 have an im
mpedance of 127Ω, as do
oes L3 which is a return ppath to virtua
al ground fo
or each sub array. L4 provides p
a rreturn path to virtual ground g
for the main “Coplanar w
waveguide (CPW)” line. T
The networkk is considerred as a serie
es resistive ccircuit with e
each sub arrray presenting an impedance of ≈ 3380Ω at 903
3 MHz, when
n the four suub array imp
pedances arre combined
d in parallel and the ““Coplanar waveguide w
(C
CPW)” discoontinuity im
mpedance co
omponent iss considered,, the result i s an impedaance of 120Ω
Ω. Due to mi nimum trackk spacing fo
or PCB fabriccation, the m
main feed im
mpedance has been raised to 85Ω wiith a track sp
pacing of 0..65mm (L5), determined using the im
mpedance calculation forrmula for finiite ground “Coplanar w
waveguide (CPW)” given in [35]. A seeries transformer of 100 Ω (line L6) is used to trransform from this to th
he feed line o
of 85 Ω (L2). A further λλg/4 transform
mation of 755 Ω (L7) is req
quired to Ω input impe
edance of thhe passive co
oupler circuit as describeed in Chapte
er 4. The reeach the 70 Ω
paassive coupleer circuit is d
designed to ttransform fro
om the 70 Ω
Ω “Coplanar w
waveguide (C
CPW)” to a 50 Ω microstrip where the feed netw
work can be connected to
o the base sttation equipment. L3
L1
L1
L1
L1
L1
L4
L2
L2
L2
L2
L3
L1
L1
L1
L3
L3
Anten
nna L5
L6 L7
7 Po
ort 1 Sourcce
Virtuaal
ed network sschematic Fig 2‐7 the theoretical fee
2--69
del of the co
omplete plannar feed network based
d on the schhematic in Fig F 2‐7 is A circuit mod
nalysed with
h respect to phase and aamplitude att each of the
e antenna poorts. The re
esults are an
prrovided in th
he following section. 2..3.2
Circuit model re
esults Th
he input reflection coeffiicient magnittude of the ccircuit mode
el schematic in Fig 2‐7 is sshown in Fig 2‐8. The rresult is welll inside the 114 dB reflecttion coefficie
ent magnitudde specificattion over he region of interest from
m 850 – 960 MHz. The ttransmission loss charactteristics of p
ports 2 to th
7 are shown in Fig 2‐9, and a the phaase characte
eristics of these ports arre given in Fig 2‐10. n loss and phhase responsse characteristics for porrts 8 to 13 arre shown Similarly, the transmission
n Fig 2‐11 an
nd Fig 2‐12 respectively.
r
The circuitt model portt characterisstics are iden
ntical for in
th
he two halvees of the feed
d network, aas well as ports on oppossing sides of f the structurre due to syymmetry. The T
transmission loss rresponses exhibit < 1 dB d differencce at any particular p
frequency across the ba
and. The ccircuit mode
el does not consider thhe parallel coupling co
omponent th
hat will be present p
due to the close
e proximity of o the transm
mission liness. These diifferences must m
be conssidered in aan electromaagnetic mod
del and com
mpared to th
he circuit m
model results. Fig 2‐8 Input rreflection coefficient maggnitude (circcuit model)
2--70
mission loss characterist ics of ports 2
2 to 7 (circuitt model) Fig 2‐9 Transm
or ports 2 too 7 (circuit model) Fig 2‐10 phasee response fo
2--71
Fig 2‐11 Transsmission losss characteristtics of ports 8 to 13 (circuit model) or ports 8 too 13 (circuit m
model) Fig 2‐12 Phasee response fo
2--72
2..3.3
The finite groun
nd “Coplanaar waveguid
de (CPW)” and secondaary slot transmission line are determiined. As previously In
n this sectio
on the copla
anar transm ission line dimensions d
m
mentioned, th
hese lines diimensions a re selected to satisfy the PCB proceessing constrraints on gaap width, and overall arrray width lim
mited by the mechanical cconstraints oof the radom
me. In Fig 2‐‐13 depicts the proposed
d geometry ffor the 85 Ω “Coplanar w
waveguide (CPPW)” line. f
b
a c
d
e
Fig 2‐13 Geom
metry of CPW
W transmissioon line mode
el Paarameters: a ‐ source track width == 3.8 mm; b ‐ ground track gap = 0..638 mm; c ‐ ground trrack width = 3.0 mm; d ‐ substrate th ickness = 0.1
125 mm (εr =
= 2.8); e – meetal claddingg = 35 um (C
Cu); f ‐ samplle length = 293 mm. A two port simulation is conducted in CST Microwave Stud
dio to confi rm that the
e desired ch
haracteristic impedance of the “Copplanar wave
eguide (CPW)” line has bbeen achievved. The reesult of this ttest as show
wn in Fig 2‐144, indicate th
hat the line h
has a charactteristic impe
edance of 855 Ω over the pass band o
of 850‐960 M
MHz. 2--73
Fig 2‐14 Port Impedance o
of the 85Ω CCPW (Smith cchart normallised to 85Ω)) To
o characterizze the slot line impedannce, a two port p
simulation is also coonducted to
o confirm th
hat the 127Ω
Ω characterisstic impedannce of the slo
otline has be
een achievedd. A model iss created ussing the parameters as sshown in Figg 2‐15. The results of this simulatioon shown in Fig 2‐15 in
ndicate that tthe line impe
edance of 1227 Ω has been achieved.
a
b
d
c
Fig 2‐15 Geom
metry for 127
7Ω slot line m
model Paarameters: aa‐ conductor width = 3 m
mm; b – gap =
= 0.5 mm; c ‐‐ substrate hheight = 0.125 mm (εr = 2.8); d ‐ sam
mple length =
= 100 mm; e –– metal cladding = 35 um
m (PEC). 2--74
e
Fig 2‐16 Impeedance of the
e 127Ω slot lline (Smith chart normaliised to 127Ω
Ω) 2..3.4
Closse proximity of the transsmission line
es Ass the “Coplaanar wavegu
uide (CPW)”” and slotline are in clo
ose proximitty, it is nece
essary to ch
haracterize the coupling and losses aassociated with this. As tthe “Coplanaar waveguide
e (CPW)” grround trackss are not co
onventional iin that theyy have a finite width groound plane, there is exxpected to b
be some influ
uence on thee properties of the slotlin
ne. Addingg to this complexity is th
he exploitation of the ground tracck of the “Coplanar “
waveguide w
(CCPW)” to fo
orm one co
onductor of tthe slot line. o investigatee this hypothesis, a moddel comprising of the “C
Coplanar waaveguide (CP
PW)” and To
th
he two paralllel slotlines is created, as shown in
n Fig 2‐17. There T
are fivve ports use
ed in this m
model. It waas not possib
ble to feed tthe “Coplanaar waveguide (CPW)” seection directly with a w
wave port, as its width would exceed that of the “Coplanar w
waveguide (CCPW)” groun
nd tracks, in
nteracting with the close
ely spaced sslotlines. To
o alleviate this, t
Port 1 is connected to the “C
Coplanar waveguide (CPW
W)” input viaa a short coaaxial cable w
with the charracteristic im
mpedance off the “Coplanar waveguide (CPW)” liine. The me
easurement plane is shiffted forward in order to
o de‐embed tthe coaxial liine phase coomponent. 2--75
2 to 5. As thhe slot line iss actually A wave port is connected to each endd of the slottline, Ports 2
aveguide (C PW)” geome
etry, the po
orts are arraanged to lim
mit mode paart of the “Coplanar wa
in
nteraction. TThis is achievved by addinng a 90 degree bend to the outputt ports such that the slot line portss and “Coplanar waveguiide (CPW)” p
ports have 90 degree oppposed meassurement own in Fig 2‐‐17. Each 990° elbow exxtension hass the same eelectrical len
ngth. The pllanes as sho
seeries λg/4 maatching transsformer is ussed to transfform the fee
ed point imppedance mea
asured at th
he “Coplanarr waveguide (CPW)” disccontinuity to
o the main “Coplanar waaveguide (CP
PW)” line im
mpedance. TThe slots in the ground ttracks of the
e “Coplanar waveguide ((CPW)” are designed to
o transition tto an odd m
mode and couuple energy to the close
e proximity sslotlines, cha
annelling th
he energy to Ports 2 to 5
5. With Port 1 as the sou
urce, the pro
operties of innterest are th
he phase an
nd amplitude measured at each subbsequent po
ort with resp
pect to Port 1 and the reflection r
co
oefficient maagnitude me
easured at PPort 1 as the
e “Coplanar waveguide ((CPW)” grou
und track w
width, short ccircuit stub, a
and slotline ttrack width aare altered. Source
Port
P 3
Portt 1
Meeasurement plane
Port 2
Ser
eries
Slot discontinuity
Port 5
Shorrt circuit
Port 4
traansformer
in ground trracks of
Fig 2‐17 Schematic layoutt for the 5 poort feed netw
work evaluattion Ass there is latteral symmettry in the deesign, the am
mplitude and phase shift ffrom input tto output sh
hould be identical at the parallel porrts. There is expected to be a slight pphase and amplitude sh
hift when co
omparing th
he non‐paraallel ports as the feed network haas included a series trransformer and a a short circuit stub which breaak the length
hwise symm
metry. The following f
seections detaail simulations conducteed to analyse the pow
wer divider ssection of the t
feed neetwork in iso
olation. 2--76
2.3.4.1 Pow
wer divide
er characte
eristics witth change
e in “Copllanar wav
veguide
(CPW)” stub length
hows the dimensional reference for f the “Coplanar waveeguide (CPW
W)” stub Fig 2‐18 sh
component of the feed network. TThe stub, wh
hich is ideallly λ/2 long, is used to provide p
a phase refereence and lim
mited impedaance matchin
ng. As it is an extension from the fee
ed point, it is importaant to characcterize this ccomponent iin terms of p
phase and am
mplitude as the stub length is varried from 160
0‐175 mm. Stub len
ngth 160-175
5 mm
Fig 2‐18 CPW termination stub length reference Th
he maximum
m phase error at the outpput ports of tthe power divider sectio n with respe
ect to the so
ource input aas the “Copla
anar waveguuide (CPW)” stub length is varied is ppresented in Fig 2‐19. Th
he results indicate that a
a stub lengthh change of 15mm has vvery little inffluence on th
he phase reesponse, pro
oducing a pha
ase differencce of less thaan 1.5 degrees at the outtput ports. The stub leength of 171 mm was se
elected (dottted black traace) as it exh
hibits the leaast overall im
mpact on th
he phase ovver the desired band ffrom 850‐96
60 MHz. In
n Fig 2‐20 tthe input reflection r
co
oefficient maagnitude of tthe power d ivider section indicates tthat the stubb length can tune the in
nput matchin
ng without im
mpacting onn the phase response. The dotted linne corresponds with th
he final stub length of 171 mm. 2--77
STUB LENGT
TH (MM)
Fig 2‐19 Maxim
mum port ph
hase differennce with variiable CPW sttub length (inn mm) STUB
B LENGTH (M
MM)
Fig 2‐20 Refleection coefficcient magnittude with varriable CPW sstub length (m
mm) 2--78
he relative siignal magnitude at each output port is presented
d in Fig 2‐21.. The signals at each Th
off the output ports had a maximum ddifference in level of less than 0.5 dB The available power att the input iss split four w
ways, and a trransmission level of 7 to 8 dB is seenn at the outp
put ports. Th
his level of lo
oss is typical of a four waay splitter. STUB
B LENGTH (M
MM)
Fig 2‐21 Maxim
mum port m
magnitude diffference with
h variable CP
PW stub lenggth (mm) wer divide
er characte
eristics witth change
e in “Copllanar wav
veguide
2..3.4.2 Pow
(CPW)” ground
g
trac
ck width
Sh
hown in Fig 2‐22 is the gground trackk width referrence used to characteri ze the effects of this paarameter on
n the feed ne
etwork perfoormance. Th
he “Coplanar waveguide (CPW)” grou
und track w
width is parametrically ad
djusted whillst monitorin
ng the reflecction coefficcient magnitude, and trransmission p
phase and am
mplitude at eeach port. 2--79
Ground track widtth Fig 2‐22 CPW ground trackk geometry rreference Th
he maximum
m phase error at the outpput ports of tthe power divider sectio n with respe
ect to the so
ource input aas the “Copla
anar waveguuide (CPW)” ground track width is vaaried from 2 to 4 mm is presented in
n Fig 2‐23. Fig 2‐23 Maxim
mum port ph
hase offset w
with change in CPW ground track widdth Th
he “Coplanar waveguide
e (CPW)” groound track w
width has an influence onn the phase rresponse ass it can define the leve
el of couplinng between the “Coplanar wavegu ide (CPW)” and the slotlines. A smaller phase
e differentiaal is seen fro
om lower values of the ““Coplanar wa
aveguide 2--80
CPW)” groun
nd track wid
dth, produci ng a phase difference of less thann 1.5 degree
es at the (C
ou
utput ports for a 2mm width. Hoowever, the results sho
own in Fig 22‐24 indicate that a naarrower ground track width w
causes an increase
ed signal ma
agnitude varriation at the output po
orts. In Fig 2‐21 the in
nput reflecti on coefficient magnitud
de of the poower divider section in
ncrease for lo
ower values o
of the “Coplaanar wavegu
uide (CPW)” ground trackk width. CPW GROUN
ND TRACK WID
DTH (MM)
FFig 2‐24 Portt level with change in CPW
W ground traack width (in
n mm) CPW GROUN
ND TRACK WIDTH
W
Fig 2‐25 |S11| vs. frequency with channge in CPW gground track width (in mm
m) 2--81
2..3.4.3 Pow
wer divide
er characte
eristics witth change
e in “Copllanar wav
veguide
(CPW)” discontinuit
d
ty gap wid
dth
In
n this sectio
on the “Cop
planar wavegguide (CPW)” discontinuity as deppicted in Figg 2‐26 is an
nalysed. The maximum phase erro r at the output ports off the power divider secttion with reespect to the source input as the “Coplanar waveguide w
(C
CPW)” grounnd discontin
nuity gap w
width is varied from 1 to 7.75 mm is presented in
n Fig 2‐27 T
The “Coplanaar waveguide
e (CPW)” diiscontinuity gap width has a slight innfluence on the phase response typpically <2.0 d
degree of ph
hase variatio
on over this range as sh own in Fig 2
2‐27. A grea
ater discontiinuity gap w
width also caauses an inccreased signal magnitudde variation at the outp
put ports, ass shown in Fig 2‐28. However thesse values are
e still small, remaining b
below 0.6 dB
B In Fig 2‐29,, larger disco
ontinuity n coefficientt magnitude of the powe
er divider gaap widths are seen to reduce the inpput reflection
seection, as a m
more significant discontinnuity is prese
ented to the propagatingg signal. Fin
nite ground CPW
C
Slot Ground
Source
Ground
Grround discon
ntinuity width
h =1 to 7.5m
mm
Fig 2‐26 CPW
W discontinu
uity planar poower dividerr 2--82
CPW
nuity gap wid
dth (in mm) Fig 2‐27 Port phase with cchange in CPPW discontin
CPW
Fig 2‐28 Port llevel with change in CPW
W discontinuiity gap width
h (in mm) 2--83
CPW
W
GAPP
d network w
with change iin CPW disco
ontinuity Fig 2‐29 Refleection coefficient magniitude of feed
gaap width (in mm) 2.3.5
ns Antenna port T‐junction
Th
he antenna port T‐junction is actuaally a short section of slot line thaat is tapped
d off the seecondary slo
ot line. For the analysis oof the antenna port T‐junctions, an eentire sub‐arrray feed neetwork consisting of thre
ee series fedd antenna po
orts is consid
dered, as seeen in Fig 2‐30
0 . The S paarameters are recorded at each iterration of slott line gap wiidth to dete rmine the amplitude an
nd phase pro
ofile across the series subb‐array. It is expected that the pha
ase slope seeen at each p
port will be d
different eithher side of th
he design frequency, bu
ut coincide a
at the designn frequency. This is typical of a seri es fed arrayy. As the ub array is a a convention
nal series feed network, the phase is in advancce above the design su
frequency and
d lags below
w the design frequency. Each of the four sub‐ar rays in the ccomplete phase profile as they havve identical structure. feeed network will have the same ampplitude and p
Eaach sub‐array on either sside of the ffinal feed network is fed back to bacck; therefore
e the two 2--84
phase profiles will be in direct opposition. This is the profile required of a centre feed array network. The phase and amplitude are measured at each of the three antenna ports with respect to the input at port one. Each sub array comprises of a slotline and three antenna ports with an impedance of 129 Ω. The variables analysed are the antenna port gap, slotline track and the slotline stub length. Slot line gap
P4
P3
Slot line track width
Antenna port gap
P2
Slotline termination
length
Port 1 feed
Fig 2‐30 Schematic of the three element slotline sub‐array feed network with variable parameters Note: The following results are for just the slotline connected to the dipole ports. These results will not represent cross coupling. The results presented in Fig 2‐31 to Fig 2‐34 depict the phase variation across the frequency band of 850‐960 MHz for the three antenna ports of the series sub‐array for antenna port slotline gaps of 0.2 to 0.7 mm. The anticipated phase slope seen at each port being different either side of the design frequency, but coincide at the design frequency of approximately 910 MHz was observed. The maximum phase taper across the sub‐array at the upper band edge (960 MHz) was around 65 degree at a gap of 0.2 mm, extending to about 75 degrees at 0.7 mm. 2-85
he results sh
how that rela
atively consiistent and flat responsess are observved for Portss 2 and 3 Th
Ass Port 4 is in
n close proximity to the termination stub, it experiences a m
more rapid cchange in th
he S‐parameeter transmission magnnitude than the other ports. A m
maximum amplitude vaariation betw
ween all an
ntenna portss of around
d +/‐ 1.5 dB
B is observeed across th
he entire frequency ban
nd. The dipole gap spacinng of 0.5 hass been selectted in the finnal arrangem
ment as it saatisfies the m
minimum track spacing annd has the le
east variation
n in amplitudde response over the paass band. M) model Fig 2‐31 Port ccharacteristics where thee dipole portt gap width is 0.2mm (EM
2--86
M) model Fig 2‐32 Port ccharacteristics where thee dipole portt gap width is 0.5mm (EM
Fig 2‐33 Port ccharacteristics where thee dipole portt gap width is 0.6mm (EM
M) model 2--87
M) model Fig 2‐34 Port ccharacteristics where thee dipole portt gap width is 0.7mm (EM
2--88
2..3.6
Phase compensation slotlin e terminatio
on A phase compensation slot line term
mination is employed at the end oof each sub‐‐array to baalance the behaviour of antenna porrts at the extreme end to the characcteristics of tthe inner an
ntenna portss. To evaluate the effecttiveness of tthis terminattion, the fouur port mode
el seen in Fig 2‐30 was u
used with parametric conntrol of the sslot line term
mination lenggth. The tran
nsmission peerformance is analysed as the stub length is varried in length from 1 to 17 mm. Th
he results arre presented
d in Fig 2‐35 tto Fig 2‐37.
ub length is 1
1 mm (EM) m
model Fig 2‐35 Port ccharacteristics where thee slot line stu
2--89
ub length is 9
9 mm (EM) m
model Fig 2‐36 Port ccharacteristics where thee slotline stu
ub length is 1
17 mm (EM) model Fig 2‐37 Port ccharacteristics where thee slotline stu
Th
he result of tthe parametric sweep off the slotline stub length indicates m inimal impacct on the ph
hase and am
mplitude. Th
hese stub lenngths (1 to 1
17 mm) represent a veryy small phase
e shift of ap
pproximatelyy 1 to 18 deggrees. As th e feed to the
e network is in series, thhe phase shifft in each 2--90
ort is related
d and minim
mizes the efffective control of phase. The slotlinne stub of 17
7mm has po
beeen selected
d as it provides similar feeed point geo
ometry to tha
at of the othher ports. 2.3.7
Parametric ssweep of the
e slotline traack width Ass the “Coplaanar waveguide (CPW)” aand slotline ttransmission
n lines are inn close proxim
mity, the efffects of thee width of the slotline ttrack must be b evaluated
d with respeect to feed network peerformance. The four po
ort model seeen in Fig 2‐3
30 was again
n used, this ttime with pa
arametric co
ontrol of thee slot line trrack width. The transm
mission performance is a nalysed as the t track w
width is varied
d from 2.4 to
o 4 mm. Thee results are shown in Fig
g 2‐38 to Fig 2‐40. Fig 2‐38 Port ccharacteristics where thee slotline track is 4mm w
wide (EM) moodel 2--91
model Fig 2‐39 Port ccharacteristics where thee slotline track is 3.2mm wide (EM) m
m wide (EM) m
model Fig 2‐40 Port ccharacteristics where thee slotline track is 2.4 mm
Frrom the anaalysis of these results, tthe slotline track width has a negliigible impacct on the neetwork performance. The T most nooticed effectt is a slight shift in thee zero tilt crross over 2--92
ed a zero de
egree relativee phase at 9
913 MHz, frequency. A 4.2mm slotline track wiidth produce
w
whereas a traack width of 2.4 mm hass a zero phasse frequencyy of 905 MHzz. This phase
e error is no
ot critical ass any mutua
al phase offsset is cancelled out as both b
sides oof the complete feed neetwork havee a phase tilt in oppos ition to eacch other. Th
he “Coplanarr waveguide
e (CPW)” grround track w
width selected is 3.5mm
m as it is close
e to the “Cop
planar wavegguide (CPW)” ground trrack width. 2..4
Elec
ctromagne
etic simula tion of the planar centre feed n
network
A detailed eleectromagnetic (EM) moddel of the en
ntire feed network was ccreated and analysed ussing CST Miccrowave Studio. The anttenna ports in this feed network w ere defined as wave po
orts to allow
w the phase and amplituude responsse to be analysed includding couplingg effects. Th
he radome aand other peripheral mouunting structture used (as explained iin Chapter 5
5) are not in
ncluded in orrder to reduce the compputation tim
me. The portt arrangeme nts for the a
array are sh
hown in Fig 2
2‐41 only one
e side of thee network has ports assum
ming mirror symmetry. Port 4
Port 6
Port 10
Port 8
Port 2 Po
ort 1
P 12
Port
Fig 2‐41 The C
CPW array fe
eed networkk port configu
uration 2--93
2..4.1
Elecctromagneticc model resuults he transmisssion loss characteristics aand the phasse response from the EM
M model are depicted Th
in
n Fig 2‐42 and Fig 2‐43 . As a cerrtain amount of cross coupling c
betw
ween the “Coplanar w
waveguide (CPW)” and th
he slotlines w
was expected
d the respon
nse of the EM
M model differs with th
hat of the ciircuit model, showing a slight asym
mmetry betw
ween the twoo halves of the feed neetwork. Thee phase resp
ponse is typpical of a fee
ed network for a centree fed array, where a reelatively consstant gradient is evidentt. The worstt case phase difference bbetween the antenna po
orts is 100 degrees, whicch occurs at 880 MHz. TThis phase error is tolerabble as each ssub array is fed back to
o back cance
elling out thee frequencyy dependent phase errorr. The circu
uit model mplitude resp
ponse exhibiits undulatio
on across simulation exhibits a similar phase errror. The am
he frequencyy band of intterest as a reesult of cross coupling. The port am
mplitude diffe
erence is th
ap
pproximatelyy 7 dB in the case of thhe EM mode
el; whereas the circuit m
model was 4 4 dB The diifferences in
n amplitude are most likkely the resu
ult of cross coupling c
bettween the “Coplanar w
waveguide (CPW)” main liine and the sslotlines. 0
‐500
‐550
‐600
‐700
‐20
‐750
‐800
‐850
transmission loss dB
difference in phase ˚
‐10
‐650
‐30
‐900
‐950
‐1000
‐40
0 880 890 9900 910 92
20 930 940 950 960 9970 980
8550 860 870
Frequency ((MHz)
arrg s21
args41
args61
s21
s41
s61
Fig 2‐42 Phasse and amplittude of the l ower array p
ports (EM mo
odel) 2--94
‐500
0
difference in phase ˚
‐600
‐1
10
‐650
‐700
‐750
‐2
20
‐800
‐850
‐3
30
‐900
transmission loss dB
‐550
‐950
‐1000
‐4
40
850 860 87
70 880 8900 900 910 920 930 94
40 950 960 970 980
Frequency
y (MHz)
a
args81
args101
args121
1
s81
s101
s121
ports (EM mo
odel) Fig 2‐43 Phasee and amplitude of the uupper array p
Note: Figure 2
2.42 and 2.43
3 are the ressults from EM
M simulation of the uppeer and lower array po
orts including the “Copla
anar wavegu ide (CPW)” ffeed. These models will ppresent crosss co
oupling. Th
he |S11| of the EM mo
odel is preseented in Figg 2‐44 the feed netwoork exhibits a 10 dB reeflection coeefficient maggnitude banddwidth comm
mencing at approximate
a
ely 872 MHzz. This is in
nside the finaal bandwidth
h specificatioon of 850 – 9
960 MHz. Ho
owever, as m
mentioned previously th
hese results are generate
ed with no rradome or p
peripheral mounting com
mponents inccluded in th
he model. Th
hese compon
nents act as a dielectric loading to tthe feed netw
twork and ha
ave been sh
hown to shiftt the response approxim
mately 40 MH
Hz lower in frrequency. Thhe final ante
enna port sp
pacing allowing for dielecctric and raddome loadingg was one w
wavelength, oor approxima
ately 290 m
mm at 910 MHz. 2--95
Fig 2‐44 |S11| of the plana
ar feed netwoork (EM mod
del) 2..5
Sum
mmary
A feed netwo
ork suitable ffor a centre fed collineaar array was introduced in Section 2
2.1 which his project. The feed network saatisfies the dimensionall and electrrical requirements of th
ch
haracteristicss have been analysed ass a planar m
multi‐port nettwork with tthe aid of [4
40]Design sttudio for a ciircuit model simulation, and [40] microwave stu
udio. By dessigning the “Coplanar w
waveguide (CPW)” main feed f
line at high characcteristic impe
edance, the overall widtth of the arrray can be rreduced, mee
eting the meechanical con
nstraints set forth. he limitation
ns of the “C
Coplanar waaveguide (CP
PW)” feed network n
havve been disccussed in Th
Seection 2.2. The printed
d network h as low effecctive therma
al dissipationn when compared to sttate‐of‐the‐aart metal tub
be constructtion, and he
ence is suita
able for low power base
e station ap
pplications. The minimu
um track spaacing of 0.5m
mm imposed by standardd reel to ree
el printed circuit processsing is accou
unted for in tthe design. work has bee
en created in Section 2.4 and port A [40]Design sstudio circuitt model of thhe feed netw
haracteristicss were analyysed. This m
model has be
een used as to benchmaark performa
ance and ch
2--96
was further developed with associated electromagnetic models of particular sections of the feed network. In Section 2.4, an electromagnetic model of the entire feed network was generated. All electrical expectations for the feed network were achieved, including low transmission loss, operation in the desired frequency band, and relatively stable phase profile. The linear phase profile has been verified by experimental testing of the complete base station array in Chapter 5.3. 2-97
C
Chapter 3
A broadband prin
nted dipole
e element fo
or a collineear array
3..1
duction
Introd
pter, the devvelopment ppath from a conventiona
al cylindricall dipole to the novel In this chap
planar dipolle element for f a printedd collinear array is descrribed. The ccharacteristics of the printed dipo
ole element ffed in series by a slotline
e transmissio
on structure are investigated and validated ussing numerical simulationn in CST Micrrowave Studio. dipole is a sim
mple assembbly with two
o λ/4 wires or tube elemeents on eithe
er side of A standard d
the source. A single dipole on its own has relatively a sm
mall proporttion of the available nergy directe
ed at the hhorizon due to the wid
de elevationn beam widtth of its radiated en
omnidirectio
onal pattern
n, which limiits its appliccation. To increase thiss radiation le
evel, the pattern shape must be compressedd in the elevvation plane
e. This is achhieved by crreating a ed in Chapter 1. The com
mplexity in creating a vertical array of a number of dipoless, as discusse
conventionaal omnidirectional arraay of vertically oriented dipoless for base station applicationss lies in the rrouting of thhe feed liness/cables to each dipole eelement. To preserve the omnidirrectional perrformance, tthe array geometry musst be as closse as possible to axis symmetric. To achieve this, the feeed network is normally routed in thhe core of th
he array, with the dip
poles mounte
ed concentriically as slee
eves on a cyllindrical suppport pipe. Th
his is the most comm
mon implem
mentation foor conventional omnidirectional bbase station
n arrays. Increasingly,, array anten
nnas are beinng designed using printed circuit tec hnology and
d as such, the dipole elements are normally y planar and
d integrated
d with the ffeed network. The advantage o
of this directiion is lower ccost, high repeatability, a
and a reducttion in the nu
umber of soldered co
onnections leading to l ess passive intermodulation. Speecialised dessigns are required to mask the trransmission lines from the t radiating
g elements. For omnidirectional collinear basse station arrays, these aadvantages aare countere
ed by the com
mplexities asssociated with feedingg the array elements andd processing a high aspecct ratio printeed substrate
e. p
dipole which is suitable forr printed This chapter introducess a high peerformance planar onal collinea
ar arrays. TTo prove thaat this new topology t
ha s more conttrol over omnidirectio
3--98
impedance and has a more uniform radiation pattern characteristic than currently available planar dipoles, a number of investigative tests are conducted. Section 3.1 presents a review of broadband planar dipoles, and explains how the planar dipole is the analogue of a cylindrical dipole in terms of current distribution and radiation pattern. Section 3.3 includes the capabilities of planar dipoles for impedance matching and self‐choking to prevent out of phase radiation, and the ability to reduce aperture blockage. The unique structure of the proposed planar dipole for printed omnidirectional collinear arrays is introduced in Section 3.3, along with an explanation of the bandwidth enhancement techniques applied to the geometry in the form slots and notches. The enhancement techniques are analysed and validated with the aid of electromagnetic simulations [40]. The model of the proposed planar dipole element is analysed separately from the array in order to characterize the feed point impedance and bandwidth characteristics. The dipole performance when in an array configuration is also discussed. The current distribution resulting from series feeding and the effect of mutual coupling on the array pattern stability are analysed. The requirements for the use of this dipole element in an omnidirectional array are also detailed. A summary of the investigations and results are presented in Section 3.4. 3.1 A review of existing broadband and planar dipoles
For conventional dipoles, bandwidth can be primarily controlled by the geometric length to diameter ratio. Smaller ratios i.e. Fat dipoles have lower Q factor (and hence broader bandwidth) but also result in reduced cross‐polar performance as there is less distinction between E and H plane geometry. The resonant length for these dipoles is shorter than for thin dipoles. Dipole bandwidth can also be increased by introducing slots and notches into the element [44], in this example as shown in Fig 3‐1 the dipole is fed at across A and B by a slot line with a gap spacing of 0.5mm. This example comprises of two sets of elements fed in parallel by a short section of slot line. Finding the right balance between match and band position will 3-99
result in a broad impedance bandwidth. The slots are used to create a dual band response. A parasitic element placed in front of one of the dipole arms, further increased the bandwidth This dipole without parasitic element can achieve a bandwidth of 22% 1.65 to 2.05 GHz with a reflection coefficient magnitude of ‐10dB.With the addition of a parasitic element, the bandwidth is 50% 1.66 to 2.71 GHz. The radiation pattern for this arrangement is directed forward with front to back ratio of approx. 10 dB this results in a realized gain > 6dB The asymmetry is the direct result of placing the parasitic element in front of one of the elements and slightly towards the feed gap. A
Arrangement for 50% bandwidth asymmetric radiation
B Arrangement for 22% bandwidth symmetrical radiation
Fig 3‐1 The broad banding technique used for [44] Parasitic elements were considered in the proposed design, these would have resulted in an increase of >10mm in the array width. These geometric obstacles delay the propagation of current around the dipole perimeter, independent of the physical length of the element. This increases the inductance which cancels out the capacitance. The net effect is to reduce the reactance that is impacting on the bandwidth. Dipoles treated in this way usually will have a lower resonant frequency. The bandwidth of a dipole is increased by introducing a disturbance to the Q of the dipole. This element being closely coupled modifies the resonance to form a second resonance. Careful adjustment of the geometry can yield a broad response in terms of reflection coefficient. 3-100
he bi‐conicall dipole show
wn in Fig 3‐2 (a) can achie
eve a multi o
octave bandw
width as its ggeometry Th
is able to occu
upy a large volume withi n the radian sphere. In tthis configurration, the fe
eed point mpedance reemains relattively constaant with chaange in freq
quency [16].. The large
e volume im
geeometry resttricts its use
e to heavy duuty applicatiions such as broadcast ppanel antenn
nas. The pllanar analogue of the bi‐‐conical dipoole is the bow
wtie dipole, shown in Figg 3‐2 (b). Th
his dipole do
oes not havve as broad bandwidth performance
e as the bi‐cconical dipoole, as radian
n sphere occcupation is only possible
e with a >600° dual trianggular elemen
nt [16] in onl y one plane.. Plan
n
(a)
viiew
(b)
Fig 3‐2 (a) Bicconical dipole
e (b) Bowtie dipole [16]
nother form
m of dipole which w
is com
mmonly used
d in base sta
ation arrays is the sleeve dipole, An
sh
hown in Fig 3
3‐3 [31]. This dipole dessign has low frequency rresonant ele ments which
h are not acctive at higheer frequencies, resultingg in a dual ressonant respo
onse [16]. BBy careful selection of tu
ube diameters and length
hs this can a pproach a broad band re
esponse. A ccomparison between du
ual band and
d broad band
d tuning of ssleeve dipole
e is shown in
n the inset off Fig 3‐3. In this case th
he two stronger resona
ant responsees are mergged togethe
er forming aa flatter, broadband reesponse. Thiss effect is sim
milar to broaad band‐passs filter tuning
g. 3--101
Virttual source a
at high Feedd point at low
w
frequuency
Dual
resonant
B
Broadband
Fig 3‐3 Sleeve dipole with virtual higgh frequencyy source: Inset ‐ synthesized brroad band re
esponse Pllanar dipoless found in th
he literaturee are generally the 2D an
nalogue of ccylindrical structures, faabricated on
n low cost printed p
circu it material. All are dessigned to m
minimize the azimuth diistortion in order o
to pro
ovide the beest omnidire
ectional performance poossible from a planar ellement. Som
me examples of these dippoles can be found in [24
4] [33] [34]. TThese arrayss all have “H
H” shaped diipoles. This sshape createes a safe path
h for the feed network off either micrrostrip or “C
Coplanar wavveguide (CPW
W)” to be roouted where there is no m
masking of t he radiation pattern. Th
he geometryy of the “H” shaped dipoole is such th
hat the radiating elemennts are space
ed out on eiither side of the feed nettwork. In thiis arrangeme
ent, the radia
ation patternn from each element is directed aw
way from the
e feed netwoork and com
mbined with a 180 degreee phase rela
ationship omnidirectionnal performaance. prroducing a sllightly oval o
TThe major diffference bettween these dipole desiggns and the proposed diipole design is in the in
ntegrated eleement match
hing and brooad banding approach. The proposedd dipoles are
e slot fed 3--102
nd passive coupled wh
hich providees broad baand perform
mance with a balanced current an
diistribution. wn that radia
ating surfacees of an ante
enna are highly reactive and feed cables that It is well know
mity to them
m will radiate out of phasse. This cablee radiation can cause arre routed in close proxim
deestructive interference to the primarry antenna p
pattern. The planar arrayy architecturre in [24] atttaches a section of coaxial cable too a grounded
d section in the planar ggeometry to
o remove th
he potential difference between thee feed cable
e ground an
nd dipole eleement groun
nd. This allows the cab
ble to be routed behind ddipoles with minimal inte
eraction. M
Much of the previous ressearch conceentrates on implementin
ng broad baand matchingg circuits raather than directly d
incre
easing the im
mpedance bandwidth off the dipoless (for examp
ple [36]). Th
he use of planar p
dipole
e elements allows for more flexibility in the creation off lumped co
onstant com
mponents witthin the dipoole geometrry. These ca
an be createed using stra
ategically po
ositioned taggs and notches within tthe dipole geometry. The T proposeed broadban
nd planar diipole structu
ures in this Chapter are ffed using “Co
oplanar wave
eguide (CPW
W)” primary ffeed line. Th
he feed netw
work discusssed in Chaptter 2 is desiggned to isola
ate the mainn feed network from th
he dipoles by b coupling to t a second ary set of slotline s
transsmission struuctures. Th
he planar diipole elemen
nts are tappe
ed off in seriees using a sh
hort section o
of slotline. 3..2
gn considerations fo
or an om
mnidirection
nal colline
ear array dipole
Desig
elem
ment
Ass elements aare added to
o an array, m
mutual coupliing of the ad
djacent dipolles effectively lowers th
he resonant ffrequency off all the dipooles in the arrray. Both tthe cylindricaal and planar dipoles tyypically are shorter than
n a half waave as the array a
size in
ncreases up to about 8 dipoles. An
or that mustt be consideered is radom
me loading. E.g. A fibergglass radom
me with a nother facto
6m
mm wall hass enough loading to reduuce the dipole resonant ffrequency byy 25MHz at 8
850 MHz. Kn
nowing the loading effects and offssets can sign
nificantly reduce simulat ion task as it can be trreated as a post process [41]. Th
he proposed
d array dipolles are desiggned to suit their positio
on in the ar ray and as such, s
the diipole elemen
nts at the farr ends of thee array are d
different. Thiis is due to tthe inside dipole arm beeing loaded by the pre
evious dipol e and the outside dipo
ole arm expposed to fre
ee space 3--103
onditions. Increasing the
e length of thhe central co
onductive support abovee the dipole, will then co
exxpose the dip
pole to similar loading ass the inside e
elements. 3..2.1
The transition frrom cylindriccal to planarr topology To
o allow the dipole to be
e used in plaanar type arrays and takke advantagee of a printe
ed circuit prrocess, the element mu
ust undergo another traansition phasse from cyli ndrical to planar, as sh
hown in Fig 3‐4. Planar elements arre an advantage as theyy can be fabbricated as a a printed circuit using highly h
repea
atable proceesses. This to
opology is suitable for iintegrating matching m
circuits into the dipole. The first chaange is feed
ding a monopole above a coaxial sle
eeve (Fig eed multiplee series fed e
elements 3‐‐4a), to a feeed through cable form tthat can be routed to fe
(FFig 3‐4b). Thee cylinder ele
ements are tthen replace
ed with plana
ar elements as shown in Fig 3‐4c. Finally, the co
oaxial cable is replaced w
with a “Coplaanar waveguide (CPW)” ttransmission
n line (Fig pole and feed
d are now suuitable for fabrication ontto a PCB. Thee next stage, and the 3‐‐4d). The dip
su
ubject of the thesis, is to further enhance th
he antenna performancee. This is done by prroviding a ceentre feed fu
unction for tthe planar aarray. To ach
hieve this, thhe dipoles arre fed by paassive coupleed slot lines.. The methodd allows for energy to be
e coupled off
ff the main “Coplanar w
waveguide (CPW)” line in
n a controlle d way. In Figg 3‐4e, the p
passive couppled dipole iis fed via tw
wo slot lines energised b
by the “Coplaanar wavegu
uide (CPW)” discontinuitty discussed in detail in
n Chapter 2. 3--104
(a) (b)
(c)
(d)
(e)
Fig 3‐4 Cylind
drical to planar element eevolution (aa) A cylindrical dipole – ce
entre fed (b
b) A cylindrical dipole – se
eries feed thhrough (cc) Cylindrical dipole plana
ar analogue –– series feed
d through (d
d) CPW fed p
planar dipole (ee) The proposed slotline ffed planar diipole topologgy A model of a cconventional cylindrical dipole and aa planar dipo
ole was creatted as in Fig 3‐5. The pllanar dipole is supposed
d to be an a lternative to
o the cylindrrical dipole. As the plana
ar dipole haas less volum
me, it will normally fall shhort in bandw
width when compared too a cylindrica
al dipole. To
o increase th
he bandwidtth of a planaar dipole, th
he Q must be b reduced. TThis can be done by in
ntroducing a second parasitic elem ent or smalll notches in
n the forcinng resonance
e at two frequencies, sspreading the
e band. here is a bro
oader response than the equivalent cconventional cylindrical ddipole, as seen in the Th
reesults in Fig 3
3‐6. The width of the dippole has been increased to give the ddipole the eq
quivalent flaat plate areaa of a cylindrical dipole. i.ee. a cylindriccal dipole off 31mm diam
e electrical equivalent e
too the area of o 62mm meter is the
w
wide planar dipole of the same lengthh [16]. 3--105
Φ32 (a) 172mm
(b
b)
φ 63
Fig 3‐5 A cylin
ndrical sleeve
e dipole and the analogu
ue planar dip
pole Fig 3‐6 |S11| for the conve
entional cylinndrical dipole
e compared to the plana r dipole 3--106
3..3
p
planar
p
dipo
ole
The proposed
Th
he structure of the proposed dipole is designed tto integrate with a linea r array feed network ass described in Chapter 2. The width of the final planar dipole collinear a rray is limite
ed by the m
mechanical diimensions off the propossed radome. This restrictts the distannce the dipolle can be aw
way from the t
main fe
eed structurre, (a) in Fig F 3‐7, and
d defines thhe azimuth pattern om
mnidirection
nal performa
ance. If the spacing is to
oo large, the azimuth pa ttern becom
mes more ovval shaped rather r
than circular (forr an omnidirrectional array). The dippole gap spacing (b) deetermines th
he feed point impedancee for the dipole, which sets the charracteristic im
mpedance off the slotlinee (c). Slots placed in the ends of the dipole arms (d) are usedd to help reco
onstitute th
he omnidireectional azim
muth patte rn. Planarr elements naturally hhave a com
mpressed om
mnidirection
nal azimuth p
pattern due to their 2D sstructure. Th
he slots are pplaced parallel to the leeading edge of the dipole
es. To broad en the dipolle impedance, notches (ee) are removved from th
he tips of each dipole arm. To prov ide inter‐ele
ement chokin
ng of the dippoles, the inside arm leength of the dipole (f) is adjusted too cancel out the currents that may bbe induced onto the trransmission ssections between the diipoles. This dipole is coupled to a ““Coplanar wa
aveguide (C
CPW)” main feed networrk. A slot lin e transmission line (c) iss formed bettween the “Coplanar w
waveguide (C
CPW)” ground tracks (g) and a secon
nd track run
nning in paraallel. This iss used to feeed the dipolles as a second layer trannsmission ne
etwork and p
prevents crosss talk from the main “C
Coplanar waaveguide (C
CPW)” feed coupling directly to the dipolees, suppresssing any un
ndesirable radiation. The dipole leength (h) haas been sele
ected to ressonate at the centre frequency of 903 MHz, taking t
into aaccount the radome loa
ading and m
mutual coupling. The diipole startingg geometry is based on aa convention
nal half wave dipole. The dimensions are then ad
djusted to co
ompensate for dielectric loading and mutual coupling effects s with CST microwave sttudio [40]. The final dimensions werre optimised with the assistance of nnumerical simulation (C
CST microwave studio) [4
40]. 3..3.1
Prop
posed planar dipole struucture Ass the slotlinee transmissio
on lines feedding the prop
posed dipole are balanceed, the plana
ar dipoles m
maintain a balanced b
currrent distribbution, leading to an efficient e
arraay pattern which is co
omparable to the cylindrical form. A compariso
on of the surface currennts is made between th
he conventio
onal coaxial collinear serries feed ele
ement in planar form [344] and the proposed p
feeed in Fig 3‐8
8. In Fig 3‐8
8(a), a large pportion of th
he available current is cooupled to the dipole, leeaving less avvailable for ssubsequent ddipoles alongg the series ffeed. The cuurrents on th
he dipole arrms are also
o asymmetricc. A sectionn of the prop
posed topology shown iin Fig 3‐8(b) displays m
much more evvenly distributed currentts. Only a co
ontrolled portion of the eenergy is coupled off th
he main feeed network, allowing thhe subseque
ent dipoles to have moore evenly matched am
mplitude in tthe series fed array. In thhis situation, the currentts in the arraay will be via
able over a longer distance enablingg a large num
mber of elem
ments. The e
extra slotlinee tracks betw
ween the diipole and main m
feed lin
ne effectivelyy separate the t current paths, prevventing out of phase in
nterference ffrom main fe
eed. 3--107
(hh) (b) (a) (d)) (e) (f)
(g) (c)
(a) Element sp
pacing from ssupport (patttern) (b) Dipole gap spacing (patttern) (c) Slotline (d) Dipole slotss (pattern) (e) Dipole notcch depth (maatching) de length (oddd mode cho
oke) length =
= λ/4*Vp (f) Dipole insid
nd track (g) CPW groun
Fig 3‐7 Propo
osed planar d
dipole schem
matic and inte
erface to feed network (w
with parame
eters) 3--108
(a) (b)
Fig 3‐8 Curren
nt distribution for dipoless with differe
ent CPW feed methods (aa) Planar dip
pole fed directly from “C oplanar wavveguide (CPW
W)” (b
b) The prop
posed metho
od of feedingg a planar dip
pole with slo
otlines couplled from a “Coplanar w
waveguide (CPW)” 3..3.2
posed planar dipole charracteristics
Prop
Th
he final topo
ology for the
e proposed pplanar dipole
e is shown in
n Fig 3‐9. Thhis dipole is ffrom the up
pper section
n of the fina
al collinear aarray antenn
na; hence the centre traack of the “Coplanar w
waveguide (CPW)” cannott be seen. Thhe planar dip
pole is designed to work k over a band
dwidth of att least 10%. 3--109
dimensions o
of the propossed planar dipole (in mm
m) Fig 3‐9 Final d
TThe structuree was simulatted in CST m
microwave studio, and the
e |S11| resul ts are depictted in Fig 3‐‐10. The pro
oposed dipole achieves aa 15 % bandwidth centre
ed at 916 M
MHz, which m
meets the sp
pecification tto cover the 850‐960 MH
Hz band. Th
he key to broad b
band performancce over and
d above a standard s
dippole is in reactance caancellation. TThe main rea
active impacct for this dip
pole comes ffrom its closee capacitive coupling to
o the feed network strrips. This afffect is canccelled out by introducinng some me
ethod of in
ncreasing thee inductance
e without ad ding to the capacitance. This is don e by placingg notches an
nd slots in th
he perimeterr of the dipolle to make itt look electrically longer tthan its physsical size. Th
he effectiven
ness of the techniques t
uused improvve the performance of t he planar dipole are also evaluated
d using Fig 3‐10. The strructure wherre either the dipole slotss or notches are filled in
n Fig 3‐10 (a) is compared
d to the propposed dipole Fig 3‐10 (b). 3.3.3
The planar dipole optimisattion method
Th
he dipole staarting geome
etry is basedd on a conventional half w
wave dipole . The dimensions are th
hen adjusted
d parametrica
ally to comp ensate for dielectric load
ding and muttual couplingg effects. Th
he broad baand perform
mance of thee radiation pattern p
of th
he completee array is monitored m
du
uring this prrocess. with CST microw
wave studio [40] optimissation and pparametric sw
weep. A paarametric sw
weep is used
d to define useful geom
metric limitss for the opptimiser. Pa
arametric sw
weep of the slot depth an
nd notch filliing in Fig 3‐1
10 and notch depth is shoown in Fig 3‐‐12 3--110
(a) (b
b)
models used (a) slots and
d notches filled in (b) no filling Fig 3‐10 The ccomparison m
he |S11| peerformance seen in Figg 3‐11 is virtually v
uncchanged witth and with
hout the Th
au
ugmentation
n slots, exce
ept for a sligght variation
n in bandwiidth from 166.5 % to 15
5 %. The beenefit of thee slots is evid
dent in the aazimuth patttern perform
mance. If th e dipole nottches are reemoved, a laarge shift in |S11| perform
mance is observed as sh
hown in Fig 33‐11. To reccover the deesired frequency coveragge, the dipo le length needs to be made shorter.. These nottches are to
o compensatte for reacta
ance associaated with the close proxximity of thee dipoles to the slot lin
nes edge traack [23]. In FFig 3‐11, thee notched dipole has 15%
% bandwidthh, whereas tthe same diipole with th
he notches filled has onlyy 12.4% band
dwidth. 3--111
Fig 3‐11 |S11| of the proposed dipole with and witthout notch and slot elem
ments he dipole no
otches are placed p
in thhe tips of th
he dipole arms to fine ttune the co
orrelation Th
beetween the required patttern pass baand performance and the |S11| bandd position. A
As shown in
n Fig 3‐11, th
he reflection coefficient magnitude performance
e of the notcched dipole element caase falls direectly over th
he desired ppass band off 850‐960 MHz, M
depicte d by the blu
ue band. W
Where the no
otches have been remooved, the reflection coefficient maggnitude perfformance occcurred somewhat lowerr in the bandd as depicted
d by the red band. The eeffect of notch depth beetween thesse two extre
emes is show
wn in Fig 3‐‐12 , where the |S11| chharacteristicc is more |S11|dB
seensitive to lo
onger length notches. Required band Notch
h removed Half-length
H
Full lengtth
3--112
oposed planaar dipole as the notch de
epth is adjustted Fig 3‐12 The |S11| of a pro
3..3.4
nar dipole inside radomee Plan
Faactors such as mutual coupling annd reflections make it difficult to analyse the dipole peerformance when in the array. A
A model of the dipole is i created too analyse th
he single ellement performance of the final c onstruction. The mode
el is shown in Fig 3‐13
3; with a w
waveguide po
ort placed accross the sloot line to fee
eds the dipo
ole independdently. The radome, su
ubstrate and part of the ffeed networrk are also included to provide realisttic loading. Su
ubstrate is 0.125 mm thicck PET with εr = 2.8 and tanδ = 0.00112
W
Waveguide p
port placed aacross slot lin
ne Radome with
R
h εr = 4.2 tanδ
δ = 0.012 Fig 3‐13 Single element planar dipole with radome loading Reeferring to C
Chapter 2, a feed netwo rk port impe
edance of 12
29Ω is assum
med. The im
mpedance reesults given in Fig 3‐14 show s
that thhe mean fee
ed point imp
pedance of tthe dipole is close to 1229Ω, Smith C
Chart is norm
malized to th is value. The
e |S11| of the radome lo aded planar dipole is sh
hown in Fig 3
3‐15, where there is a m
minimum reflection coeffficient magn itude of >10
0 dB with 100% bandwidtth. 3--113
903MHz//1
129Ω d point imped
dance of a pllanar dipole mounted insside 76mm ddiameter rad
dome Fig 3‐14 Feed
0
‐2
‐4
|S11|dB
‐6
‐8
‐10
‐12
‐14
‐16
‐18
‐20
7550
800
850
900
950
100
00
Freq
quency (M
MHz)
Fig 3‐15 Normalised plan
nar dipole reeflection coe
efficient mag
gnitude imp edance (refe
erence is 1229Ω) Th
he E‐field current distribution was annalysed at th
he extremitie
es of the des ired frequen
ncy band, an
nd the resultts are shown
n in Fig 3‐16 and Fig 3‐17
7. At the lo
owest frequeency of 850 M
MHz, the 3--114
onger curren
nt path arou
und the outter edges of o the dipole
e can be seeen. At the higher lo
frequency of 960 MHz, th
he current aactivity is mo
ore confined
d to the geoometric lengtth of the diipole. This characteristicc highlights hhow the dipole geometrry has been modified to
o present th
he broader reesponse. Fig 3‐16 E field
d current disstribution at lowest frequ
uency= 850M
MHz d current disstribution at the highest frequency = 960MHz Fig 3‐17 E field
3--115
3..3.5
Dipo
ole performa
ance in the aarray W
When antenn
na elements form an arraay, mutual ccoupling of the adjacent elements efffectively lo
owers the resonant frequ
uency. As aa consequen
nce, the plan
nar dipoles m
must be shorrtened in leength when arranged in the final coollinear arraay to mainta
ain the coveerage of the
e desired frequency ban
nd. Th
he proposed
d dipole design has a slootline feed, hence there
e are no exttra balancingg devices reequired. Thiss type of feed insures eequal curren
nt distributio
on leading tto efficient radiation from the arrayy. The propo
osed array d ipoles are de
esigned to su
uit their posiition in the a
array and ments at the far ends of the array are differennt than thosse in the ass such, the dipole elem
m
middle. This is due to the inside di pole arm be
eing loaded by the prevvious dipole and the ou
utside dipolee arm expossed to free space conditions for ele
ements at thhe end of th
he array. In
ncreasing thee length of the t central cconductive tracks t
of the
e feed netwoork exposes the end diipoles to sim
milar loading conditions a s the inner e
elements. 3..3.6
Radiation performance of thhe planar dip
pole array elements Raadiation properties of th
he dipole eleement and fe
eed network design are tthe main controls on paattern shapee in an array situation. A
A squinted d
dipole pattern will influennce the squiint in the arrray; and inccorrectly phased elementts will introd
duce major p
pattern disto rtion indepe
endent of th
he dipole peerformance. If the dipolee does not choke c
curren
nts sufficienttly, the out of phase cu
urrents beco
ome dominan
nt and greatlly influence tthe radiation
n pattern disttortion. Th
he radiation
n patterns of the propoosed planar dipole arrayy elements a seen in Fig F 3‐18a ellevation and The azimuth pattern in Fig 3‐18(b) shows a maximum distoortion of 1.6 dB peak o peak. This distortion is typical of a planar collin
near array element with a width app
proaching to
a λ/4. The ind
dividual arrayy element gaain is approxximately 3.2 d
dBi. 3--116
(a) (bb)
Fig 3‐18 The ttypical (a) ele
evation and (b) azimuth radiation pattern of dipoole 3..4
Sum
mmary
W
Within an eleectromagnetic modellingg environment, the operrational paraameters of the novel pllanar dipolee element proposed p
fo r use in a collinear arrray have bbeen analyse
ed. The trransition fro
om a cylindrical dipole to a planaar dipole wa
as discussedd, and then
n further deeveloped to allow for use
e in the colli near array p
proposed in tthis thesis. It t was shown how this pllanar dipole is related to
o coaxially cconstructed dipoles in te
erms of currrent chokingg, and an exxample of th
he proposed
d dipole withh its balance
ed feed displays evenly distributed currents. Th
he planar dip
pole structurre is also effeective at cho
oking off surface currentss. Th
he proposed
d planar dipole is designned to meett the broad bandwidth and omnidirectional paattern speciifications ou
utlined in CChapter 1 by b implemen
nting augmeentation tecchniques, naamely slots aand notches.. The slots a re used to co
ompensate ffor reactancee associated with the close proximitty of the dip
poles to thee edge of the
e slotline feed [23]. Thhe slots augm
ment the ove directivi ty and reducce azimuth rripple, and too lesser exte
ent assist raadiation patttern to impro
in
n impedancee matching. The propoosed planar dipole with notched tipps was show
wn to be caapable of a 1
15% bandwid
dth for VSW R < 2:1, whe
ereas the con
nventional pplanar dipole shape is on
nly capable of a 12% bandwidth. It was show
wn that this planar dipoole design meets m
the baandwidth required for base station aapplication, and is simila
ar to a cylinddrical dipole in terms off impedance and radiatio
on pattern p erformance.. A study off the currentt distribution
n showed th
hat the dipole is active a
at its extrem
mities for the
e lower frequency and m
more active closer to th
he feed point at the higher h
frequuencies. In
ndividual pla
anar collineaar element sections diisplayed an aazimuth ripple of 1.6 dB aand a maxim
mum gain of 3
3.2 dBi.
3--117
C
Chapter 4
Robus
st Coplanarr wave guid
de to micro
ostrip transsition
4.1
Introd
duction
A passively co
oupled “Cop
planar wavegguide (CPW))” to Microsstrip transitioon with DC isolation op
perating in the band from
m 850 – 960 MHz is pressented. The transition is specifically designed to
o solve a meechanical stress fracturee risk associated with the t soft soldder attachm
ment of a co
oaxial cable to the “Coplanar wavegguide (CPW)”” transmissio
on line feed of an array antenna. Th
he transition
n provides a low loss con nection to th
he “Coplanar waveguidee (CPW)” fed antenna arrray, with a robust launcching platforrm for a microstrip line/ccoaxial connnector interfa
ace. The ad
dded advanttage of passivve coupling iis a reduced number of ssolder conneections and a
a gradual trransition to tthe “Coplana
ar waveguidee (CPW)”. Siignificantly, tthis transitioon is unique in that it is totally DC issolated, leading to improoved PIM perrformance. n Section 4.2,, a literature
e review detaails research relating to b
broad band ““Coplanar wa
aveguide In
(C
CPW)” to Microstrip tran
nsitions, all oof which are
e fabricated on the samee dielectric layer and ussing a comm
mon ground p
plane or a coommon source connection. In Sectiion 4.3 the p
proposed paassive coupling circuit is introduced. The unique
e performancce capabilitiees are discusssed. The prroposed tran
nsition is un
nique as it p rovides an impedance transformatio
t
on, DC isola
ation and en
nhanced PIM
M protection by using passive coupling. Secction 4.4 deetails the simulated peerformance of the transition, along w
with a param
metric investigation of keyy structural features. Seection 4.5 exxplains the p
physical test setup requiired to accurrately characcterise the p
proposed trransition. Measured results are pressented, and compared to
o the simulaations. In Section 4.6 th
he assembly of the tran
nsition is exxplained, alo
ong with an analysis of tolerance effects e
in co
onstruction. A summary of the invesstigations is p
provided in S
Section 4.7. 4--118
4.2
ature revie
ew of bro
oad band
d “Coplan
nar waveg
guide (CP
PW)” to
Litera
micro
ostrip transsitions
Th
he vast majo
ority studies into “Coplannar waveguid
de (CPW)” to
o Microstrip ttransitions a
are found to
o be based on o uniform thickness hoomogenous substrates. These trannsitions are of great in
nterest to th
he research community as they allow for a low loss conttrolled transsition for m
module intercconnections up to 40 GHzz. “C
Coplanar waaveguide (CP
PW)” can poose difficultie
es when inte
erfacing withh a coaxial cable c
via co
onventional connectors. A close pitcch track spaccing in most cases makess it difficult tto attach a connector or o cable with
hout the riskk of short circuit. A low
w loss transittion to non‐‐coplanar trransmission lline can be a better optioon for mechaanical and electrical interrfaces. he design sh
hown in Fig 4‐1 is a traansition with
h strong elecctromagneticc field coupling. The Th
trransition has capacitive surface patchhes to couple
e the active llines of the ““Coplanar wa
aveguide (C
CPW)” and th
he Microstrip [45]. The ground plan
ne is commo
on to both ““Coplanar wa
aveguide (C
CPW)” and M
Microstrip. T
This transitio n has a relattively low reflection coeffficient magn
nitude (< ‐110 dB) from 3
3.2 GHz to 11
1.2 GHz, andd a transmisssion loss of 0.2 dB 4--119
(a)
(b)
Microstrip
(c)
CPW
W interface
interfacce
a) Miccrostrip inte
erface b) Tran
nsition area
c) CPW
W interface
Fig 4‐1 Capaciitive coupled
d CPW to miccrostrip transition [36]. Th
he “Coplanaar waveguid
de (CPW)” to microsttrip transitio
on shown in Fig 4‐2 has an ellectromagneetically coupled ground and a com
mmon source
e track. TThis transitio
on has a GHz, with a reflection co
oefficient trransmission aattenuation of 0.3 dB att 5 GHz and 1 dB at 7.8 G
m
magnitude off 15 dB from
m 2.8 ‐ 7.5 G
GHz [46] and is proposed
d for use in flip chip “m
monolithic m
microwave inttegrated circcuit (MMIC)”” applicationss. 4--120
(c)
(b)
(a))
Micro
ostrip
CPW
C
interface
i
Interfface
Section vieews
a) Microstrip Interfacee b) Transition area c) CPW intterface nsition with ccommon souurce track and electromagnetically cooupled groun
nd [37] Fig 4‐2 A tran
In
n Fig 4‐3, a transition design with sstepped grou
unded CPW (GCPW) secttions is show
wn which also has a common micro
ostrip and “CCoplanar waaveguide (CP
PW)” signal cconductor [4
47]. The nterface is a a convention
nal microstrrip line, and
d the transition is madde through a GCPW in
trransformer to t “Coplanarr waveguidee (CPW)”. Th
he presented
d performannce is not clear, the reeflection coeefficient maggnitude is aapproximately 10 dB, th
he insertion loss < 1 dB
B over a baandwidth of 38%. 4--121
(a) (b)
(c)
(
Micro
o
CP
PW
strip
in
nterface
a) Microstrip in
nterface Section views
b) GCPW Transsition area c) CPW interface Fig 4‐3 A tran
nsition with ccommon souurce track and electromagnetically cooupled groun
nd [38] ound “Coplanar wavegu ide (CPW)” to Microstrip transition is shown in
n Fig 4‐4. A tapered gro
c
trracks is well known bro
oad‐banding technique. This transitiion is an Taapering of conductor im
mprovement over the previous desiign as it hass a modified
d ground traack and source track in
ntroducing a more gradua
al transition from one mode to the o
other [48]. 4--122
(a)
(b)
(c)
Microstrip
CPW
interface
interface
Section views
a) Microstrip interface b) GCPW Transition area c) CPW interface Fig 4‐4 A transition similar to Fig 4‐2 but with tapered conductors [39] The transition depicted in Fig 4‐3 uses electromagnetic coupling for the source of the Microstrip line. The “Coplanar waveguide (CPW)” source track is at ground potential at the transition. The Microstrip coupling has fan stub and the “Coplanar waveguide (CPW)” has a return to ground [49]. The reflection coefficient magnitude is 10 dB with insertion loss of 0.3 dB at 5 GHz and 1 dB at 7.5 GHz. This transition uses an RT/Duroid substrate with a permittivity of 10.2 to reduce the physical size of the transition. As the source track of the Microstrip is DC isolated from the “Coplanar waveguide (CPW)”, this type of transition avoids the need for DC blocking capacitors in some applications. The fan stub meshing with the fan 4-123
haped apertture provide
es the graddual transition from “C
Coplanar waaveguide (CPW)” to sh
M
Microstrip ovver a broad band exhibbiting similar results to the previouus transition
n with a taapered geom
metry. (a)
(b)
((c)
Microsttrip
CPW
interfacce
Section vieews
a Microstriip interface
a)
b Transition area b)
c CPW inte
c)
erface Fig 4‐5 An eleectromagnetically coupleed CPW to M
Microstrip transition, com
mmon ground
d [40] he “Coplanar waveguide
e (CPW)” to Microstrip transitions sh
hown in Fig 44‐6 are exam
mples of: Th
(aa) a straight‐‐stub transitiion with a b andwidth off 20%; and (b) a radial sttub transitio
on with a baandwidth of 25%. Both transitions have a reflection coefficcient magnit ude of 17 dB and an in
nsertion losss of 0.18 dB
B in the ra nge near 85 to 110 GHz G [50]. TThe transitio
ons have ellectromagneetically coupled ground pplane, and sh
hare the source conductoor. The mettal tracks osited on a si licon wafer ssubstrate. arre gold flasheed and depo
4--124
Ground
Ground (a) CPW Interface
(b)
Microstrip Interface
Fig 4‐6 These are examples of state of the art performance coupler circuits [50] (a) straight‐stub transition (b) radial‐stub transition A very simple transition is shown in Fig 4‐7, where both the “Coplanar waveguide (CPW)” and Microstrip have continuity in ground and source tracks [51]. Via are used to complete the connection between the “Coplanar waveguide (CPW)” ground tracks and the Microstrip ground. The source tracks are directly connected. As there are no resonant circuits involved, this coupler works over a very large bandwidth (0.04 GHz up to 25 GHz). The insertion loss is typically around 1.5 dB at 20 GHz and the reflection coefficient magnitude is 15 dB Values of insertion loss and reflection coefficient magnitude are not stated at the lower frequencies. Microstrip Finite ground CPW
Fig 4‐7 Example of directly connected CPW to Microstrip transition [42] 4-125
Perfformance go
oals for the ““Coplanar wa
aveguide (CP
PW)” to Mic rostrip transsition 4.2.1
he desired coupler will m
make a transsition from aa flexible PCB collinear aarray to a substantial Th
co
oaxial cable and low PIM P
connecttor. As pe
er other cou
uplers descrribed, the proposed p
trransition willl be passive (i.e. has no DC continuitty) and has a
a resonant sttructure to ffine tune th
he coupler paass band perrformance. TThe following specificatio
ons are requuired: pler Taable 4‐1 Thee performancce specificat ions for the passive coup
Baandwidth: In
nsertion lo
oss: 850 to 960 MHz (minimum)
0.25
5 dB Reeflection co
oefficient > 15
5 dB m
magnitude: ≥ ‐140 dBc mini mum, ≥ ‐150
0dBc ideal w
with 7/16 DIN
N PIIM: nector conn
Po
ower: 100W maximum
m Arrray port: CPW
W (approx. Zo = 69 Ω) Eq
quipment 7/16
6” DIN con nector via RG142 double shieldedd po
ort: cablle. 4.3
The structure
s
and
a
operattion of the passive co
oupling circ
cuit
Th
his section in
ntroduces a passive cou pling transition from Microstrip to ““Coplanar wa
aveguide (C
CPW)” lines, where a heavy duty cconnector or cable can be attacheed to a robu
ust, rigid M
Microstrip PC
CB. The de
esign is uniqque, as both
h the source and grounnd of the “Coplanar w
waveguide (C
CPW)” transm
mission line are effectivvely DC isola
ated from thhe coaxial co
onnector asssembly. Thee shape of the t “Coplanaar waveguide (CPW)” co
oupling padss is the resu
ult of the an
nalysis of [46
6], and othe
er literature on “Coplanaar waveguide
e (CPW)” to microstrip ccouplers. 4--126
uired in orde
er to couple the ground currents Sttrong capacitive couplingg was foundd to be requ
efffectively fro
om the arrayy to the miccrostrip grou
und. The geometry of thhe coupling probe is baased on thee analysis of [45]. The dimensions of the final coupler weere restricted by the avvailable width of the ra
adome. It iss well know
wn that smo
ooth transitioons are req
quired to en
nhance the b
bandwidth achieved hen ce the curve
ed geometry.. TThere is no solder proccessing requuired on the
e “Coplanar waveguide (CPW)” side of the trransition, with the coaxial cable onnly attached to the morre robust M
Microstrip sid
de of the trransition. In the specificc application for the “Coplanar wave
eguide (CPW))” fed colline
ear array prroposed in this thesis, DC continuityy is specificallly avoided to improve t he PIM performance off the entiree assembly. A schemaatic of the proposed “C
Coplanar waaveguide (C
CPW)” to M
Microstrip traansition is de
epicted in Figg 4‐8. The transition consists of twoo substrates: the 125 μm
m PET flexible substratte (εr = 2.88) of the “C
Coplanar waveguide (CPPW)” colline
ear array prresented in tthis thesis; and an Arlon AD250C [52
2] substrate ((εr = 2.5 and tanδ = 0.0014). This riggid Arlon fibreglass composite maaterial is mo
ore suitable to supportt a coaxial cable or co
onnector intterface (via a Microstripp line) than the flexible
e PET, and also provide
es stable ellectrical prop
perties, low PIM and higgh operatingg temperaturre allowing ffor high temperature so
oldering. Th
he two materials are laaminated together with no direct eelectrical connection beetween the ttwo transmisssion lines. CPW
W cou
upling CPW ground coupling Microstriip
source
Heeavy duty
CPW
W
con
nnector
inteerfa
Microstrip probe length Arlon
AD250
125 μm PE
ET
flexible
Microstrip
M
grround
a) CPW
W interface W coupling a
area ground
b) CPW
c) CPW
W coupling a
area source
d) Microstrip interface Se
ection viewss Fig 4‐8 Schem
matic layout o
of the propoosed passive CPW to Microstrip transsition 4--127
he broad ban
nd performa
ance of this ttransition is ffacilitated byy a gradual cchange in the
e ground Th
pllane geomettries of the M
Microstrip linne and “Coplanar wavegu
uide (CPW)”.. The tapering of the grround planes gradually changes c
thee “Coplanar waveguide (CPW)” (
moddes to the Microstrip M
m
modes over th
he λ/4 wave
elength of th e transition.. The optimu
um slot deptth in this transition is 133.75mm. Thiis tapered sh
hape maximiises the bandwidth durin
ng the transsition from “Coplanar w
waveguide (CPW)” to Miccrostrip as seeen in the lo
ower plan vie
ew of Fig 4‐99. The flarin
ng of the “C
Coplanar wavveguide (CPW
W)” and Miccrostrip grou
unds are overlapped to foorm low Q capacitive co
oupling platees. The shap
pe of the traansition plate
es was optim
mised using [40]. The o
optimum m
microstrip pro
obe length iss 32.5mm fo r a “Coplanaar waveguide
e (CPW)” proobe length o
of 38mm. Th
he slot lengtth and micro
ostrip probe length have been selected based onn the highestt average Reeflection coeefficient maggnitude respoonse and the
e minimum loss through the transitio
on. Microstrip ground Slot de
epth CPW
1.6mm
m
Arlon
λ/4 couplinng AD250
rigid substrate
s
εr = 2.5
Fig 4‐9 The layyout of the ccoupling trannsition bottom layer show
wing couplinng pad alignm
ment he specially designed ed
dge connect or is presen
nted in Fig 4‐10 allows aa coaxial cab
ble to be Th
so
oldered direcctly to the A
Arlon Microsstrip line with more subsstantial mec hanical stren
ngth and ellectrical connection than
n when direcctly soldered
d to a “Coplanar waveguuide (CPW)””. Unlike m
most microw
wave connectors which are fabricatted with ind
dividual diellectric supports and seeparate inneer conducto
or componennts, this edge connecto
or was speccifically designed to in
nclude the atttaching cab
bles dielectriic and inner conductor as a part of thhe assemblyy thereby reeducing the manufacturring parts coount and co
ost. The co
onnector moounting grooves are deesigned to slide s
onto th
he Arlon AD
D250 substrate [52], and
d precisely s et the heigh
ht of the 4--128
Microstrip atttached cable such that the inner coonductor passses through at the surfaace of the M
trrack. The brraid of the cable can be securely soldered into the body of the connector. The Arrlon PCB is recessed to
o allow for tthe length of o the connector to be inset into the PCB, in
ncreasing thee alignment a
accuracy andd mechanical strength. (a) (b
b) (c) Fig 4‐10 low p
pim edge con
nnector appllication (aa) Low PIM eedge connector model (bb) a fabricated edge con
nnector (c) thhe applicatio
on of the co
onnector to tthe passive ttransition to the collinear array 4.4
Broad
dband passsive transiition perforrmance
waveguide ( CPW)” to microstrip tran
nsitions and couplers are
e used to Traditionallyy “Coplanar w
provide low loss interconnects for m
microwave m
modules. This proposed ccoupler provvides this mplete DC issolation, as well as solving a mecchanical issu
ue when performancee with com
interfacing ““Coplanar wa
aveguide (CPPW)” using a cable or con
nnector. he passive cooupler in the
e back‐to‐ba
ack configuraation was crreated to A parametric model of th
ptimise the geometry. This is a well
T
l‐known metthod to charracterise tra nsitions thatt are not op
500 Ohms as shown in [5
53]. The traansition structure in Fig 4‐11 was ssimulated using CST ellectromagneetic software
e [40]. Thee back‐to‐baack test fixture comprissed of two identical M
Microstrip to “Coplanar waveguide w
(CCPW)” transsitions conne
ected as a m
mirror image about a seection of “Co
oplanar wave
eguide (CPW
W)” equivalen
nt to that ussed in the coollinear arrayy feed of seection Chapter 2.5. 4--129
Line of sym
mmetry pler test fixtuure Fig 4‐11 Back to back coup
A parametric sweep of the
e slot depth located betw
ween the co
oupling platees as shown iin Fig 4‐9 w
was conducteed. Minimum insertion loss is achiieved over a a wide bandd for a slot depth of 133.75mm is o
observed from Fig 4‐12. This configu
uration results in an inseertion loss o
of 0.1‐0.2 dB
B and a min
nimum and reflection r
cooefficient maagnitude 29 dB. From thhese results,, the slot deepth is seen to have a m
minor influencce on the inssertion loss, but greatly aaffects the rreflection co
oefficient maagnitude. Th
he effect asssociated with
h the Microsttrip probe le
ength (as see
en in Fig 4‐8) is shown in Fig 4‐13. Th
his is a critical component in the m
matching an
nd efficiencyy of the couupling coefficients. A M
Microstrip pro
obe length of o 32.5mm pprovides a minimum m
inssertion and rreflection co
oefficient m
magnitude values. 4--130
ons in the “Coplanar w
waveguide (CPW)” (
prob
be length (ddepicted in Fig 4‐8) SSmall variatio
dual changess in insertionn and large changes in reflection r
cooefficient ma
agnitude, geenerate grad
An
n optimal value v
of 38
8 mm prod uces a min
nimum inserrtion and rreflection co
oefficient m
magnitude of 0.2 dB and 29dB respecctively when combined w
with a Micro strip probe length of 322.5mm and sslot depth off 13.75mm.
Fig 4‐12 Backk to back coupling test wiith parametrric slot depth
h 4--131
eep of the paassive couple
er with vario
ous Microstriip probe lenggths Fig 4‐13 A parametric swe
4.5
Back
k to back trransition m
measureme
ent
he back‐to‐b
back transitio
on was meassured from 8
850 ‐ 960 MH
Hz, with two 100 mm lon
ng coaxial Th
caables used to connect to an Agi lent 5070B network analyser to characterise
e the S‐
paarameters. A A compariso
on to the moodelled resu
ults (with co
oaxial cable ssections included) is prrovided in Fiig 4‐18. By halving the iinsertion losss measured in the back‐‐to‐back tran
nsition, a single passive coupler exh
hibited an inssertion loss o
of around 0.3
3 dB, with a reflection co
oefficient m
magnitude greater than 1
15 dB. The w
worst case m
measured inssertion loss iis 0.4 dB at 8
850 MHz Th
he measureed reflection
n coefficien t magnitude response is differennt to the modelled m
reesponse, alth
hough it remains within tthe specificaation outlined
d in Section 4.1.1. The increased lo
oss most likelly caused by the PCB reggistration and
d alignment accuracy in tthe fabricatio
on of the prrototype passsive couplerr. 4--132
easured S pa rameters of transition ch
haracterised back to back Fig 4‐14 Modeelled and me
he performaance of the
e coupler iss comparable to simila
ar transitionns presented
d in the Th
litterature. A comparison is provided in Table 4‐1
1. However, this couple r also has th
he added ad
dvantage of total DC isollation, allow ing the transsition to funcction with arrbitrary subsstrates at eaach interfacee whilst main
ntaining low insertion losss and high reflection coeefficient maggnitude. 4--133
mparison the
e proposed ppassive couplling transitio
on to referennces (measurred back‐
Taable 4‐2 Com
to
o‐back resultts) Proposed [50] [51] [4
46] 85
50‐960 85‐11
10 0.04‐25 2.8‐7.5 MHz GHz GHz GHz GCPW ‐> CPW microstrip microsstrip microstrip < 0
0.5 dB 0.18dB
B 15 dB 17 dB
15 dB 15
5 dB Ground Passive Passive
e Direct Direct tracks coupled couple
ed connection onnection co
Direct Pa
assive connection co
oupled co
oupler Frequency Transition Insertion loss CP
PW ‐> ‐> < 1.5 dB CP
PW ‐> microstrip 1 d
dB @20 GHz Reflection coefficient magnitude Source tracks Passive coupled Direct connecctio
n 4.6
The passive
p
coupling asse
embly
Th
he passive trransition sho
own in Fig 4‐115 comprises of the follo
owing compoonents: 
Microstrip
p line passive
e coupling ci rcuit: Arlon A
AD250C [52] 
Alignmentt/compressio
on plates: Poolycarbonate
e 
Low PIM eedge connecctor: Silver pplated brass

RG 142 caable 
Non‐cond
ductive fasten
ners: Nylon

Low PIM 7/16 DIN Jacck If DC groundin
ng of the coa
axial interfacce is required
d, a short cirrcuited paralllel stub can be fitted to
o the low PIM
M connector as shown iin Fig 4‐15. This option may be reqquired to pro
otect the eq
quipment fro
om lightningg or other staatic charge sources, such
h as areas deesignated inttrinsically 4--134
petro‐chemiccal sites. Th is proposed coupler provvides this peerformance a
as well as saafe areas in p
co
omplete DC isolation to
o solve a m echanical issue when in
nterfacing ““Coplanar wa
aveguide (C
CPW)” using a cable or co
onnector. R
RG 142
3mm polycaarbonate
125 um plates
PE
ET C
Cable
R
RG 142 cablee stub
Arlon AD250
RG 142 cable
Fasteeners
Loow PIM 7//16 DIN o the PET CPW feed PCB Fig 4‐15 Side vview of passiive transitionn attached to
Th
he Arlon Miccrostrip circu
uit is secureed to the PETT collinear array substraate by polyca
arbonate alignment/com
mpression p
plates and noonconductive
e fasteners. The fasteneer holes in th
he plates arre aligned prrecisely to th
hose on the PET array substrate, keying the aliggnment and ensuring co
onsistent cou
upling performance. To test the imp
pact of misalignment of the substratte layers, a parametric sweep of th
he longitudinnal alignmen
nt was cond
ducted. Fig44‐16 depicts the test w
monittoring the reflection r
sttructure useed. The substrates weree offset by ± 3 mm whilst co
oefficient maagnitude the
e passive couupling transittion. The ressults of this ssimulation arre shown in
n Fig4‐17, wh
here an offse
et of ± 3 mm
m changes th
he insertion loss value byy a maximum
m of 0.15 dB
B. The reflecction coefficient magnituude exhibits more significcant change.. 4--135
+/- 3mm
m
Fig4‐16 Passivve coupler alignment tesst configuration metric sweep
p of the passsive coupler ssubstrate alignment Fig4‐17 Param
he PIM performance and PIP perforrmance resu
ults for the passive p
couppling transitiion were Th
teested as partt of the complete collineear array, an
nd are detaile
ed in Chapteer 5. As the low PIM co
onnector is h
hand crafted from raw b rass, it has o
only achieved
d a PIM valuue of ‐148dBcc. This is exxpected to faall well below
w ‐150 dBc inn the final prrofessionally machined aand finished product. 4--136
4.7
mary
Summ
he majority of “Coplan
nar waveguidde (CPW)” to microstrip transitionns presented
d in the Th
litterature are aimed at fre
equencies abbove 1 GHz ffor MMIC (“m
monolithic m
microwave in
ntegrated circuit “) appllications. In Section 4.22 the literatu
ure has show
wn that the ttransition to
opologies ncountered had DC con
ntinuity in a t least one of the cond
ductors. In Section 4.3 a novel en
“C
Coplanar waveguide (CP
PW)” to micrrostrip transsition is presented. The proposed transition w
was implemented in orde
er to increasse the mechanical streng
gth of the ccable connecction and elliminate sold
der processes. This trransition hass low loss, low PIM, inncidental im
mpedance m
matching, and
d is totally DC D isolated ffrom the inp
put to outpu
ut transmissiion lines. It provides su
ufficient mecchanical isola
ation to enabble a robust coaxial cable
e connectionn. In
n Section 4..4 the prop
posed coupl er performaance is determined ussing electrom
magnetic simulation. Th
he insertion and reflectiion coefficient magnitud
de performa nce is explo
ored with e (CPW)” reespect to keyy parameters that affectt the coupling between tthe “Coplanaar waveguide
an
nd microstriip transmission lines. A
As the outp
put impedance of the ““Coplanar wa
aveguide (C
CPW)” is different to the VNA (““Vector Nettwork Analyyser”) meas uring equip
pment, a co
ommonly ussed back‐to‐back measuurement technique was explained iin Section 4.5. 4 The paassive coupling transition performannce was then verified byy testing a rrealized prottotype of th
he back‐to‐baack structure
e, with reasoonable congrruence to the simulated results. An insertion an
nd reflection
n coefficient magnitude of < 0.5 dB and > 15 dB
B respectivelly was achieved over th
he frequencyy band of 850
0‐960 MHz.
Th
he transition
n assembly w
was detailed in Section 4.6, which con
nsists primarrily of a flexib
ble 0.125 m
mm “Polyethyylene terephthalate (PETT)” PCB lamin
nated with th
he more subbstantial Arlo
on AD250 [552] rigid miccrostrip subsstrate. The lamination and a alignme
ent of the pllanar substrates was in
nvestigated tto determine
e the effect oof manufactturing tolerance. The cooupling structure was sh
hown to be relatively in
nsensitive too small offse
ets of ± 3 mm. m The sollder‐less asssembly is exxpected to im
mprove the PIM perform
mance signifiicantly. With
h complete ppassive coup
pling, the trransition circcuit is capable of couplinng efficientlyy through disssimilar subsstrate dielectrics and th
hicknesses. This allows the collinea r array desccribed in thiss thesis fabrricated on 0..125 mm th
hick “Polyeth
hylene tereph
hthalate (PETT) “to be eassily cabled du
uring the asssembly proce
ess. 4--137
Chapter 5
“Copla
anar waveg
guide(CPW
W)” fed omnidirection
nal collinea
ar array
with fre
equency independentt beam dire
ection
In
n this chapteer the plana
ar feed netw
work, dipole elements and passive ccoupler com
mponents in
ntroduced in
n previous chapters c
aree integrated to form th
he proposedd omnidirecttional six ellement “cop
planar wavegguide (CPW))” fed collinear array. The T array haas unique attributes w
which expand
d the possible uses for aa relatively low cost PC
CB based anttennas, with
h pattern peerformance comparable to much moore complex array designs. Taable 1‐4 thee characterisstics of a sellection of sttate‐of‐the‐art omnidirecctional antennas are co
ompared wiith the advantages andd disadvanttages for ea
ach technoloogy, and how they co
ompare to th
he characteriistics of the pproposed array in TABLE 1‐1. In
n Section 5.1 the plana
ar structuredd feed netw
work, dipole elements aand passive coupler co
omponents aare reviewed
d, and then united to fo
orm an omniidirectional ccollinear array. The peerformance specification
ns for the arrray are also rrevisited in Section 5.2. Seection 5.3 deetails the sim
mulated and measured aarray perform
mance. Simuulated and m
measured reeflection co
oefficient magnitude reesults are compared. Base stattion array antenna peerformance is ultimatelyy assessed bby the analyssis of the rad
diation patteern. Factorss such as beeam tilt, gain and side lobe level aare the maiin characteristic measurres discussed in this seection. Th
he power handling pe
erformance is anotherr important factor in base statio
on array ap
pplications. Section 5.4 evaluates thhe maximum
m power hand
dling of the aarray. The behaviour off the array w
when PIM (Pa
assive Interm
modulation) p
products are generated uusing an exciitation of tw
wo carriers, and the PIP
P (Peak Instaantaneous Power) P
as a result of thhe combinattion of a nu
umber of traansmitters into the one aantenna are aalso analysed
d. Seection 5.5 exxplores the m
mechanical li mits that base station arrrays are req uired to with
hstand in th
heir installation environm
ment. The calculated wind w
survival ratings andd projected area are 5--138
ong with the
e design connsiderations put in place to protect tthe array fro
om these prresented, alo
sttresses. A discussion o
on the metho
ods used to provide a prre‐calculated
d beam tilt baased on pha
ase offset n base station arrays is presented in Section 5.6. Modelling rresults for a proposed arrray with in
a fixed 4 degrree down tiltt are provideed, and the im
mpact this m
modification has on the rreflection oefficient maagnitude perrformance is analysed. A
A comparison
n is made beetween the tilted and co
un
n‐tilted arrayy in terms o
of reflection coefficient m
magnitude, pattern shappe and realizzed gain. Finally, the ch
hapter is sum
mmarized in SSection 5.7.
5.1
Struc
cture of the
e omnidirec
ctional arra
ay with bea
am stability
ty
Th
he array feed
d network ba
ased on cop lanar transm
mission lines presented inn Chapter 2 u
utilised a ceentrally located “Coplanar waveguidde (CPW)” discontinuity,, which in coombination with the du
ual secondarry coplanar sstrip transm ission lines p
provides the phasing reqquired for ze
ero beam tillt functionality. The dipoles d
elem
ments used in the arra
ay are optim
mized for maximum m
baandwidth an
nd compatibility with thee feed netwo
ork in Chapte
er 3. The speecially design
ned cable co
onnection in
nterface desccribed in Chaapter 4 provvides a low P
PIM interfacce from the attached caable to the edge of the co
oupling PCB.. This couple
er alleviates one of the m
most commo
on causes off PIM in an array; the solder joints beetween the ffeeding coaxial cable andd the array itself. Th
he proposeed “coplanar waveguidde(CPW)” fe
ed omnidire
ectional arrray with frrequency in
ndependent beam direction is depictted in Fig 5‐1. The arrayy can be desscribed as tw
wo series feed dipole arrrays placed back to bacck (Part A an
nd Part B) fo
or an upper sub‐assemb
bly and a lo
ower sub‐asssembly with a lateral linne of symmetry. Each series fed aarray contain
ns dipole an
ntenna elem
ments with the geometrry shown in Fig 5‐9 and has an offsset cardioid azimuth raadiation patttern. There iss a 180 degrree phase offfset between each seriees fed array, resulting in
n near omnid
directional pa
atterns from each sub‐asssembly. Th
he sub‐assem
mblies are fe
ed back to bback with a line of symm
metry througgh the centrre axis of th
he array, creaating the gab
ble phase pr ofile require
ed for centre feeding andd zero beam tilt. This ph
hase profile results in th
he upper suub array asse
embly tilting
g in oppositi on to the lo
ower sub arrray assembly when freq
quency is vaaried. In this configuration, the vert
rtical omnidirectional m
main beam broadens b
ratther than abbruptly tiltin
ng (as it would in a serries feed arrray) with ch
hange in freq
quency. “Co
oplanar waveeguide (CPW
W)” is well kn
nown to provvide very go
ood cross 5--139
ave circuits. This is prim
marily due to
o the groundd tracks bloccking the taalk immunityy in microwa
su
urface wave propagation
n between acctive lines. TThis attribute
e of “Coplanaar waveguide
e (CPW)” haas been exp
ploited to prrovide a leveel of isolatio
on between the main ““Coplanar wa
aveguide (C
CPW)” feed currents c
and
d the seconddary parallell strip transm
mission line currents. The T array w
would not fun
nction correcctly if these ssignals were allowed to ccross paths uuncontrolled. To
o maximize tthe PIM and PIP perform
mance and to
o provide a gradual transsition from “Coplanar w
waveguide (C
CPW)” to microstrip moodes, the arrray is DC issolated by aa specially designed d
“C
Coplanar waveguide (CPW
W)” to microostrip transittion as show
wn in Fig 5‐1 The transittion uses caapacitive cou
upling to botth ground annd source, and hence this coupler efffectively DC
C isolates th
he array. Coupling is pro
ovided by meechanically laminating th
he specially m
modified interface of th
he array to th
he transition, providing aa low loss connection to the array. CPW
Part A
discontinu
uity
poweer
Part A
Part B
U
Upper
sub-arrray
CPPW to micrrostrip
traansition
Part B
Lower
L
sub-arrray
Fig 5‐1 A scheematic of the
e overall arraay componen
nts 5--140
5.2
ges of th
he six ele
ement “C
Coplanar waveguide
e (CPW)”
” array
Imag
components.
Th
he following images are sshowing the major comp
ponents of th
he array In
n Fig 5‐2 The array is pressented alonggside an existting array an
ntenna using the same ra
adome. CC8
806‐09 anten
nna 76m
mm diameterr radome High density ffoam spacerss 6 element CPW array anten na onents of the
e “Coplanar waveguide (CPW)” arrayy with an anttenna using tthe same Fig 5‐2 Compo
raadome In
n Fig 5‐3 Thee high density foam spaacers are sho
own either side s
of the aarray PCB. The T foam sp
pacers are deesigned to support the PPCB concenttrically in the
e radome to prevent asyymmetric lo
oading and possible azimuth pattern distortion. TThe passive ccoupler circuuit is shown attached to
o the array. 5--141
Microstrip to CPW transition
coupler High density foam support pads Active track coupling probe
Ground coupling pads Low PIM edge connector Fig 5‐3 The foam spacers used to support the array PCB In Fig 5‐4 The passive coupling circuit attached to the array sandwiched between two polycarbonate plates and secured with four fasteners. Fig 5‐4 The passive coupler circuit attached to the array 5-142
In Fig 5‐5 shows the full length of the array components. The cross section of the foam spacers is a “D” with a 12mm diameter core removed to prevent direct contact with the “Coplanar waveguide (CPW)” main line. Fig 5‐5 The full length of the “Coplanar waveguide (CPW)” array In figures 5‐6 to 5‐8 are the impedance matching and phase control elements CPW stepped impedance transformers Fig 5‐6 close‐up of “Coplanar waveguide (CPW)” transmission line 5-143
C
CPW and slot
tline CPW phase reference shunt
up of upper d
dipole Fig 5‐7 close‐u
Slot line
e ground retuurn stubs
Fig 5‐8 close‐u
up of top element 5..2.1
Arraay constructiion Th
he array is enclosed in a pull‐extruuded fiberglaass radome 76mm outeer diameter with an in
nternal diam
meter of 63m
mm. This raadome has been selecte
ed as it is uused for oth
her array 5--144
ntennas of aa similar freq
quency bandd. The arrayy has been fa
abricated onn a low cost,, flexible, an
0..125 mm PETT substrate cclad with 0.0035 mm cop
pper. The pa
assive coupliing circuit usses Arlon AD250 high temperature
t
PIM frienddly substrate
e. High den
nsity polyethhylene (HD‐P
PE) foam pacers are used u
to supp
port the arraay concentrically inside the radomee. A mountting tube sp
su
upport is theen fitted and
d secured too the radom
me. The arra
ay constructiion is detailed in Fig 5‐‐10 . The com
mplete techn
nical drawinggs of the array are given in Appendixx A through tto E. Dipolee offset spaccing Dipole
e gap controls the feed ppoint impeda
ance contro
ols the azimu
uth Dipole slots for ripplee and dipole ffeed pattern sh
hape point impedance Integraal choke
CPW
W GROUND
D Slot line feed Fig 5‐9 Planar dipole struccture 5--145
Fasteners
Feed cable interface
Polycarrbonate plates top and bottom
m support the
e coupling ci rcuit
cpw arrray main circcuit
Fig 5.3 Passivve coupling circuit laminaated to the array PCB Mountin
ng
flange
HD-P
PE foam
supp orts
R
Radome
Radiatoor entation of t he array Fig 5‐10 Cut aaway represe
5--146
5.3
Perfo
ormance expectation
ns
Taable 5‐1 is the performa
ance expecteed from thiss antenna ba
ased on low
w power applications. Th
his antenna was not exxpected to hhandle high power levels, as the m
materials use
ed in the prrototype aree intended fo
or temperatuure controlle
ed circuits. The specificattions are suiitable for lo
ow power (less than 100 Watts) cell eextension and low powerr telemetry aapplications. Taable 5‐1 Elecctrical speciffications (froom Chapter 1
1) Characcteristic Freque
ency band Specification 850 ‐ 960 M
MHz Gain 10.0 ± 1 dB
Bi over 10% b
bandwidth Side lo
obe suppresssion (typicaal) Azimu
uth patttern stability ‐10 dB ± 0.8 dB ovver 360˚ 10%
% bandwidth Tilt 0 ± 1 degre
ee over 10% bandwidth Powerr rating 100 Watts maximum at 26˚ C VSWR
R < 1.5:1 ove
er 10% bandw
width Conne
ector interface 7/16 Din jaack PIM ‐150 dBc, 2 x 30dBm carriers Peak Powerr Instantaneous 1 kW dBi is specifie
ed over a 10 % bandwidth. A typical 6
6 element coollinear arrayy vertical A gain of 10 d
of 10 to 12 degrees allow
ws for range of possible installation loocations witthout the beeam width o
neeed for accu
urate placem
ment. The ppower rating specified off 100 W enaables multiplexing of trransmitters with w a maxim
mum total ccombined po
ower of 70 Watts W
at low
w duty cycle. A peak in
nstantaneouss power rating of 1 kW enables coiincidental diigital symbools when the
e array is 5--147
CDMA or otther digital m
modulation techniques. Zero beam
m tilt is requ
uired for ussed with WC
brroad band performance
p
e independennt of operattional freque
ency within the band 850 ‐ 960 M
MHz. A 7/16 DIN connecttor is essenti al to obtain the associated PIM spec ification of ‐150 dBc. Six collinear radiator r
elem
ments are reequired with
h an elementt spacing of 0.881λ to achieve a
a Bi at centre frequency f
off 903 MHz referring r
to Eqn. 10. Ann element sp
pacing of gaain of 10 dB
0..881λ is used
d based on net dielectric loading offsets from sub
bstrate and rradome loading. Esstimation of number of e
elements reqquired | |
(10) = number o
of elements, = directivityy in dBi, = wavelengtth, = distance b
between elem
ments. 332
0.881
291
6 5.4
Arra
ay Antenna
a performa
ance
Th
he analysis of o the collinear array is conducted using Finite Difference Time Domain solver [440]. An acccurate mode
el of the anntenna dime
ensions has been consttructed, including all m
material electtrical propertties. The finnal design arttwork is exported to cre ate a prototype PCB. 5--148
his PCB is then t
assemb
bled into th e complete antenna arrray, includi ng the rado
ome and Th
m
mounting fixtures. The an
ntenna reflecction coefficient perform
mance is thenn tested in frree space ussing a Vectorr Network An
nalyser. Th
he simulated
d and measured results ffor the collin
near array wiith frequency
cy independe
ent beam diirection are sshown in Figg 5‐11. Both the numerical model an
nd the fabriccated prototyype have a reflection co
oefficient ma
agnitude in eexcess of 14 dB within th
he specified bandwidth ffrom 850 ome minor discrepanciess in the shape of the simulated and m
measured results can ‐ 9960 MHz, So
bee observed. These werre found to be due to tolerances t
in
n the manu al fabricatio
on of the |S11| co
oupling interrface. nitude for thhe proposed collinear Fig 5‐11 Meaasured and simulated refflection coeffficient magn
arrray 5.4.1
diation patte
ern measureement techn
nique Rad
M
Microwave arrrays of sma
all physical ssize and smaall aperture can be meaasured indoo
ors in an an
nechoic chamber. The proposed aarray is 1900
0mm long, approximate
a
ely 6 wavele
engths at 5--149
t aperture
e size (D) att the wavelength of 9003MHz. The range length is deterrmined by the op
peration: usiing Eqn. 11 Faar field (11)) Fo
or an array o
of this size tthe range lenngth given b
by (10) is mo
ore than 40 meters. Du
ue to this large range length, it is more practica l to use a fre
ee space ground reflectioon range as sshown in osed array. Inn this configu
uration, the ground refleection itself is used to Fig 5‐14 to teest the propo
mounted horrizontally at a height caancel out thee reflectionss on the rangge. The sourrce array is m
w
where the speecular groun
nd reflection and the incident wave ccoincide at ttest position creating a reflection qu
uiet zone. Th
he array undeer test is tilte
ed forward tto be on boree sight to the
e base of th
he source support. To m
measure the rradiation patttern of an a
array outdooors, it must b
be placed in
n relative freee space to prevent p
refleections that may impactt on the res ults. In this case the raange area is located on 20 hectares of cleared ffarm land at Tooborac, N
North of Me
elbourne, Au
own in Fig 5
5‐12 is the tyypical radiattion pattern logging equuipment and rotation ustralia. Sho
co
ontrol. Fig 5‐12 rangee test control and loggingg equipmentt Th
he antenna under test iss mounted oon an insulatted mount to
o measure tthe elevation
n pattern ass shown in Fig 5‐13 the a
antenna is m
mounted high
h above grou
und to preveent ground re
eflection, an
nd requires aan elevation platform to make adjusttments. 5--150
Fig 5‐13 settin
ng up antenn
na for testingg To
o measure vertically v
pollarized omniidirectional arrays, the array a
is placced horizontally on a no
on‐metallic p
pedestal abo
ove the grouund. To stim
mulate the array under teest, a signal source is ap
pplied. Due to reciprocity, the arrayy under test or the sourcce array can be either acctive (TX) orr passive (RX
X) in the systtem. The chooice of active and passivve antennas is normally b
based on ellimination off interferencce from or too other servvices. The arrray is rotateed on its cen
ntral axis hrough 360 d
degrees. This angular traansducer gen
nerates the X
X axis data. TThe amplitud
de of the th
reeceived levell output from
m the netwoork analyser is the Y axis data. The rradiation patttern is a reepresentation of the X axxis data withh respect to Y axis data. It can be p resented in either of tw
wo forms, Caartesian (in Fig 5‐15), or ppolar (in Fig 5
5‐16) 5--151
Sou
urce
X
Axis
Posiitioner
945
5.3mm 4.2m
X Axis
A angular ddata
Y Axiis relative levvel data
Fig 5‐14 Grou
und reflection range Fig 5‐15 Carteesian plot of array elevattion pattern
5--152
Fig 5‐16 Polar axis plot of elevation pattern shown in fig 5.15 The directivity of the array is calculated by integration of the main lobe. The gain of the array (in dBd) is simply the directivity minus the gain of a dipole in free space which is approximately 2.14 dBi. The total gain of the array is this value minus the reflection and resistive losses. Another method to evaluate gain which complies with TIA 329C standard is the gain by comparison technique. In this case, a known standard gain array is measured by the same process as the array under test and received levels compared. The array under test gain is determined relative to the measured value of the standard gain array. 5-153
5.4.2
ulated and m
measured raadiations pattterns Simu
he measureed and simu
ulated radiaation patterns for the collinear arrray with frrequency Th
in
ndependent beam directtion are pressented in Figg 5‐17 to Fig
g 5‐21. Thee results disp
play very go
ood similaritty with the exception of slightly higher h
level side lobes at 960 MH
Hz in the m
measured casse. The main
n lobe eleva tion pattern
n is on the ho
orizon withinn ± 1˚ over tthe 850 ‐ 9660 MHz bandwidth. The
e measured and simulated beam tiltt and beam w
width characteristics arre compared
d in Table 5‐2
2 and Table 55‐3. Fig 5‐17 Modeel and measu
ured elevatioon patterns 8
850 MHz 5--154
Fig 5‐18 Modeel and measu
ured elevatioon patterns 8
870 MHz Fig 5‐19 Modeel and measu
ured elevatioon patterns 9
910 MHz 5--155
Fig 5‐20 Modeel and measu
ured elevatioon patterns 9
930 MHz Fig 5‐21 Modeel and measu
ured elevatioon patterns 9
960 MHz 5--156
Taable 5‐2 Beam tilt characcteristics (in degrees) Frrequency (MHz) 850 870 910 930 960 M
Model (Array)) ‐1.0 ‐1.0 0.0 0.0 1.0 M
Measured(Arrray) ‐0.9 0.0 0.1 0.2 0.6 TTable 5‐3 Half power bea
am widths (i n degrees)
Frequen
ncy(MHz) 850 870 910
0 9330 Model (A
Array) 10.5
10.7 10.8
8
100.8 11.5 10.6
10.6 10.2
2
9.33 9.3 Measureed (Array) 960 5.4.3
Gain characteristics he gain of the prototyp
pe collinear array with frequency f
in
ndependent beam direction was Th
evvaluated usin
ng the gain b
by comparisoon technique
e, employing
g a pyramidaal horn stand
dard gain an
ntenna (700‐‐1200 MHz).. The loss inn the array w
was determin
ned by compparing the ca
alculated diirectivity and the measured gain oof the array, and the efficiency of the array was w thus caalculated. Table T
5‐4 su
ummarises tthe measure
ed pattern characteristi
c
cs of the prototype co
ollinear arrayy. Taable 5‐4 Patttern charactteristics for tthe six element array with frequencyy independe
ent beam diirection 5--157
Six element collinear array (measured) Frequency HH, Path = 45 m, AUT height = 4.2 m, Po = +10dBm test range 45 850 870 910 930 960 HPBW (°) 10.6 10.6
10.2
9.3 9.3
Beam Tilt (°) ‐0.9 0.0
0.1
0.2 0.6
9.05 9.52 9.9 10.2 9.7 9.0 9.4 9.9 10 9.5 0.05 0.12 0.0 0.2 0.2 95.3 95.7
96.0
97.9 98.9
(MHz) Directivity (dBi) AUT gain by comparison (dBi) Residual loss (dB) (coupling, resistive and VSWR losses) Efficiency (%) The peak measured gain is 10.2 dBi, and the results over the 850 – 960 MHz frequency band are within the 10 ± 1 dBi tolerance specified in Table 5.1. The realized gain from the CST simulation is plotted in Fig 5‐22 and compared to the measured gain results. The results show good congruence, with a maximum difference of only 0.4 dB. The variation could be attributed to accuracy tolerances in the measurement technique. 5-158
12
11
realised gain dBi
10
9
Measuredd gain
8
Modelled gain
7
6
5
850 8660 870 880 890 900 910 9920 930 940 950 960
Frequency(M
MHz)
nd modelled realized gain for the six element arraay Fig 5‐22 Meassured gain an
n summary, the collinear array withh frequency independen
nt beam direection has a a vertical In
beeam stabilityy of ±1° overr the definedd bandwidth, which is wiithin the speecifications. The side lo
obe level is typically 10 d
dB down on the main lo
obe, except ffor at the eddge of the ba
and (960 M
MHz) where iit reaches 7.8 dB. The bbeam width is between 9 ‐ 11° acro ss the defined band. Th
he maximum
m azimuth deviation for the array iss ± 0.7 dB att 960 MHz. TThe azimuth
h pattern deeviation is diiscussed in C
Chapter 3, whhere it was sshown to be caused by ggeometric asyymmetry co
ommonly fou
und in plana
ar element aarrays. Corre
ection techn
niques for thhis azimuth d
deviation haave not been
n implementted, as they w
would resultt in the arrayy being too w
wide for the specified raadome. 5.4.4
Peakk instantane
eous power pperformance
e measured
W
With advanceed radio netw
works using m
multi‐carrierr digital sche
emes, there iis a statistica
al chance th
hat multiple transmitterr carrier pow
wers can combine in‐ph
hase to geneerate a high
h voltage grradient within an array ccombiner/feeed network [2]. This volttage is capabble of destro
oying the in
nsulation com
mponents in an array, inccluding coaxiial cable dielectrics and i nsulators. Fo
or all base sttation arrayss that suppoort multiple ccarrier systems, electricaal acceptance testing sh
hould includee a non‐desttructive high pot test at aa peak level of approxim ately 2.1 kV [2]. This teest places th
he compone
ents under w
working stre
ess and can detects fauulty solder joints or in
ntermittent ccontacts, mo
oisture and ccontaminated dielectrics. Failure to detect these quality po
oints can ressult in early ssystem failurre. 5--159
For non‐destructive device testing, the applied voltage is gradually increased towards the peak voltage whilst monitoring the leakage current as shown in Fig 1‐12. If the current exceeds a pre‐set threshold or an arc occurs before reaching the test voltage the device has failed and the applied voltage resets. The voltage spike associated with PIP is not readily detected by conventional power meters. It is normally seen as noise affecting the bit error rate. The measurement of PIP is not practical, and prevention is preferable. Eqn. 12 can be used to calculate the potential value for PIP based on the power, modulation index and the number of transmitters in use at a transmitter site. The proposed collinear array has been subjected to high pot testing and passed the 2.1 kV limit as shown in Fig 1‐12, that validating the electrical performance of the prototype. (12) M = modulation Peak to Average ratio dB Pca = Power per channel at antenna port in watts N = Number of channels in use Fig 5‐23 Non‐destructive ionisation test of the passive coupler 5-160
5.4.5
odulation pe rformance m
measured Passsive intermo
he PIM of the prototyp
pe collinear array with frequency f
in
ndependent beam direction was Th
evvaluated exp
perimentally (refer to Figg 1‐11 for a schematic o
of this experrimental setu
up). The PIIM Test Brid
dge consistss of two traansmitters combined c
ussing a low PIM diplexe
er set to frequencies that will gen
nerate a prooduct in the
e pass band
d of the recceiver chann
nel. The reeceiver consists of a small amplifierr connected to a spectru
um analyser,, which is filtered to prrevent the fu
undamental TX carriers ffrom de‐senssitising the re
eceiver. Thee PIM test eq
quipment haas residual P
PIM contribu
ution; and thhis is accountted for in the results. Thhe values off PIM are exxpressed as dBc, relativve to the funndamental carrier c
levelss. For exam
mple, if two 46 dBm caarriers are ap
pplied to the
e device undder test, and the measurred level at tthe mixing frrequency is ‐104dBm, th
hen the diffe
erence betw
ween them is the PIM levvel = ‐150 dBBc. Typical vvalues for d ‐160dBc. PIIM are betweeen ‐140 and
Th
he PIM value measured for the proototype was ‐148 dBc, fa
alling just shhort of the desired d
‐
1550dBc. The substrate ed
dge connecttion used in this prototyype was macchined by ha
and from brrass and is not n plated. It is expectedd that when
n this compo
onent is preccision machined and silver plated the PIM value for the prototype collinear c
array with freqquency inde
ependent on will meet or exceed st ate‐of‐the‐art value of le
ess than ‐1500dBc [10]. beeam directio
5.5
Pow
wer handlin
ng perform
mance
Th
he power haandling capabilities of baase station arrays must b
be determineed to ensure
e they do no
ot fail under peak load co
onditions. TThe governingg factors rela
ated to poweer handling a
are: 
ng, coaxial cablee power ratin

dielectric lossses, 
VSWR, 
heat dissipation, 
material theermal properrties, 
duty cycle and peak pow
wer. 5--161
or high pow
wer applications above 2200 Watts at a a frequen
ncy of 800 MHz, the co
onductor Fo
vo
olume must be large in
n order to handle in excess e
of 70
0 amps of RRF current. Ground co
onnections aare normally bonded dir ectly to the mounting tu
ube as it proovides excelllent heat diissipation. One of the prrimary conse
equences of poor powerr handling is a gradual ddrift in the m
measured VSSWR performance. On
nce the VSW
WR reachess approximately 1.8:1, the generatted heat in
ncreases and
d can go intto thermal rrunaway. This VSWR drift can be caused by dielectric d
heeating or meechanical disstortion whi ch detunes the array with increasedd temperatu
ure. The en
nd result can
n be VSWR sh
hut down of a transmitte
er site. 5.5.1
Pow
wer measurement methood he maximum
m power han
ndling capab ility of the ccollinear arra
ay feed netw
work was dettermined Th
byy an actual p
power test. The transm itter source comprises o
of four Andreew feed forw
ward cell exxtender transmitters, phase locked aand combine
ed to deliver a maximum power of 40
00 watts, ass shown in FFig 5‐24. The control coombiner has an inbuilt V
VSWR bridgee connected in series w
with each transmitter. Th
his allows th e monitoring of the VSW
WR perform ance of the antenna un
nder load. A directiona
al coupler c onnected to
o a spectrum
m analyser i s used to verify the fo
orward and reflected power levelss of the com
mbined output. The rreflection co
oefficient m
magnitude is tthen calculatted and recoorded. 5..5.2
Pow
wer test results he power tessting results are summarrized in confiiguration Th
Taable 5‐5 the starting pow
wer level forr the test is approximate
ely 80 Wattss, and is held
d for one ho
our. At thee completion
n of this staage, the tem
mperature re
ecorded by tthe thermal imaging caamera is 54◦ C, or 27°C above ambiennt (27°C). Th
he power is then increassed to appro
oximately 1000 watts for one hour. At this level the array in
nternal temp
perature is m
measured to be 70°C w
which is 43◦ C
C above ambient at the ccompletion o
of the test. If the ambiennt temperatu
ure were to
o reach 40 degrees, this in combinattion with the
e 43◦ C interrnal temperaature would increase 5--162
he net temperature to 83°C. 8
Even aat this high ambient tem
mperature, tthe net temperature th
reemains well sshort of the material worrking temperature of app
proximately 130°C. TX Tx1 TTx2 Tx3 Tx
Tx4 100w 1100
100w 1 00w w
PHASE Ambien
n
Therrmo couple monitors Hybbrid com
mbiner D
Direction
Dire
ectional a power al cou
upler DUT m
meter
Thermal cam
mera Fig 5‐24 Poweer testing configuration
Taable 5‐5 Pow
wer measurement with thhermal moniitoring Forward power dBm Reeflected po
ower dBm Forward Watts po
ower Reflectioon coefficieent magnitu de dB Max recorrded °C Ambient chamb
ber °C abovee ambieent °C Total power appliied Watts ‐30.7 ‐43.2 85.1 ‐12.5
54
27
27 78.8
‐29.5 ‐42.1 112.2
2 ‐12.6
(70)
27
(43) (101.1)
‐28.6 ‐41.5 138.0
0 ‐12.9
100
29
71 127.8
‐27.5 ‐41 177.8
8 ‐13.5
136
29
107 161.9
‐26.6 ‐37.3 218.8
8 ‐10.7
150
29
121 203.2
Th
he VSWR diid not change significanntly during the power testing, andd hence the
e printed co
ollinear arrayy can be rate
ed to a maxim
many low mum of 100 Watts. Thiss rating is suffficient for m
5--163
ower applicaations. The use of the low cost PET substrate limits the ppower ratingg. If the po
su
ubstrate is ch
hanged to Ka
apton or Pol yamide, the power ratin
ng would moost likely incrrease but w
would also siggnificantly increase the coost. 5..6
Mechanical Characteris
C
stics
To
o protect thee collinear array with freequency inde
ependent be
eam directionn against me
echanical sttresses, the rradiator is su
upported byy high densitty polyethyle
ene foam pa ds and enclo
osed in a heeavy duty 76
6mm OD rad
dome with a 6mm wall. The array is mounted onn a low proffile radial flaange affixed to the radom
me, using sil icone sealan
nt and grub screws. Ass most collin
near array an
ntennas are i nstalled high
h above grou
und, they aree subjected tto severe w
weather include large swings in tempeerature strong winds and
d rain. Windd survival speed is an im
mportant figu
ure, as it allows the seleection of an array type a
and construcction that iss suitable fo
or a particular environm
ment. Wind load is calculated as an
n equivalentt flat plate projected p
arrea in accord
dance with TIA329C, annd wind gustt ratings in accordance a
w
with AS1170
0.2:2002. Th
he Torque is the twistingg moment appplied to the mounting hardware. Th
he proposed
d printed colllinear array has been designed to fit f the internnal dimensio
ons of an offf‐the‐shelf rradome. This radome is a 1.9 m secttion of the normal 6.5 m
m stocked length used fo
or high gain ccorporate fe
ed collinear aarrays. As su
uch the mechanical strenngths of thiss radome arre well withiin what wou
uld be requirred for an antenna of th
his size (refeer to Table 5
5‐6). The w
wind gust ratiing for the p
proposed anttenna array ensures it ca
an survive inn excess of 2
240 km/h w
wind gusts witthout failure
e. 5--164
aracteristics
Taable 5‐6 Mechanical cha
Projeccted area 1270 cm
m2 Lateral thrust @160 km/h 160 N
Wind gust rating > 240 km/h Torque @
@ 160 km/h 73 Nm Length 1850mm
m Rad
dome 76mm
Mou
unting 89 mm diam
meter We
eight 3kg 5.7
Fixe
ed beam tiilt array
W
Where an array is mounted high abo ve the inten
nded coverag
ge area, a fixxed beam tilt can be ap
pplied to prrevent interference to other servicces beyond the radio hhorizon allowing for frequency reu
use and efficcient networ k planning. TThere are tw
wo primary ooptions for tilting the m
main beam of an array antenna. A m
mechanical tilt t is normally applied tto directiona
al panels. Th
his is provideed by physically tilting thhe antenna vvia the moun
nting hardwaare design. Electrical beeam tilt can be applied b
by offsetting the phase b
balance in the
e antenna. U
Under this co
ondition, th
he phase fro
ont of the arrray is now forced to a gradient rather than p arallel, the tilt in an om
mnidirection
nal antenna is imposed on the elevaation pattern and distribbuted radiallly in the azzimuth patteern. o achieve electrical tilt, delay sectioons are place
ed in the feed network.. In a corpo
orate fed To
arrray this shifft can be inte
egrated into each powerr divider arm
m, resulting i n lower distortion to th
he side lobe structure th
han when onnly the first branch is offfset, and ass such allow
wing for a grreater tilt range. For a centre c
fed arrray based on series o
fee
ed sub‐arrays
ys the delay is simply pllaced at thee central feed point. Thee upper sub array has a a phase advaance over th
he lower arrray. This ch
hange in phasse has an im pact on the matching cirrcuit and thee side lobe levels, and 5--165
smaller tilt ranges are realized when compared to a corporate fed array. Series fed arrays can be tilted over a wider range but can only hold the tilt angle at a single frequency. In Table 5‐7, a table of collinear arrays with various feed methods and the typical tilt stability over a 10% bandwidth is presented. The worst case is the series fed array where the beam tilt at the lowest frequency is more than 10°. Table 5‐7 Tilt stability characteristics of omnidirectional array feed technologies Frequency (MHz) Centre fed array Corporate fed array Series fed dipole array 850 870 900 940 960 ‐5° ‐4° ‐4° ‐4° ‐5° ‐3˚ ‐4˚ ‐3˚ ‐4˚ ‐3˚ ‐13˚ ‐10˚ ‐8˚ ‐6˚ ‐4˚ A fixed down tilt is created in a centre fed array by applying phase shift at the central feed distribution as shown in Fig 5‐25 (the “Coplanar waveguide (CPW)” discontinuity in the feed network) where the feed slot has been re‐positioned in order to produce a differential offset in the phase. The phase shift magnitude and sign controls the amount and direction of tilt in the main lobe. In the proposed collinear array, down tilt is controlled by the location of the “Coplanar waveguide (CPW)” discontinuity slots relative to centre axis of the array. (13) is used to calculate the required phase delay, and (14) determines the physical length difference needed to present the correct phase offset. A fixed phase offset is used to introduce an electrical down tilt to the array which is independent of frequency. ° 5-166
°∙ ∙
∙
physiicalshiftinm
mm
(13) (14) Zero tilt feeed Feed po
oint offset
for beam
m tilt
Finitee ground CPW
W
Grouund Souurce
Grouund
ed point posiition to conttrol tilt Fig 5‐25 Adjjustment of tthe “Coplanaar waveguide (CPW)” fee
5.7.1
Dow
wn tilt perforrmance he proposed
d planar array can be cconfigured to t provide fixed beam ttilt by adjussting the Th
po
osition of the feed pointt. The limit for beam tilt is the upp
per side lobee level. Too m
much tilt w
will result in aa side lobe e
equal in heig ht to the maain lobe and a loss in dirrectivity assu
uming no otther change in geometryy. Practical ccentre fed arrrays can tiltt to approxim
mately 4◦ from zenith beefore losing pattern stab
bility. To
o show the eeffectivenesss of this methhod, a polar plot of a zerro tilt collineaar array of Fig 5‐26 is co
ompared to the same arrray with 4° down tilt sh
hown in Fig 5‐27 The 44° beam dow
wn tilt is sttable within 4
4 ± 1°, and the first uppeer side lobe is 3 dB down
n from the m
main lobe at 9
960 MHz. A 4° down tilt is sufficient for most basse station ap
pplications. 5--167
Having upset the phase balance, the application of beam tilt normally impacts on the reflection coefficient magnitude performance, as can be seen in Fig 5‐28 for the 4° down tilt collinear array where the reflection coefficient magnitude has been reduced down to 9 dB. The realized broadband gain at beam maximum is also reduced by approximately 1 dB, as shown in Fig 5‐29 It is possible compensate for the mismatch associated with beam tilt by optimizing the pattern shape to reduce the side lobes for a predetermined beam tilt. This can be achieved by reducing the element spacing. Fig 5‐26 Vertical radiation pattern for the zero tilt collinear array 5-168
ollinear arrayy
Fig 5‐27 Vertiical pattern ffor a 4° fixedd beam tilt co
5--169
mpared to arrray with Fig 5‐28 Reflection coefficient magniitude of arrray with 4°beam tilt com
°beam tilt 0°
Fig 5‐29 Gain comparison
n (pattern maaximum) 4°ttilted and 0° tilt array 5--170
5.8
Sum
mmary
n this chapter an omnidirrectional six element “co
oplanar wave
eguide(CPW
W)” fed colline
ear array In
w
was assembleed from the
e planar fee d network, planar dipole elementss and planarr passive co
oupler comp
ponents prevviously discuussed in Chapters 2, 3 and 4. The aarray has the unique atttributes wide band, effectively zeero beam tilt over 10%
% bandwidtth, and low
w passive in
ntermodulation. It has be
een shown inn simulation as well as experimental ly that the “Coplanar w
waveguide (C
CPW)” based
d collinear aarrays main objective of o frequencyy independent beam diirection has been achievved. The raddiation patte
ern and gain of the arrayy are typical for a six ellement centtre fed arrayy. The meaasured reflecction coefficcient magnittude perform
mance is 14 dB acrosss the require d bandwidth
h, although tthe measure d response iis slightly grreater than 1
diifferent to th
he simulation
n due to han d fabrication
n tolerances.. Th
he intermod
dulation performance has been tested t
using
g passive inntermodulattion test eq
quipment. TThe objective
e of low passsive intermo
odulation ha
as not been achieved, fa
alling just sh
hort of the ‐1
150 dBc state
e‐of‐the‐art levels. For low power syystems, a PIM
M value of ‐1
140dBc is accceptable. Itt is expected
d that when tthe coaxial in
nterface com
mponents aree machined tto higher to
olerance and
d silver plated, the speciffied level of ‐150dBc can
n be met. Thhe coupling interface haas also been
n hi pot teste
ed to 2.1 kV
V without faiilure. At this level the aantenna can be used w
where PIP is eexpected to b
be less than 1 kW. Th
he mechaniccal specificattions for thee array have
e also been discussed. The array has h been deesigned usin
ng a radome diameter coommonly em
mployed to ssupport anteennas greate
er than 5 m
meters long. The designed wind loadiing for the prototype collinear array is 240 km/h.. n example o
An
of a fixed dow
wn tilt in thee proposed aarray has been presenteed. The impa
act of tilt on
n the critical parameters such as reeflection coe
efficient mag
gnitude, gainn and side lo
obe level haave been analysed and ccompared too the standard configuration of the ““Coplanar wa
aveguide (C
CPW)” fed co
ollinear arrayy with disturrbance to th
he reflection coefficient magnitude o
observed an
nd a gain red
duction of ap
pproximatelyy 1 dB record
ded. The ma
aximum tilt ffor a centre ffed array is limited by two t
factors, side lobe leevel and reflection coeffficient magnnitude. The side s
lobe leevel must be low to prevent interrference to other services, and th e desired reflection r
co
oefficient maagnitude perrformance m
must be main
ntained to minimize refleection losses that will haave a negativve impact on
n the realizedd gain. 5--171
Chapter 6
Thesis overview a
and propos
sed furtherr work
Th
he objective of this thessis is to inveestigate enhaanced performance colliinear antenn
na arrays w
with effectively zero be
eam tilt veerses freque
ency. The array is too have com
mparable peerformance to the state of the art pllanar collinear arrays. To
o the authorr’s knowledgge, a high gaain omnidireectional arra
ay of greate r than two elements ba
ased on a ““Coplanar wa
aveguide (C
CPW)” centree feed archittecture has nnot been rep
ported. To p
progress thiss development, some gu
uidelines an
nd specifica
ations were drawn against existin
ng tubular metal construction an
ntennas. The complete a
array includi ng dipole ele
ements is to be construccted on a single sided in
nexpensive flexible f
subsstrate. Thee design takes full advvantage of modern ree
el‐to‐reel m
manufacturingg techniquess for large sccale flexible p
printed circuits. Electromagneetic simulatio
on (using CSST Microwave Studio) [40] is the key design method, esentation oof the array performance
e after initiaal circuit sim
mulations. prroviding a reealistic repre
Th
his accounts for the ope
en and comppact architeccture of the feed networrk, which has a small leevel of interaaction with the radiationn pattern. As such the ra
adiation patttern of the a
array is a keey parameteer for conside
eration durinng the design
n of the feed network an d dipole elem
ments. 6.1
Chap
pter 1 Summary - Bac
ckground information
i
n
Ch
hapter one reviews th
he state of f the art in
n collinear array antennnas. Perfformance co
omparisons between the
e fundamenntal collinearr designs (i.e
e. centre fedd, corporate fed and th
he lower costt series fed a
arrays) are p resented. Th
he key appliccations for eeach type of array are given, with a d
discussion on
n the limitatiions for each
h type of arra
ay. 6--172
6.2
Chap
pter 2 Summary - Fee
ed Network
k
w
CPW)” feed network architecture suitable s
for a centre fed planar A “coplanar waveguide(C
ollinear array is presentted in Chappter 2. The feed design satisfies tthe dimensio
onal and co
ellectrical requ
uirements off the projectt. The theorretical circuit schematic of the feed network w
was devised, including all a node imppedances an
nd transform
mer sectionss. The feed network ch
haracteristicss have been analysed ass a planar multi‐port nettwork with tthe aid of CSST Design Sttudio and verified by an e
electromagnnetic model u
using CST Microwave Stuudio. In
n section 2.1, it was show
wn that by ddesigning the
e “Coplanar w
waveguide (CCPW)” main feed for hiigher impedaance the ove
erall width oof the array ccan be reducced. This alloows for a wider track sp
pacing and reelatively narrrower widthh conductorss to be used,, resulting inn an array that meets th
he mechaniccal constraints set by thee selected raadome tube. A secondaary slot tran
nsmission lin
ne was used to distribute
e the energyy from the main “Coplana
ar waveguidee (CPW)” fee
ed line to th
he dipole anttenna elements of the arrray. Th
he limitation
ns of the “Co
oplanar waveeguide (CPW
W)” feed netw
work was disccussed in Section 2.2 reelated to thee low effectiive thermal dissipation when w
compa
ared to convventional me
etal tube co
onstruction. The minimu
um track spaacing of 0.5m
mm imposed
d by standardd reel to ree
el flexible prrinted circuitt processing was also coonsidered a cconstraint off this work. The method
d used to su
upport the feed f
networrk with foam
m spacers is presented with a conssideration fo
or the RF trransparency of this material. A CST Design studio circu
uit model of f the feed ne
etwork has been createed in Section
n 2.3 and po
ort characteeristics were analysed. TThe feed ne
etwork is bro
oken into itss functional sections, an
nd analysed to ensure ap
ppropriate am
mplitude and
d phase distrribution to thhe antenna p
ports. el of the enttire feed nettwork was ge
enerated in Section 2.4 to verify n electromagnetic mode
An
th
he circuit simulations of o Section 2.3. The feed netwo
ork meets aall of the electrical e
sp
pecifications,, including o
operation in the desired frequency b
band, low trransmission loss, and reelatively stab
ble phase pro
ofile. 6--173
6.3
Chap
pter 3 Summary - Pla nar dipole
e
W
Within an eleectromagnetic modellingg environment, the operrational paraameters of the novel pllanar dipolee element proposed p
fo r use in a collinear arrray have bbeen analyse
ed. The trransition from
m a cylindriccal dipole to a planar dip
pole was disccussed, and ffurther deve
eloped to allow for use in the collinear array pproposed in this thesis. It was shoown how this planar n terms of cu
urrent chokinng. diipole is related to coaxially constructted dipoles in
Th
he proposed
d planar dipole is designned to meett the broad bandwidth and omnidirectional paattern speciifications ou
utlined in CChapter 1 by b implemen
nting augmeentation tecchniques, naamely slots aand notches. The notchees are used tto compensa
ate for reactaance associa
ated with th
he close proximity of the dipoles too the edge of o the slotlin
ne feed. Th e slots augm
ment the raadiation patttern to impro
ove directivi ty and reducce azimuth rripple, and too lesser exte
ent assist in
n impedancee matching. The propoosed planar dipole is shown s
to bee capable of o a 15% baandwidth for VSWR < 2:1, whereas tthe conventional planar dipole shapee typically acchieves a 122% bandwid
dth. The no
ovel planar dipole desiggn meets th
he bandwidtth required for base sttation applications, and iis similar to a cylindrical dipole in te
erms of impeedance and radiation paattern performance for tthe same vol ume. 6.4
pter 4 Summary - “Co
oplanar wa
aveguide (CPW)”
(
to M
Microstrip Passive
Chap
coup
pler
In
n an effort to
o improve the mechaniccal stability of the coaxial cable connnection to the feed neetwork, a novel passive
e coupler trransition waas developed
d. A secon dary feature
e of this trransition is co
omplete DC isolation of tthe array fro
om the transm
mission equiipment. Th
he majority of “Coplan
nar waveguidde (CPW)” to microstrip transitionns presented
d in the litterature are aimed at fre
equencies abbove 1 GHz for” monolitthic microwaave integrateed circuit (M
MMIC)” app
plications. In Section 4.2 a literaature review
w indicated that the transition to
opologies geenerally have DC continnuity in at least one off the conduuctors (signa
al and/or grround). 6--174
In Section 4.3 the novel “Coplanar waveguide (CPW)” to microstrip transition is presented. The proposed transition was implemented in order to increase the mechanical strength of the cable connection and eliminate solder processes. This transition has low loss, low PIM, incidental impedance matching, and is totally DC isolated from the input to output transmission lines. It provides sufficient mechanical isolation to enable a robust coaxial cable connection to the thin and flexible PET substrate employed. In Section 4.4 the proposed coupler performance is determined using electromagnetic simulation. The insertion and reflection coefficient magnitude performance is explored with respect to key parameters that affect the coupling between the “Coplanar waveguide (CPW)” and microstrip transmission lines. As the output impedance (of the “Coplanar waveguide (CPW)”) is different to the VNA measuring equipment, a commonly used back‐to‐back measurement technique was explained in Section 4.5. The passive coupling transition performance was then verified by testing a realized prototype of the back‐to‐back structure, with reasonable congruence to the simulated results. An insertion and reflection coefficient magnitude of < 0.5 dB and > 15 dB respectively was achieved over the frequency band of 850‐
960 MHz. The transition assembly was detailed in Section 4.6, which consists primarily of a flexible 0.125 mm PET PCB laminated with the more substantial Arlon AD250 rigid microstrip substrate. The lamination and alignment of the planar substrates was investigated to determine the effect of manufacturing tolerance. The coupling structure was shown to be relatively insensitive to offsets of ± 3 mm. The solder‐less assembly is expected to improve the PIM performance significantly. With complete passive coupling, the transition circuit is capable of coupling efficiently through dissimilar substrate dielectrics and thicknesses. This allows the collinear array described in this thesis fabricated on 0.125 mm thick PET substrate to be easily cabled and inspected during the assembly process. 6-175
6.5
pter 5 Summary - Co mplete arrray perform
mance eva
aluation
Chap
In
n Chapter 5 an omnidire
ectional six eelement “coplanar wave
eguide(CPW))” fed colline
ear array w
was assembleed from the
e planar fee d network, planar dipole elementss and planarr passive co
oupler comp
ponents prevviously discuussed in Chapters 2, 3 and 4. The aarray has the unique atttributes wid
de band, efffectively ze ro beam tilt over > 10
0% bandwidtth, and low
w passive in
ntermodulation. It has be
een shown inn simulation as well as experimental ly that the “Coplanar w
waveguide (C
CPW)” based
d collinear aarrays main objective of o frequencyy independent beam diirection has been achievved. The raddiation patte
ern and gain of the arrayy are typical for a six ellement centtre fed arrayy. The meaasured reflecction coefficcient magnittude perform
mance is 14 dB acrosss the require d bandwidth
h, although tthe measure d response iis slightly grreater than 1
diifferent to th
he simulation
n due to han d fabrication
n tolerances.. Th
he intermod
dulation performance has been tested t
using
g passive inntermodulattion test eq
quipment. TThe objective
e of low passsive intermo
odulation ha
as not been achieved, fa
alling just sh
hort of the ‐1
150 dBc state
e‐of‐the‐art levels. For low power syystems, a PIM
M value of ‐1
140dBc is accceptable. Itt is expected
d that when tthe coaxial in
nterface com
mponents aree machined tto higher to
olerance and
d silver plated, the speciffied level of ‐150dBc can
n be met. Thhe coupling interface haas also been
n hi pot teste
ed to 2.1 kV
V without faiilure. At this level the aantenna can be used w
where PIP is eexpected to be less thann 1 kW. The proposed co
ollinear arrayy has been subjected to
o high pot testing t
and passed the 2.1 kV limitt as shown in Fig 1‐12,, that valida
ating the ellectrical perfformance of the prototyppe. Th
he mechaniccal specificattions for thee array have
e also been discussed. The array has h been deesigned usin
ng a radome diameter coommonly em
mployed to ssupport anteennas greate
er than 5 m
meters long. The designed wind loadiing for the prototype collinear array is 240 km/h.. An
of a fixed dow
wn tilt in thee proposed aarray has been presenteed. The impa
act of tilt n example o
on
n the critical parameters such as reeflection coe
efficient mag
gnitude, gainn and side lo
obe level 6--176
aveguide haave been analysed and ccompared too the standard configuration of the ““Coplanar wa
(C
CPW)” fed co
ollinear arrayy with disturrbance to th
he reflection coefficient magnitude o
observed an
nd a gain red
duction of ap
pproximatelyy 1 dB record
ded. The ma
aximum tilt ffor a centre ffed array is limited by two t
factors, side lobe leevel and reflection coeffficient magnnitude. The side s
lobe r
leevel must be low to prevent interrference to other services, and th e desired reflection co
oefficient maagnitude perrformance m
must be main
ntained to minimize refleection losses that will haave a negativve impact on
n the realizedd gain. 6.6
Conc
clusion
A high perforrmance centre fed collinnear array with w planar architecture hhas been prresented. Th
he modelled
d and measured radiatio n patterns sshow good congruence w
with a gain o
of 10 dBi an
nd beam tilt less than on
ne degree ovver a 10% bandwidth. A passive interrmodulation
n value of ‐1148 dBc has been achievved due mainnly to the re
educed numb
ber of electrrical joints. A
A PIP test haas shown thaat the transm
mission line ccan withstan
nd an ionisatiion test at 2..1 kV. Th
his is as a ressult of increa
asing the linee impedance
e and minimu
um track spaacing >0.5mm
m. Th
he design co
onstraint where the arra y must fit an
n existing 76
6mm diametter radome h
has been m
met. The arraay width has been restrricted to 32
2mm. The pa
assive couplling circuit has h been efffective with < 0.5 dB insertion loss w
with reflectio
on coefficientt <‐14 dB oveer 10% band
dwidth. 6.7
posed future work
Prop
Arreas of furrther develo
opment for the enhan
nced perform
mance collinnear antenn
na array prroposed in th
his thesis are
e based on thhe requirements for futu
ure commun ications netw
works. 6.7.1
Furtther develop
pment to impprove radio site efficienccy Ass major neetwork pro
oviders aim to increase the cap
pacity for high speed
d digital co
ommunicatio
ons and keep
p ahead of ddemand, the requiremen
nts for smalleer cell sizes and high deensity lower power mesh
h networks aare likely to increase [3]. Th
his current d
developmentt in printed ccircuit based collinear antennas allow
ws for more e
elements to
o be fed with
hout increasiing the widthh of the asse
embly. This p
passive couppled architectture may also be developed to pro
ovide multibaand perform
mance as the
e CPW transsmission line
e is not a nsitive part of the arrayy. Shorter slot lines cou
uld independdently suppo
ort other frequency sen
nds interleavved or in serries. In this ccase, the cha
allenge will be to reduce
e mutual frequency ban
oupling detuning effects.. co
6--177
n the directivve gain is alsso possible as more elem
ments are addded. The cha
allenge is n increase in
An
to
o provide a suitable matcching circuit and further improve the phase stabi lity. Byy increasing the numberr of fed elem
ments to 12, results in a directivity oof 15 dBi. This centre feed planar deesign allows for the exppansion to greater numb
bers of elem
ments an exa
ample of w
which is show
wn in APPEND
DIX E and raddiation patte
erns in APPEN
NDIX F throuugh to M. H
Higher gain antennas with low ppower transmitters can improve the powerr supply co
onsumption of a radio site, interferrence incidents can be re
educed by ddecreasing th
he power off the transmitters and increasing thee gain of the antenna. Th
he design can be scaled tto higher bannds where th
he physical size can be reeduced. Th
he antenna b
bandwidth o
of the array ccan also be sscaled to anyy frequency rrange betwe
een up to 1..2 GHz. The operational frequency iis limited byy the minimum track sppacing of the
e printed circuit fabrication technique at 1.2 GH
Hz. 6.7.2
Beaam tilt contrrol using new
w technologyy. n chapter 5..7 there is an examplee of how beam tilt can be implem
mented in a planar In
arrchitecture array. a
This was shown in modellin
ng by shifting
g the feed ppoint relativve to the ceentre axis. Reecently therre have been
n some encoouraging devvelopments into voltagee variable dielectrics. Th
hese materiaals are poten
ntially able too provide a remotely varriable phase shift which could be ussed to contro
ol phase offsset within th e array ante
enna therebyy controlling the amount of beam tillt. Currentlyy the voltage required to shift the die
electric consttant effectiveely is >=5 kV
6.7.3
Beam direction control to a llow for freq
quency reuse
e An
nother tool u
used by servvice providerrs to increase
e the capacitty of a servicce is frequency reuse. Th
his can allow
w a service to
o occupy thee same chann
nel frequenccy in a neighbbouring cell. This can bee provided b
by restrictingg the beam ddirection or by controllin
ng the elevaation beam a
angle. An exxample of beam directio
on control iss shown in APPENDIX
X M with thhe radiation patterns sh
hown in APPENDIX O to. This could bbe of particu
ular use to sspatial diverssity schemess such as M
MIMO servicees. A numbe
er of elemen ts can be mo
ounted to prrovide de‐coorrelation suiitable for M
MIMO applicaations. 6--178
An investigation into the use of high temperature substrates and thermal sinking in the design would allow for a power rating of up to 250 Watts. 6-179
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APPENDIX A The topology of the lower coupling and second dipole of the planar collinear array 6-185
APPENDIX B The topology of the central feed point and inner two dipoles of the planar collinear array 6-186
APPENDIX C The topology of the second last and last dipole in the planar collinear array 6-187
D The compllete planar ccollinear arraay topology
APPENDIX D
6--188
APPENDIX E First model trial of a ten element array APPENDIX F The broad band gain response of a ten element array 6-189
APPENDIX G The ten element “Coplanar waveguide (CPW)” array at 840 MHz APPENDIX H The ten element “Coplanar waveguide (CPW)” array at 850 MHz 6-190
APPENDIX I The ten element “Coplanar waveguide (CPW)” array at 860 MHz APPENDIX J The ten element “Coplanar waveguide (CPW)” array at 870 MHz 6-191
APPENDIX K The ten element “Coplanar waveguide (CPW)” array at 880 MHz APPENDIX L The ten element “Coplanar waveguide (CPW)” array at 890 MHz APPENDIX M The “Coplanar waveguide (CPW)” panel array (850‐960 MHz, 15 dBi) 6-192
APPENDIX N The reflection coefficient for the “Coplanar waveguide (CPW)” six element panel array APPENDIX O The azimuth pattern for the “Coplanar waveguide (CPW)” six element panel array 6-193
APPENDIX P The elevation pattern for the “Coplanar waveguide (CPW)” six element panel array APPENDIX Q The broad band directivity response for the trough reflector array APPENDIX R The spacing from the reflector to the array for the trough reflector array 6-194
S(11) dB APPENDIX S The angle of the trough reflector APPENDIX T The reflection coefficient for omnidirectional and unidirectional array Reflection coefficient
6-195